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Tag: 2N7000

One of the facts which I’ve been blogging about, concerns the software called ‘NG-SPICE’, which stands for ‘Simulation Program with Integrated Circuit Emphasis’. When using it, I can sometimes seem to recognize the parameters with which it defines components, those parameters will correspond to concepts which I already know about – such as capacitance, or ‘transconductance’ – but in the modern context, those parameters can stand for the real-world properties of a MOSFET, that do not match old-world properties, by the same name.

So as an example of this, I can demonstrate the following ‘SPICE’ definition, of an old-fashioned, discrete MOSFET, which was named the ‘2N7000′:

In this specification there is a parameter named ‘CGSO’ , of which I might say, ‘It stands for Gate-Source capacitance’. Its value is close to 1.79·10-7 . If this value was in Farads, it would actually mean that the transistor to be modeled has 180nF input-capacitance, when not active. This would be a very high capacitance-value, which in turn, would lead me to think that the transistor-type was of very low quality. But in reality, this parameter is in Farads /Meter. And so what I would now think, after noticing this detail, is that because the transistor in question only has a width of 100μM, its passive input-capacitance is really only 18pF.

The same goes for ‘CGDO’.

This will make a huge difference, in terms of how fast circuits can become, that use this transistor. And I am back to having faith, in how NG-SPICE simulates its circuits.

At the same time, when such a transistor is functioning in an active circuit, it will exhibit the property of voltage-gain indirectly. This voltage-gain will reduce the apparent capacitance, between the Gate and Common, because it reduces the voltage-changes between the Gate and the Source of the transistor, which determine how much current will flow due to its real Gate-Source capacitance, with respect to how large the changes are, in Gate-to-Common voltages. However, this phenomenon needs to be held in suspicion, if we are expecting it to reduce input capacitance by more than a factor of 1000, because:

The real voltage-gain of the transistor-circuit is only rarely that high, and

There are additional factors which will increase input-capacitance, so that such an ideal will not be reached.

The transistor needs to be a fast one, with low Gate-Source capacitance, if that’s what the circuit-designer is looking for.

One of the facts which I’ve been writing about, is that I possess the open-source version of ‘SPICE’, that is named ‘NG-SPICE’, and that this acronym stands for ‘Simulation Program, with Integrated Circuit Emphasis’. The full, associated suite of programs allows me to edit schematic diagrams graphically, but to export ‘Netlists’, so that I can then simulate the circuit – and see if it works.

And one of the facts which I have also been contemplating, is that by default, SPICE will put transistors, which correspond to micron-sized transistors, which will therefore never be able to drive output-loads, from a hypothetical IC, unless an explicit attempt is made, to design output-buffers, which can. These output-amplifiers have as function, that they should merely follow their input voltage, but draw as little current from their respective inputs as possible – that are outputs of other, more interesting ICs – while allowing low load-resistances to be connected to their own outputs, which correspond to plausible external components, such as 100Ω load-resistors.

I had posted an earlier, conceivable design, of such an output-buffer, which had a major flaw, that I also pointed out in the preceding posting: That amplifier could only produce a range of voltages, which was a direct function of what the Gate-Source threshold voltages would be, of the component transistors used. Hence, because I had also specified low-quality, outdated MOSFET transistors with high threshold-voltages, the output-voltage-range, was also modest but reasonable. But, newer transistors will have lower threshold voltages by design, which would, oddly enough, reduce the voltage-range of that amplifier. This would be an important consideration if the transistors were not in fact discrete, but needed to be incorporated onto the IC, where low-threshold-voltage transistors are already standard. Which means, that I needed to design a better output-buffer.

So below is a better output-buffer, schematic:

And these are the SPICE definitions, of the discrete transistors which I decided to base my design on again, both enhancement-mode MOSFETs:

The main disadvantage of this latest design would be, that the transistors which I labeled ‘X2′ and ‘X3′, do in fact conduct current to their combined inputs, which makes the additional transistor ‘X1′ necessary, since this amount of current would already be excessive, to connect to an output, of any pre-existing IC circuits. But then, the advantage goes so far, that ‘X2′ now models a level-shift, which exactly mirrors the level-shift of ‘X4′, and the voltage-level-shift of ‘X3′ now mirrors ‘X5′. There is design beauty in this. But one disadvantage now is, that the Gate-Source threshold-voltage of (1) n-Channel MOSFET (2.2V) plus (1) p-Channel MOSFET (3.2V) gets subtracted from the input-voltage, so that the available voltage-range still suffers, with respect to both the supply, and the input-voltage. Input-voltage now ranges from 5.4V to approximately 12.5V, which is closer to the range of supply-voltages than what the previous circuit allowed, and the resulting output-voltages are graphed below:

(Update 06/20/2018, 0h20 : )

There is another observation which I should add:

In the days of vacuum tubes, ‘transconductance’ was measured in Amperes / Volt, and was therefore given in ‘Mhos’, which were the reciprocal of Ohms. Apparently, in modern days, the transconductance of a MOSFET, also given as its ‘KP’, is in Amperes / Volt2 . This conscious design-decision must follow the real-world behavior of MOSFETs, but makes my earlier Math, of multiplying such a component-property by the series-resistance, to arrive at gain, incorrect. Gate-Source voltage-changes lead to current-changes, but greater Drain-Source voltages, lead to greater current-gain. This is good, because the actual gain of a MOSFET, reduces the apparent capacitance at its Gate.

In This earlier posting, I described how I can use the program ‘NG-SPICE’, along with its associated GUI-applications, to create and edit circuit diagrams, and then to simulate approximately how those would function if actually implemented as circuits.

‘SPICE’ stands for ‘Simulation Program, with Integrated Circuit Emphasis’. And one of the facts which I had lamented in the posting linked to above, was that it tends to produce default transistors, that correspond to transistors one would find on an IC, those having sizes of 1 micron and smaller. Therefore I faced the challenge which I believe professional IC-designers also face, which is, to make circuits small that contain many transistors, yet, to drive output-pins of the ICs in such a way, that at least, discrete components can be connected to those output-leads, without doing so collapsing the output-waveforms.

But, my hypothetical solution to this problem will be different, from the professionally-chosen solution. What I think gets done commercially, is that special output-transistors are put on the IC, that take up greater surface-area there than unit transistors do, and which are difficult to diagram in a schematic faithfully, to how schematics depict discrete-transistor circuits. My solution was to accept the legacy n-Channel, ‘2N7000′, and the legacy p-Channel, ‘BS250P’, discrete, enhancement-mode MOSFET transistors, in the spirit that bipolar-transistor solutions are to be avoided, if the main chip-technology was MOSFET, but to mark these components as being ‘special’ – even though ages ago, the discrete components would have been regarded as ‘normal’, while the IC components would have been regarded as ‘special’, I defined those discrete components within NG-SPICE, as subcircuits, which are therefore labeled with descriptors that begin with the letter ‘X…’ , as opposed to native MOSFETs, that are to be labelled with the letter ‘M…’ .

Here are the subcircuit, Netlist-definitions of these two, now-defunct transistor-types:

I understand that this was to be an exercise, at amplifying the output of a hypothetical IC, as if with discrete components that would need to be connected to the IC externally, but remember, that equivalent, low-quality MOSFETs can be incorporated into any chips which are to be manufactured, so that in the event that they are, external components would not be necessary.

I also understand that my schematic is imperfect, in that it creates voltage-level-errors, but that such errors would become irrelevant, if the output-amplifier was put inside a feedback-loop, that has high, open-loop gain, as such a feedback loop would also just correct the voltage fed in to the circuit of this posting, so that correct output-voltages nevertheless follow.

This is the schematic:

One reason for the voltage-level errors is the fact that in this schematic, ‘X1′ and ‘X3′ do not match, and that equally, ‘X2′ and ‘X4′ do not match. Even though ‘X1′ and ‘X2′ were meant to model the threshold-voltage, with which each transistor turns on, and then to apply the correct level-shifts to the gates of ‘X3′ and ‘X4′, the main problem in my design is, that ‘X1′ is of the p-Channel type, while ‘X3′ is of the n-Channel type, and vice-versa. There was never any guarantee, that the positive threshold-voltage of the n-Channel transistor, exactly mirrors the negative threshold-voltage, of the p-Channel transistor! But, unless such an amplifier is to place a load-current onto the input-terminal, I found no simple way to avoid this situation. As it stands, the input-terminal would offer minimum load on the circuit designed in the posting linked to above, because all it drives is floating gates. All this does is add some capacitance, while the driven output-resistance of 100Ω, is a direct function of the fact that the chosen, discrete MOSFETs, were only able to handle 200mA maximally.

Interestingly enough, the direction of the level-shift between ‘X1′ and ‘X3′ is the same, as the direction in which ‘X2′ and ‘X4′ shift the voltage-levels, which means that ‘VSD(X1)’ + ‘VDS(X2)’ == ‘VGS(X3)’ + ‘VSG(X4)’, for which reason the circuit can be expected to work anyway. And, neither the Source-Drain current through ‘X1′, nor that through ‘X2′, ever reverse, over the operating range.

One subject which I wrote about in an earlier posting, was that software exists by the name of ‘SPICE’, which stands for “Simulation Program with Integrated Circuit Emphasis”. There are several variants of this software in existence, but the version which I am focusing on for now is the Open-Source ‘NG-SPICE’ system, which needs to be bundled with numerous other packages under Linux, really to be useful. One important package is ‘ngspice-doc’, but there is a whole suite of Linux packages referred to as ‘gEDA’.

Simply having tested a few demo-projects, is not the same thing as actually having designed a circuit, and having witnessed that project ‘work’, at least according to the simulation. Just last night, I did the latter, in order to get a better, working grasp of how to use the software, and also, some idea of the sort of error messages and problems which invariably occur on a first-time basis. What this means is that I actually designed a circuit using the ‘gEDA Schemtic Editor’, which is also known as ‘gschem’, and then ran multiple simulations of the circuit, discovering at first that it had performance issues as I had imagined it, modifying it numerous times, and ending up with a version of the circuit, which I could be satisfied with for now.

The circuit which I was designing, actually involved MOSFETs, because those are the most important components in circuit-design today, and surely enough, I did run into initial problems. One of the tasks which we must complete, when using active components in SPICE, is to define the component, which is as fundamental as the fact that we also don’t just put a resistor, but must also specify what the Value of the resistor is in Ohms. Well with active components, we must do something similar, which also goes under the GUI heading of the Value attribute for the component. Therefore, MOSFETs, be they NMOS or PMOS, also have values, and by default, those values are defined by a Model Card, from which the computer can predict such physical properties about the NMOS or the PMOS transistor, as what its gate-capacitance is, how well it conducts when switched to conductive, conversely when the gate-voltage is zero, etc., etc., etc..

But, because NG-SPICE (v26) is advanced software – though still not the latest version – it may not require that the user defined all these parameters each time he or she considers designing a circuit, because standard component specifications exist.

By default, our MOSFETs have Reference Descriptors that begin with the letter “M”, and not with the letter “Q”, which would stand for a Bipolar Transistor, but which the GUI of ‘gschem’ suggests for the user when he first clicks a MOSFET into his circuit. So we override that, by editing the RefDes into a text-string that has the letter “M” followed without spaces by a number.

What I next proceeded to do, was to put MOSFET-transistors into my circuit, which from the GUI, only had 3 pins. This is a common way in which MOSFETs are often diagrammed, and looked something like this:

Believing that I could just accept what the GUI had constructed, I next tried to simulate the circuit, and received the error, which roughly stated “Unable to find Definition of Model.” This error-message wasted much of my time trying to solve, because I had in fact created a Model Card for the transistors which I was going to use, and at first, I despaired that NG-SPICE might not be as good as paid-for software. But I soon learned that indeed, the following example is a sufficient Model Card for an arbitrary NMOS transistor, with which circuits can be designed:

In actual circuit-design, we’d drop the .TXT Filename-Extension, that makes the above examples readable in a Web-browser. Not only that, but we can also use the ‘gschem’ GUI, to embed such definitions directly into the Netlist, by giving them as a ‘Model’ attribute. So what was causing this error message, in my example? The fact is that MOSFETs are 4-pin components by nature. They have a hypothetical Source, a hypothetical Drain, a Substrate Electrode, and a Gate. It’s the voltage between the Gate and the Substrate Electrode, that finally determines how conductive the MOSFET is to become. By convention, many practical MOSFET-packages tie the Substrate Electrode together with the Source lead, which also happens to make the Source different from the Drain.

By telling NG-SPICE that we’re including a ‘MOSFET_TRANSISTOR‘ in our circuit, we’re telling this program to read 4 Nodes from the Netlist, to parse what the transistor is to be connected to. But, when the GUI only provides 3 arguments, an error ensues, that garbles the attempt of NG-SPICE to parse the Netlist. That’s all. Curiously, the reverse error does not happen. If I conjure a 4-lead MOSFET-symbol from the GUI, but specify a 3-lead MOSFET (more on that below), then I obtain a well-managed error message, that tells me what the problem is.

(Updated 06/14/2018 … )

Actually, the symbols above are also different in another way. In theory, one stands for an enhancement-mode, and the other, for a depletion-mode transistor. But, because under Linux, ‘this software is divided into two departments’, effectively, this does not matter.

The GUI allows schematics to be drawn in such a way, that Netlists result, while the actual NG-SPICE software emulates what these Netlists define. It’s in the emulation of the Netlists, that the decision is also made, as to whether a component is an enhancement-mode or a depletion-mode, or a subcircuit component… ( :1 )