QRP HomeBuilder - QRPHB -

Sunday, 27 November 2016

Radio astronomy requires a receiver output data stream to make sense of what you receive. Computer programs may convert the data into graphs, FFTs, spectrograms, or pool it with other data in hopes of digging the signal out of the noise. At this point, I'm happy with a simple X-Y graph and hopefully my methods will evolve over time.

After much reading + thinking and some hardware failures, I settled on a commercial product to stream data into my computer via a USB cable and then use homebrew + commercial programs to massage my data.

The on-board USB-to-Serial Bridge Controller = a Prolific PL-2303 HX. Click for Prolific's web site. The PL-2303 contains an on-chip clock, so I put the board in a RF shielded container fashioned with 2-sided copper clad board. Unlike some ADCs sold today, the PL-2303's drivers were MS Windows certified; they installed and worked on my first try.

Above — I carved out a hole for the USB, Type B connector with a hammer and chisel before soldering it to the rest of the enclosure. My USB cable has ferrite beads on both ends. On the RCA input port, I placed a 100n bypass capacitor.

I'm currently only using Channel 1 of the ADC, so the other 3 inputs are shorted to ground.

Data Streaming and Code

To run the ADC--4U15, you start a supplied application that establishes communication to the ADC and an output number between 0 and 32767 ( 15 - bits) appears for each of the 4 data channels.

Above — The ADC-4U15 controller application. Once I plug the ADC to its USB port and start this application, I normally hide it -- and it runs in the background.

Above — setup dialog box for the ADC-4U15 controller application. Once you set the com port, sample rate and maximum voltage of your data channel(s), it's set and forget. This application provides a data logger output, but you don't have to use it. EECI provides some source code to get you going in both C# and VB. The ADC-4U15 application is a combination user interface and USB driver -- you read the 4 digital channels from a Windows memory label

In Microsoft Visual Studio ( free version) I followed Bert from EECIs instructions and created an application that read the Windows memory map and it compiled without errors!

Then it's up to you to add code to convert the 2 byte pairs of each channel into a 16 bit integer -- and then further, to convert it to some usable unit to graph. Of course, you'll need to write code to write to file -- or graph within your application. I chose the former for now.

I kept my units as DC volts since this allowed me to test and calibrate my receiver system to ensure my code was working. At some point when I get my noise figure system going, I'll plot the Y axis in degrees Kelvin.

using System;
using System.Collections.Generic;
using System.ComponentModel;
using System.Data;
using System.Drawing;
using System.Linq;
using System.Text;
using System.Threading.Tasks;
using System.Windows.Forms;
using System.IO;
using System.IO.MemoryMappedFiles;

Above — Channel 1 is expressed as a DC voltage. All other channels are shorted to ground and read 0. Once the app starts up, elapsed time gets displayed. The only purpose of this app is to read the data from memory, massage the data bytes into something usable, and write that data to a CSV file in a format that meets my requirements. The file name = the current timestamp.csv. This allows me to keep track of the CSV files by a unique namesake.

Above —The 6 column CSV file allows me to graph in UTC or local time. Microsoft Excel seems to suffer some time stamp issues, so the source code was adjusted to provide an accurate output for my location.

Above — a test plot of my Jupiter receiver during the early evening when my local noise is louder than after midnight. I've calibrated my receiver's DC output voltage against known amplitude input signals from -121 to - 47 dBm and wrote some of them on the graph in red.

My graphing process will likely change over time. It's nice to finally have data to work with.

Sunday, 13 November 2016

After many experiments and 4 versions, I settled on my simplest design.

The AD8307 splendidly responds to input signals from DC to ~ 500 MHz, so it can work well for a DC or ZIF receiver. However, any pulsing noise on its positive rail, or at the input may get converted to a DC output and wreck your log response and accuracy. Not to mention -- any detected RF will likewise wreck your day.

Above all, I learned it's essential to put it in a metal box and filter the DC rail and input heavily. My analog meter registers a receiver input signal from - 121 dBm ( S1 ) to - 43 dBm ( S9 + 30 dB ) Note: measures made with a low noise + distortion continuous wave available power connected to the 50 Ω port of the filter / preamplifer with the receiver tuned to generate a 1 KHz AF tone at the receiver AF output.

Above — Final schematic of my Jovian logarithmic amplifier module.

DC Filtration

I'll first discuss the DC filtration. In past log amps featuring the AD8307, I ran battery power to eliminate any noise on the DC rail. In other versions, I added active ripple filters and for this project compared the results to just using big capacitors to filter the AF ripple. I measured no difference and opted for the 470 + 1000 + 470 µF shown. Further, careful RF filtration delivers reasonable attenuation of RF signals from HF to UHF.

Recall that when we combine 2 RF caps in parallel , the inductance of 1 capacitor resonates with the capacitance of the other to form a parallel resonance — leading to a high impedance — that wrecks RF bypass at a specific frequency, although this peak will also exhibit skirts like any band-pass filter. Hence, I've placed either a resistor, or a choke in between the shunt 1n or 100n RF capacitors.

Input Signal Filtration

My inspiration came from Gary, NA6O ( ex-WB9JPS ) who penned An Accurate S-Meter for Direct Conversion Receivers for QST, February 2008. I recommend his article as a gold-standard reference on this topic. With Gary's permission, I copied his fast attack, slow delay post AD8307 detector circuitry, and the 5 volt supply rail-rail op-amp scheme. Further, Gary nailed the problems ailing the AD8307 at AF with particular insight.

I ended up making 4 circuits and settled on the 1 presented above. This, an experimenters circuit, and likely requires your own nuance. Based on my AD8307 projects, much discussion and emails over the years, accurate log amps may prove thorny and difficult for some builders.

Gary found common-mode noise afflicted his design when receiving small amplitude signals.
He opted for a front-end differential amplifier to reduce this noise while providing adjustable gain to optimize the AD8307 input amplitude to any direct conversion receiver with known gain to get the best possible dynamic range. I found his conclusion about needing variable gain true plus invaluable. Gain adjustment can be done in the receiver, or the log module; or both.

Further, employing a low noise, rail-to-rail op-amp proved vital.

Let's quickly review common mode noise in differential op-amps since sometimes when I discuss this, the other builder's eyes glaze over. Differential amps done right reject common mode noise. In differential op-amps, the inverting [-] and non-inverting inputs [+] are driven 180 degrees out of phase. Since unwanted noise or hum appears in-phase on each input ( common mode ), this noise/hum gets rejected and does not appear at the op-amp output. Of course, you need to 1% match the input circuit capacitors and resistors to yield exemplary common mode suppression in an op-amp filter.

In my lab, with a differential amp/filter I obtained good results, but still, my AD8307 lacked sensitivity to measure signals when I applied < -110 dBm to the front end of my single-signal receiver. For my particular receiver, I found that brute-force R-C filters worked better.

Simple R-C filters get a bad rap; likely because for a first order filter the roll off is only 6 dB per octave and we get so much better with our popular active filter designs.

The 0.33 µF series cap provides some hum immunity into the op-amp unity-gain buffer. An identical 68 Ω R ( labelled RF in the schematic ) plus a 3.3 µF polyester low-pass filters the buffer's output. If your receiver output looks pretty noisy, you may increase RF to as high as 200 Ω to drop that filter's 3 dB cutoff point. Brute force indeed. Set the buffer input and output R-C filter values to what you require.

With the analog meter calibrated with a signal generator, and the zero and calibration pots adjusted correctly, I can measure down to -121 dBm or S1. With a voltmeter, I measured signals down around -130 dBm before getting clobbered by the receiver noise floor. Since my
receiver monitors low-level Gaussian noise for radio astronomy, I wanted the meter sensitivity focused towards the low end.

Calibrating my analog S - meter proved easy. I set a calibrated signal generator to 20.190 MHz and tuned the receiver for a 1 KHz AF output signal. I injected - 73 dBm or S9 and adjusted the meter cal pot to go about 60% of my particular panel meter. Then I decreased power by 6 dBm to find S8 and so on. When satisfied, I had the range correct. I marked my meter with a black marker for S1- S9 -- and then S9 + 10 dB ( -63 dBm) , S9 + 20 dB and then S9 plus 30 dB. I tested the meter by inserting in-line 6 and/or 10 dB attenuators and so forth. It check out and my meter marks account for any non linearity in the AD8307 circuit.

This gives you a calibrated measurement receiver. I love the S-meter -- and Gary's fast-attack, slow delay detector further boosts the experience. With it, no AD8307 output DC low-pass circuitry is required ( see the datasheet ).

The other output goes to a panel jack for a connection to the an ADC for data acquisition. The 2K7 shunted pot will help calibrate the noise power output on a graph.

Above — First "real world" test with analog meter before I painted its face white and then after, marked the S units with a felt marker in a calibration set up. When I close the -6 dB switch on my preamplifier - filter module, the meter drops by 1 S-unit. I'm happy!

I've attached 2 of my discarded circuits for experimental fodder.

Click for Module 8, the ADCClick to return to the main index for this project

After re-reading Chapter 9 of Experimental Method in RF Design (now out of print), a sketch arose and I ran to the bench to make some sparks. I closely followed much of the circuitry designed by Chapter 9 author Rick, KK7B and also incorporated stuff I've learned from Wes, W7ZOI.

Above — First real-world test of the new (unlabelled) receiver in my modular radio astronomy system. 1 less noise side band to deal with makes me happy.[1] Input filter, mixer/amps and AF phase shift networks

Above — Schematic Part 1 shows the design from the RF input to the outputs of the I and Q channel arm all-pass networks.

I opted to in-phase split the RF signal with the popular, broadband, 3 dB hybrid splitter because it's simple and worked well in past adventures.

Above — A sweep of the input filter + 3 dB hybrid splitter after I affixed 2 temporary SMA RF connectors and connected a 51 Ω shunt resistor across the unused output port. The measure is what really matters in RF design ! After a little tweaking of the inductors by spreading or squishing the windings, I tweaked the stage's attenuation to 3.22 dB; showing my match and gain at ~20.1 MHz look great.

I wound the coils on a T37-6 and a T44-6 toroid with # 26 gauge wire.

Does omission of a diplexer and just going with a 50 Ω high-pass shunt network + an RFC seem reckless? Perhaps, just frugal. I stuck with the post mixer AF common base preamp popularized by Roy Lewallen, W7EL and promoted by Rick, Wes and others.

Above — Emitter current on my common base post-mixer AF amplifiers. 26/IE = 50 Ω. I felt jubilant I obtained this with 5% resistors. I matched my 2N4401 transistors. From a DC and AC perspective, each channel resembles the other 1. The 12K collector resistor establishes a
collector DC voltage of 5.95v -- this provides about 1/2 VCC virtual ground bias for the op-amp chain that follows each common base amp.

The final DC output voltages -- and AC test signal voltages ( when I injected an AC test signal to the input of each arm ) were within 2% of each other for the 2 arms. In a ZIF receiver, we want such parity to glean optimal opposite side band suppression. Apart from phase, the I and Q channels should look like each other when viewing with instruments and your physical layout.

The op-amps U1-U4 follow Rick, KK7B's designs. I chose Rick's all-pass network R and C values for 2 reasons. [1] Since 1995, this design enjoys constant replication and proof of function via measurement by many builders [2] I collected these 1% metal film resistor values and had them on hand. Our best circuits are always built using whatever parts we've got on hand.

These network values offer the potential for 60 dB opposite side band suppression from 270 - 3600 Hertz. As it turned out, I did OK --- at a 1 KHz offset I measured upper side band suppression at 55 dB! My best effort to date.

I've designed all-pass filters, but lack the 1% resistors to build them right now.Click for my blog page covering 90 degree phase shifts.

[2] I and Q channel summer, AF filter/amplifiers and AF splitter

Above — Schematic Part 2.

A 1K trimmer allows you to set whatever ratio of the I or Q channel needed to optimize side band suppression. A 470 Ω AF gain trimmer pot allows the receiver gain to vary from ~ 44 - 61 dB. I normally choose around 50- 54 dB gain and then make up the additional needed AF gain with the AF power amp -- or gain control circuitry in the log amplifier system.

1 op-amp virtual ground establishes 1/2 VCC on all the op-amp + pins. In my first Jupiter receiver, I chose 6 poles of low-pass filtration with a 3 KHz cut-off. In this design, I opted for 4 poles with a 1 KHz 3dB cut-off. Note the filter response = Gaussian - 12 dB. The meandering G-12 dB group delay sounds the best to my ears -- and never rings. I learned this from Wes. Choose a lower cutoff frequency, less poles and make it a Gaussian response for great sound. I also do this for crystal ladder filters. We're listening to galactic noise and not competing in the CW sweepstakes after all.

The signal gets bisected by U7. 1 signal goes to the AF amp module, the other to the log amp for output into an S-meter, plus an ADC.

Above — The output of both AF line outputs @ ~ 802 Hz offset with -73 dB available RF power at the receiver input during some testing. 1 bench cable was long, the other short and this resulted in a slight phase shift!

Above — Serendipity. I bench tested the quadrature hybrid from 20.1 to 20.3 MHz and the phase shift was exactly 90 degrees. The amplitude difference gets cancelled with the balance trimmer in schematic Part 2.

Above — Full on bench mode. Temporary RF and BNC connectors allow situ testing. With extensive test as-you-go construction, we usually suffer no surprises when placing your completed work in an enclosure. Every op-amp was tested after soldering during the build up -- entire stages were swapped in and out during this build to learn what worked best.

Above — Device in test mode. The Q original mixer was blown and got replaced with 1 I bought in ~1994 from the late Doug DeMaw. I boosted the side band suppression with the balance trimmer after originally getting 53.4 dB and writing in on the Cu+ clad board.

Above —Boxed up.

Above — Testing the module with some other Jovian modules. More to come.

Click for Module 6, the AF amplifierClick to return to the main index for this project

Tuesday, 18 October 2016

It's fun and useful to listen to your signals from Jovian reception adventures. Along with electromagnetic Jovian bursts, you may also hear bad noise like RFI or hum which might not glare apparent on your monitor.

I designed and built an AF preamp with a buffer, splitter, active AF gain control, a tone control and finally, a discrete component PA for speaker drive. The enclosure also contains 4 extra DC ports to provide DC voltage to other modules via short, shielded RCA cables.
Josh M pointed out that I really have to act carefully to avoid mixing up my AC and DC ports!

Above — Input to tone control output. This PA goes in a metal box since the last thing we want is RF from the AM BCB band getting detected and amplified. The input 39R and 0.22 µF capacitor low-pass filters all signals above 18.6 KHz including offensive RF.

To provide capability to drive a sound card with an isolation transformer, I split the input with 1 path going to the AF preamp and the other to an output RCA port.

1/2 of a TLO82 op-amp generates the virtual ground and it's ground referenced to a single point along with each of the 2 AF boards. Yay! No hum was detected in this module.

The active volume control gives very low noise at lower volume setting levels. R1 and R2 set the maximal gain and can be manipulated with the standard formula: Vout /R2 = -Vin/R1.

Speakers, cabinets and rooms color sound. I feel a tone control is necessary to allow pleasing sound in whatever speaker you ply in your listening room. This is essentially a 1 control Baxandall circuit developed by Douglas Self and modified for single supply.

Above — PA stage with its own virtual ground to use up the other half of the 5532 op-amp. Giving the PA board its own virtual ground made bench testing this board easy.

I've written about this circuit before and developed it for Regen #5. Click and scroll to read about it.

In order to get a power amp output signal swinging as close to the DC rails as possible ( between 0 and 12.3 VDC ), you must depart from a complimentary pair of emitter followers. In these, the collector-to-emitter saturation voltage plus VBE keep you from getting close to the rails. Thus, I've adopted the topology shown -- a common-emitter configuration for the output pair.

Above — a 12-bit FFT of my PA into a 8 Ω resistor. Drive is here set to give 877 mW output power and each vertical division = 10 dB. 877 mW sounds loud and all harmonics lie under 60 dB down.

Click for Module 7, the logarithmic amplifier Click to return to the main index for this project

This year, I built and designed 3 local oscillators, however, this VXO leads in simplicity. Click for my synthesizer for Jupiter receivers.

Above — Entire schematic for the VXO.

2 standard value crystals in "super VXO" format provide a delta F of 135 KHz; just enough to steer around any interference on or around 20.1 MHz. Stability tests showed average frequency variation at room temp is 1/10 of 1 hertz. No need to ovenize crystal oscillators for Jupiter reception.

You may wish to increase the delta F by increasing the inductor size and/or employing a variable capacitor with a wider delta C. It's important to test the output in a 'scope and counter to ensure reasonable frequency stability and that it remains oscillating across the swing of the variable capacitor.

For the main oscillator, I employed the medium power BC337 transistor -- a device with low flicker noise. Mine are original Fairchild parts [ now ON Semiconductor ] as terrible bootleg copies are widely sold now.

Vojtěch Janásek keeps an interesting thread on low noise AF transistors including the BC337. Click for his pdf page link.

Following the Colpitts oscillator, an emitter follower's output port provides a good input match for a common base RF amp. The CB amp provides strong reverse isolation between the oscillator and its load. Again, another emitter follower transforms the high output Z of the common base amp to a low Z which gets low pass filtered and buffered with a 4 dB pad.

Above — Output of the VXO in a DSO.

Above — Simulation of the low-pass output filter in GPLA from EMRFD.

This VXO looks pretty unremarkable -- but adopts careful RF bypass of DC power lines in every transistor amp and runs just enough current to minimize distortion as the signal moves through each RF amp.

The enclosure looks ugly -- it's a re-purposed box from some old project. I threaded in a bolt to block the hole seen on the right side in the reverse view photograph.

Monday, 17 October 2016

Above — Build of the input and RF preamplifier stage. I put in a 0 or 6 dB attenuator right at the input. That's double-sided Cu+ board with copper vias joining the 2 surfaces intermittently and also at crucial RF ground points.

Standard band-pass filters filter steeper and deeper in the high-pass skirt, so I added the input low-pass filter to bolster the low-pass transfer function. It worked.

Above — After breadboarding the preamp, I put a connector on the input and output and measured some parameters.

I ran a RFMD [ now Quorvo ] SiGe HBT MMIC amplifier; a 5 GHz part with a decent OIP3 and that well behaves with respect to parasitic oscillations. Still, then, with such a high Ft, it's suicide unless you adopt a good layout, bypass into UHF, and run 2-sided copper clad board with via wires near the RF ground terminals etc.

With the 1 nF series input and output caps, I measured gain from 1.459 to 50 MHz in this sweep. The 1 nF caps decrease the gain at 20.1 MHz and also at the top end of the broadcast AM band. The gain at 1459 KHz = only 7.59 dB. Some of these high Ft MMICs exhibit massive gain down low and this can trash IMD from AM BCB interference. With the 1 nF coupling caps and the front-end filters, I don't have to worry about the 10 KW AM transmitter located only a few Km from here.

Above — Sweep of the MMIC from 64 - 640 MHz

Above — Sweep of the MMIC from 0.65 - 3 GHz. I could find no unwanted parasitics with bench testing. Satisfied, I hooked up the preamp to the filter and enclosure mounted it.

Above — Sweep of the whole RF filter amplifier from port to port after re-tuning. Happily, the Hammond enclosure didn't add any cavity effects to trigger unwanted oscillations.
I added the 3 dB output pad to further throw away some gain and ensure a wide band 50 Ω impedance output into the mixer of the main receiver. Final S21 = 9.56 dB. The output 3 dB pad = 294.1 Ω 1% tolerance size 0805 resistors from a reel sent to me free by a kind reader + an 18 Ω 5% tolerance resistor I hand picked after measuring 17.7 Ω. The input switch attenuates the signal by 6.2 dB measured.

My goal was 10 dB gain with a stable, wide band match on the input and output. I did not go with the common gate JFET as it's difficult to get a decent output return loss ( > 20 dB ) across a wide bandwidth into 50 Ω; plus I wanted a better OIP3. Still, too, the JFET proves a good low current choice for a Jupiter receiver preamp and its output Z is well suited for easily matching a NE612 Gilbert cell mixer input Z. I also purposely avoided a choke on the MMIC to keep the gain under ~10 dB.

Above — Return loss sweeps of the output port of the RF preamp/filter. Outstanding S22 over a wide band.

Above — Another photo of the entire stage. Built with Ugly Construction, plus hand carved pads of the correct width [ 50 Ω impedance] for the MMIC.