Guide to Audio Electronics--Test instruments and measurements

Audio equipment is, by definition, ultimately intended to provide sounds
which will be heard by a listener. In spite of rather more than a century
of experience in the electrical transmission and reception of audible signals,
the relationships between the electrical waveforms into which sound patterns
are transformed and the sound actually heard by the listener are still not
fully understood.

This situation is complicated by the observable fact that there is a considerable
variation from person to person in sensitivity to, and preferences in respect
of, sound characteristics -- particularly where these relate to modifications
or distortions of the sound. Also, since the need to describe or indeed the
possibility of producing such modified or distorted sounds is a relatively
new situation, we have not yet evolved a suitable and agreed vocabulary by
which we can define our sensations.

There is, however, some agreement, in general, about the types of defect
in electrical signals which, in the interests of good sound quality, the design
engineer should seek to avoid or minimize. Of these, the major ones are those
associated with waveform distortion, under either steady-state or transient
conditions: or with the intrusion of unwanted signals: or with relative time
delays in certain parts of the received signal in relation to others: or with
changes in the pitch of the signal; known as 'wow' or 'flutter', depending
on its frequency. However, the last kind of defect is only likely to occur
in electro- mechanical equipment, such as turntables or tape drive mechanisms,
used in the recording or replay of signals.

It’s sometimes claimed that, at least so far as the design of purely electronic
equipment is concerned, the performance can be calculated sufficiently precisely
that it’s unnecessary to make measurements on a completed design for any other
reason than simply to confirm that the target specification is met. Similarly,
it’s argued that it’s absurd to attempt to endorse or reject any standard
of performance by carrying out listening trials. This is so since, even if
the results of calculations were not adequate to define the performance, instrumental
measurements are so much more sensitive and reproducible than any purely 'subjective'
assessments that no significant error could escape instrumental detection.

Unfortunately, all these assertions remain a matter of some dispute.

With regard to the first of these points - the need for instrumental measurements
- the behavior patterns of many of the components, both 'passive' and 'active',
used in electronic circuit design are complex, particularly under transient
conditions, and it may be difficult to calculate precisely what the final
performance of any piece of audio equipment will be, over a comprehensive
range of temperatures or of signal and load conditions. However, appropriate
instrumental measurements can usually allow a rapid exploration of the system
behavior over the whole range of interest.

On the second point, the usefulness of subjective testing, the problem is
to define just how important any particular measurable defect in the signal
process is likely to prove in the ear of any given listener. So where there
is any doubt, recourse must be had to carefully staged and statistically valid
comparative listening trials to try to determine some degree of consensus.

These trials are expensive to stage, difficult to set up and hard to purge
of any inadvertent bias in the way they are carried out. They are therefore
seldom done, and even when they are, the results are disputed by those whose
beliefs are not upheld.

INSTRUMENT TYPES

An enormous range of instruments is available for use in the test laboratory,
among which, in real-life conditions, the actual choice of equipment is mainly
limited by considerations of cost, and of value for money in respect to the
usefulness of the information which it can provide.

Although there is a wide choice of test equipment, much of the necessary
data about the performance of audio gear can be obtained from a relatively
restricted range of instruments, such as an accurately calibrated signal generator,
with sinewave and square-wave outputs, a high input impedance, wide bandwidth
AC voltmeter, and some instrument for measuring waveform distortion - all
of which would be used in conjunction with a high-speed double trace cathode-ray
oscilloscope. I have tried, in the following pages, to show how these instruments
are used in audio testing, how the results are interpreted, and how they are
made. Since some of the circuits which can be used are fairly simple, I have
given details of the layouts needed so that they could be built if required
by the interested user.

SIGNAL GENERATORS

Sinewave oscillators

Variable frequency sinusoidal input test waveforms are used for determining
the voltage gain, the system bandwidth, the internal phase shift or group
delay, the maximum output signal swing and the amount of waveform distortion
introduced by the system under test. For audio purposes, a frequency range
of 20 Hz to 20 kHz will normally be adequate, though practical instruments
will usually cover a somewhat wider bandwidth than this. Except for harmonic
distortion measurements, a high degree of waveform purity is probably unnecessary,
and stability of output as a function of time and frequency is probably the
most important characteristic for such equipment.

It’s desirable to be able to measure the output signal swing and voltage
gain of the equipment under specified load conditions. In , for example, an
audio power amplifier, this would be done to determine the input drive requirements
and output power which can be delivered by the amplifier.

For precise measurements, a properly specified load system, a known frequency
source and an accurately calibrated, RMS reading, AC voltmeter would be necessary,
together with an oscilloscope to monitor the output waveform to ensure that
the output waveform is not distorted by overloading.

Some knowledge of the phase errors (the relative time delay introduced at
any one frequency in relation to another) can be essential for certain uses
- For example, in long-distance cable transmission systems - but in normal
audio usage such relative phase errors are not noticeable unless they are
very large. This is because the ear is generally able to accept without difficulty
the relative delays in the arrival times of sound pressure waves due to differing
path lengths caused by reflections in the route from the speaker to the ear.

Oscillators designed for use with audio equipment will typically cover the
frequency range 10 Hz to 100 kHz, with a maximum output voltage of, perhaps,
10 V RMS. For general purpose use, harmonic distortion levels in the range
0.5-0.05% will probably be adequate, though equipment intended for performance
assessments on high quality audio amplifiers will usually demand waveform
purity (harmonic distortion) levels at 1 kHz in the range from 0.02% down
to 0.005%, or lower. In practice, with simpler instruments, the distortion
levels will deteriorate somewhat at the high and low frequency ends of the
output frequency band.

A variety of electronic circuit layouts have been proposed for use as sinewave
signal generators, of which by far the most popular is the ' Wien Bridge'
circuit shown in FIG. 1. It’s a requirement for continuous oscillation in
any system that the feedback from output to input shall have zero (or some
multiple of 360°) phase shift at a frequency where the feedback loop gain
is very slightly greater than unity, though to avoid waveform distortion,
it’s necessary that the gain should fall to unity at some value of output
voltage within its linear voltage range.

In the Wien bridge, if R1 = R2 = R and Q = C2 = C, the condition for zero
phase shift in the network is met when the output frequency,

fo = 1/(2 pi RC).

At this frequency the attenuation of the RC network, from
Y to X, in FIG. 1, is 1/3. The circuit shown will therefore oscillate at
f0 if the gain of the amplifier, A1, is initially slightly greater than three
times.

The required gain level can be obtained automatically by the use of a thermistor
(???) in the negative feedback path, and the correct choice of the value of
R3.

Since QR1 and C2R2 are the frequency-determining elements, the output frequency
of the oscillator can be made variable by using a twin-gang variable resistor
as Ri/R2 or a twin gang capacitor as Q/C2. If a modern, very low distortion,
operational amplifier, such as the LM833, the NE5534 or the OP27 is used as
the amplifier gain block (??) in this circuit, the principle source of distortion
will be that caused by the action of the thermistor (???) used to stabilize
the output signal voltage, where, at lower frequencies, the waveform peaks
will tend to be flattened by its gain-reduction action. With an RS Components
'RA53' type thermistor as TH1, the output voltage will be held at approximately
1 V RMS, and the THD at 1 kHz will be typically of the order of 0.008%. The
output of almost any sinewave oscillator can be converted into square-wave
form by the addition of an amplifier which is driven into clipping. This could
be either an opamp, or a string of CMOS inverters, or, preferably, a fast
voltage comparator IC, such as that also shown in FIG. 1, where RVX is used
to set an equal mark to space ratio in the output waveform. An alternative
approach used in some commercial instruments is simply to use a high-speed
analogue switch, operated by a control signal derived from a frequency stable
oscillator, to feed one or other of a pair of preset voltages, alternately,
to a suitably fast output buffer stage.

An improved Wien bridge oscillator circuit layout of my own, shown in FIG.
2 (Wireless World, May 1981, pp. 51-53) in which the gain blocks Ax and A2
are connected as inverting amplifiers, thereby avoiding 'common mode' distortion,
is capable of a THD below 0.003% at 1 kHz with the thermistor controlled amplitude
stabilization layout shown in FIG. 2, and about 0.001% when using the improved
stabilization layout, using an LED and a photo-conductive cell, described
in the article.

As a general rule the time required (and, since this relates to a number
of waveform cycles, it will be frequency dependent) for an amplitude stabilized
oscillator of this kind to 'settle' to a constant output voltage, following
some disturbance (such as switching on, or alteration to its output frequency
setting), will increase as the harmonic distortion level of the circuit is
reduced. This characteristic is a nuisance for general purpose use where the
THD level is relatively unimportant. In this case an alternative output voltage
stabilizing circuit, such as the simple back-to back connected silicon diode
peak limiter circuit shown in FIG. 3, would be preferable, in spite of its
relatively modest (0.5% at 1 kHz) performance in respect of waveform distortion.

FIG. 2 Improved Wien bridge oscillator.

FIG. 3 Diode-stabilized oscillator.

FIG. 4 Amplitude control by FET.

FIG. 5 Output level control.

Rather greater control of the output signal amplitude can be obtained by
more elaborate systems, such as the circuit shown in FIG. 4. In this circuit,
the output sinewave is fed to a high-input impedance rectifier system (A2/D1/D2)9
and the DC voltage generated by this is applied to the gate of an FET used
as a voltage controlled resistor. The values chosen for IVR7 and C3/C4/C5
determine the stabilization time constant and the output signal amplitude
is controlled by the ratio of R8:R9- In operation, the values of R4 and R3
are chosen so that the circuit will oscillate continuously with the FET (Q1)
in zero-bias conducting mode. Then, as the -ve bias on Q1 gate increases,
as a result of the rectifier action of Q1/Q2, the amplitude of oscillation
will decrease until an equilibrium output voltage level is reached.

In commercial instruments, a high quality small-power amplifier would normally
be interposed between the output of the oscillator circuit and the output
take-off point to isolate the oscillator circuit from the load and to increase
the output voltage level to, say, 10 V RMS. An output attenuator of the kind
shown in FIG. 5 would then be added to allow a choice of maximum output voltage
over the range 1 mV to 10 V RMS, at a 600-ohm output impedance.

A somewhat improved performance in respect of THD is given by the 'parallel
T' oscillator arrangement shown in FIG. 6, and a widely used and well-respected
low-distortion oscillator was based upon this type of frequency-determining
arrangement. This differs in its method of operation from the typical Wien
bridge system in that the network gives zero transmission from input to output
at a frequency determined by the values of the resistors Rx, Ry and Rz, and
the capacitors Cx, Cy and Cz. If Cx = Cy = Cz/2 = C, and Rx = Ry = 2RZ = R,
the frequency of oscillation will be 1/2 pi CR, as in the Wien bridge oscillator.

If the parallel T network is connected in the negative feedback path of a
high gain amplifier (??) oscillation will occur because there is an abrupt
shift in the phase of the signal passing through the T' network at frequencies
close to the null, and this, and the inevitable phase shift in the amplifier
(Ax) converts the nominally negative feedback signal derived from the output
of the T' network into a positive feedback, oscillation sustaining one.

FIG. 6 Oscillator using parallel T.

A problem inherent in the parallel T design is that in order to alter the
operating frequency it’s necessary to make simultaneous adjustments to either
three separate capacitors or three separate resistors. If fixed capacitor
values are used, then one of these simultaneously variable resistors is required
to have half the value of the other two. Alternatively, if fixed value resistors
are used, then one of the three variable capacitors must have a value which
is, over its whole adjustment range, twice that of the other two. This could
be done by connecting two of the 'gangs' in a four-gang capacitor in parallel,
though , for normally available values of capacitance for each gang, the resistance
values needed for the 'T' network will be in the megohm range. Also, it’s
necessary that the drive shafts of Cx and Cy shall be isolated from that of
Cz.

These difficulties are lessened if the oscillator is only required to operate
at a range of fixed 'spot' frequencies, and a further circuit of my own of
this kind {Wireless World, July 1979, pp. 64-66) is shown in simplified form
in FIG. 7. The output voltage stabilization used in this circuit is based
on a thermistor/resistor bridge connected across a transistor, Q1. The phase
of the feedback signal derived from this, and fed to Al5 changes from +ve
to -ve as the output voltage exceeds some predetermined output voltage level.
The THD given by this oscillator approaches 0.0001% at 1 kHz, worsening to
about 0.0003% at the extremes of its 100 Hz to 10 kHz operating frequency
range.

FIG. 7 Level stabilizer circuit.

It’s expected in modern wide-range low-distortion test bench oscillators
that they will offer a high degree of both frequency and amplitude stability.

This is difficult to obtain using designs based on resistor/capacitor or
inductor/capacitor frequency control systems, and this has encouraged the
development of designs based on digital waveform synthesis, and other forms
of digital signal processing.

Digital waveform generation

Because of the need in a test oscillator for a precise, stable and reproducible
output signal frequency a number of circuit arrangements have been designed
in which use is made of the frequency drift-free output obtainable from a
quartz crystal oscillator. Since this will normally provide only a single
spot-frequency output, some arrangement is needed to derive a variable frequency
signal from this fixed frequency reference source.

One common technique makes use of the 'phase locked loop' (PLL) layout shown
in FIG. In this, the outputs from a highly stable quartz crystal 'clock' oscillator
and from a variable-frequency 'voltage controlled oscillator' (VCO) are taken
to a 'phase sensitive detector' (PSD) - a device whose output consists of
the 'sum' and 'difference' frequencies of the two input signals. If the sum
frequency is removed by filtration, and if the two input signals should happen
to be at the same frequency, the difference frequency will be zero, and the
PSD output voltage will be a DC potential whose sign is determined by the
relative phase angle between the two input signals.

If this output voltage is amplified (having been filtered to remove the unwanted
'sum' frequencies), and then fed as a DC control voltage to the VCO (a device
whose output frequency is determined by the voltage applied to it), then,
providing that the initial operating frequencies of the clock and the VCO
are within the frequency 'capture' range determined by the loop low-pass filter,
the action of the circuit will be to force the VCO into frequency synchronism
(but phase quadrature) with the clock signal: a condition usually called 'lock'.
Now if, as in FIG. 8, the clock and VCO signals are passed through frequency
divider stages, having values of -î-M and -HN respectively, when the loop
is in lock the output frequency of the VCO will be F_out = Fck (N/M). If the
clock frequency is sufficiently high, appropriate values of M and N can be
found to allow the generation of virtually any desired VCO frequency. In an
audio band oscillator, since the VCO will probably be a 'varicap' controlled
LC oscillator, operating in the MHz range, the output signal will normally
be obtained from a further variable ratio frequency divider, as shown in FIG.
For the convenience of the user, once the required output frequency is keyed
in, the actual division ratios required to generate the chosen output frequency
will be determined by a microprocessor from ROM-based look-up tables, and
the output signal frequency will be displayed as a numerical read-out.

FIG. 8 Phase-locked loop oscillator.

Given the availability of a stable, controllable frequency input signal,
the generation of a low-distortion sine wave can, again, be done in many ways.

For example, the circuit arrangement shown in FIG. 9 is quoted by Horowitz
and Hill (The Art of Electronics, 2nd edition, page 667). In this
a logic voltage level step is clocked through a parallel output shift register
connected to a group of resistors whose outputs are summed by an amplifier
(Ai). The output is a continuous waveform, of staircase type character, at
a frequency of F_ck/16.

If the values of the resistors R4-R7 are chosen correctly the output will
approximate to a sinewave, the lowest of whose harmonic distortion components
is the 15th, at -24 dB. This distortion can be further reduced by low-pass
filtering the output waveform. A more precise waveform, having smaller, higher
frequency staircase steps, could be obtained by connecting two or more such
shift registers in series, with appropriate values of loading resistors.

Like all digitally synthesised systems, this circuit will have an output
frequency stability which is as good as that of the clock oscillator, which
will be crystal controlled. Also the output frequency can be numerically displayed,
and there will be no amplitude 'bounce' on switch on, or on changing frequency.

A more elegant digitally synthesised sinewave generator is shown in FIG.
10. In this the quantized values of a digitally encoded sine waveform are
drawn from a data source, which could be a numerical algorithm, of the kind
used , for example, in a 'scientific' calculator, but, more conveniently,
would be a ROM-based 'look-up' table. These are then clocked through a shift
register into a 16-bit digital-to-analogue converter (DAC). If the data chosen
are those for a 16-bit encoded sinusoidal waveform, the typical intrinsic
purity of the output signal will be of the order of 0.0007%, improved by the
use of low-pass, sample-and-hold filtering.

Moreover, if digital filtering is used, prior to the DAC, this can be made
to track the frequency of the output signal. As in the previous design ( FIG.
9) the output signal frequency is related to, and controlled by, the clock
frequency.

FIG. 9 Digital waveform generation.

ALTERNATIVE WAVEFORM TYPES

FIG. 10 ROM-based waveform synthesis. DAC --- Output waveform

FIG. 11 ICL
8038 application circuit.

A range of waveforms including square- and rectangular-wave shapes as well
as triangular and 'ramp' type outputs are typically provided by a 'function
generator'. The outputs from this kind of instrument will usually also include
a sinusoidal waveform output having a wide frequency range but only a modest
degree of linearity.

An IC which allows the provision of all these output waveform types is the
ICL '8038' and its homologues , for which the recommended circuit layout is
shown in FIG. 11. The output from this is free from amplitude 'bounce' on
frequency switching or adjustment, and can be set to give a 1 kHz distortion
figure of about 0.5% by adjustment of the twin-gang potentiometers RV4 and
RV5. As shown, the frequency coverage, by a single control (RV2), is from
20 Hz to 20 kHz.

Since a square-wave signal contains a very wide range of odd-order harmonics
of its fundamental frequency, a good quality signal of this kind, with fast
leading edge (rise) and trailing edge (fall) times, and negligible overshoot
or 'ripple', allows the audio systems engineer to make a rapid assessment
both of the load stability of an amplifier and of the frequency response of
a complete audio system.

With an input such as that shown in FIG. 12(a), an output waveform of the
type in 7.12(b) would indicate a relatively poor overall stability, by comparison
with that shown in 12(c), which would merely indicate some loss of high frequency.
The type of waveform shown in 12(d) would imply that the system was only conditionally
stable and unlikely to be satisfactory in use.

FIG. 12 (a) Square-wave input waveform (wide bandwidth).

FIG. 12 (b) poor amplifier stability.

FIG. 12 (c) relatively rapid (- 12 dB/octave) loss of HF gain.

FIG. 12 (d) output showing conditional stability.

FIG. 12 (e) poor LF gain.

FIG. 12 (d) output showing conditional stability.

FIG. 12 (e) poor LF gain.

FIG. 12 (f) -(h) loss of HF gain.

The type of oscilloscope waveform shown in FIG. 12(e) would indicate a fall-off
in gain at low frequencies, whereas that of FIG. 12(f) would show an increase
in gain at LF. Similarly, the response shown in FIG. 12(g) would be due to
an excessive HF gain, while that of FIG. 12(h) would indicate a fall-off in
HF gain. With experience, the engineer is likely to recognize the types of
output waveform which are associated with many of the common gain/frequency
characteristics or design or performance problems.

The other types of waveform which can be provided by a function generator
have specific applications which will be examined later. The sinewave output,
though too poor in linearity to allow amplifier THD measurements to be made
with any accuracy, is usually quite free from amplitude variation, and the
'single knob' wide-range frequency control of such an instrument allows fast
checking of the performance of such circuits as low- or high-pass filters
or tone controls.

FIG. 13 Origins of IM distortion.

DISTORTION MEASUREMENT

One of the more important characteristics of any audio system is the extent
to which the signal waveform is distorted during its passage through the system.
Where the input signal (f_in) is of a single frequency, and this distortion
is due to some non-linearity in the transfer characteristics of the signal
handling stages, this non-linearity will generate a series of further spurious
signals occurring at frequencies which are multiples of the input frequency
(or frequencies). Because these spurious signals have a harmonic relationship
(i.e. 2/in, 3/in, 4/in, 5/in, etc.) to that of the input waveform this characteristic
is known as harmonic distortion and, since all such harmonics will be present
in the residue, it’s usually referred to as 'total harmonic distortion' (THD).
Where the input waveform consists of signals at two or more distinct frequencies,
a further type of distortion will occur, in which the amplitudes of each of
the signals will be modulated to some extent at the frequency of the others
- an effect which is more easily seen if there are only two input signals
and these are widely different in both amplitude and frequency, as shown in
FIG. 13. This type of distortion is known as Intermodulation Distortion' (IMD)
and leads to a loss of clarity in the reproduction of such complex signals.
Since, in audio applications, the input signal will be typically of multiple
frequency form, IMD is very important as a design quality. There is, unfortunately,
no direct numerical relationship between IMD and THD, except that the lower
the THD of a system, the lower the IMD will also be.

FIG. 14 Parallel T notch filter characteristics.

THD meters

Two quite distinct techniques are used for such distortion measurement: Total
Harmonic Distortion' (THD) meters and 'Spectrum Analyzers'. In the THD meter,
the instrument is usually arranged so that there will be a sharp notch at
some point in its gain/frequency curve, which will give zero transmission
at some specific frequency, as shown in FIG. 14. If this notch is tuned to
the same frequency as a low-distortion sinewave input test signal, then, when
the test signal is 'notched out', what is left will be the spurious harmonics
(plus hum and noise) introduced by the equipment under test, or the test instrument
or the associated wiring connections.

In the normal method of use of this instrument, the 100% setting is found
if the notch filter circuit is temporarily by-passed, its output is taken
to a sensitive wide-bandwidth AC milli-voltmeter, and the system gain is adjusted
so that the meter reads full scale. Then, when the notch is once more switched
into circuit, the meter reading will show the residual THD + noise and hum.

In those cases where the intrinsic hum and noise in the system is fairly
low, and the input test signal is a very low distortion sinewave - with, ideally,
less than 0.01% THD - a simple THD measurement will often be entirely adequate
to determine the performance of the equipment, particularly if the residual
output signal waveform, following the notch circuit, is displayed on an oscilloscope.

FIG. 15 Crossover distortion residues.

FIG. 16 Distortion residues in high-quality audio amplifier

FIG. 17 Distortion from high-quality phonograph pickup.

With experience, the engineer can derive a lot of useful information about
the circuit performance from this type of display. For example, in FIG. 15,
the characteristic 'spikes' due to 'crossover distortion' in the output waveform
of a relatively poorly designed (or badly adjusted) push-pull transistor amplifier
can be seen very clearly in the THD meter output (and would, in practice,
be audibly unpleasant), even though the waveform distortion which caused the
spikes cannot be seen very easily in the output waveform shown above.

In FIG. 16 the THD meter output waveform shown is that from a high-quality
high-power transistor amplifier at an output voltage swing, into a water-cooled
resistive load, which is just below clipping level.

Because of some slight asymmetry in the operation of the amplifier under
test - most probably in the output stage - there is a slight flattening of
the lower part of the signal waveform, and this causes the slight undulation
in the residual THD display which occurs at these negative-going peaks.

There are also some small crossover-type waveform discontinuities just visible
in the THD trace, coincident with the mid-point of the voltage swing. The
peak-to-peak amplitude of these is about the same level as that of the random
background noise of the amplifier and meter. Because of their brief duration,
they represent a very high order harmonic content -- probably the thirteenth
or higher, and would, possibly be supersonic. Also, because of their brief
duration, the energy associated with these glitches is very small. As a matter
of interest, the actual harmonic distortion which was indicated for the amplifier
test shown in FIG. 16 was 0.006%. A much more obvious distortion residue is
that shown in FIG. 17. Since the THD residue waveform has twice the number
of peaks as the input waveform it’s therefore due to the second harmonic.
It was, in fact, measured as 0.65%, and was the output from a (high quality)
phonograph pick-up replaying a 1 kHz test groove, cut at a lateral velocity
of 5 cm/s on an vinyl LP test disc. The noise on the trace is surface noise
from the disc.

A test signal of this kind is invaluable for setting the pick-up tracing
angle, and, as shown, was almost certainly that when the cartridge was optimally
adjusted.

As can be seen from the illustrations, a useful feature of this kind of measurement
is that it allows the fault in the system to be related to that part of the
signal waveform where it occurs. A disadvantage, also shown, is that the presence
of high frequency circuit noise, as in FIG. 16, tends to conceal the error
signals, while that of hum and low frequency noise as shown in FIG. 17 (in
this case due to surface noise and 'rumble') blurs the display. For these
reasons, most typical THD meters include optional low-pass and high-pass filters
to lessen unwanted output artifacts. These filters are unsuitable for use
at the low- and high-frequency ends of the audio spectrum, and lessen the
usefulness of THD measurements under these conditions.

Almost any notch circuit can be used in an instrument of this kind, of which
the two most popular are the Wien network nulling circuit, shown in outline
form in FIG. 18, and the 'parallel T' notch circuit shown in Fig. 19. A
practical circuit of my own (Wireless World, July 1972, pp. 306-308), based
on a Wien bridge null network, is shown in FIG. 20. In order to get an accurate
result the notch must be sharp enough to remove the fundamental frequency
entirely, while avoiding any attenuation of the second harmonic frequency.
In the circuit shown, the notch is sharpened up by the use of some overall
negative feedback between the emitters of Q4 and Q1 via R13 and R6.

FIG. 18

FIG. 19

FIG. 20

IMD meters

As noted above, if the transfer characteristics of the system are non-linear,
and there are two (or more) signals present at the same time, a further type
of distortion will occur, as shown in FIG. 13. This is termed 'Intermodulation
Distortion' (IMD), and arises because , for any signal, the gain of the system
is defined by the slope of the transfer characteristic. This means that, for
example, if a large signal is present at the same time as a smaller one, it
will have the effect of moving the smaller signal up and down the transfer
curve, and its gain, and consequently the size of the output voltage due to
it, will change at the frequency of the larger one.

A variety of techniques are used to measure IMD of which the earliest (1961)
was that proposed by the ( US) Society of Motion Picture and Television Engineers
(SMPTE). In this two signals, typically 70 Hz and 7 kHz, at a 10:1 magnitude
ratio, are fed to the system, the 7 kHz signal is separated by filtering and
then demodulated to recover the 70 Hz intermodulation component which is measured
using an RMS reading meter.

An alternative technique which allows the measurement of system performance
in the 11-20 kHz region , for which harmonic distortion measurements would
not be easily applicable, is the CCIR 'difference frequency' method in which
two tones, of equal amplitude, at , for example, 19 kHz and 20 kHz are used
as test signals, and the 1 kHz difference tone due to IM effects is isolated
by filtering and measured. This type of test can be performed with various
input signal frequencies, or even with the input signals swept over the 4-20
kHz range at a constant difference frequency (1 kHz or 70 Hz). A more recent
approach, called the Total Difference Frequency' method, proposes test frequencies
of 8 kHz and 11.95 kHz. The simple difference frequency product from these
will be 3.95 kHz, while the second harmonic of the 8 kHz test tone will generate
a 4.05 kHz difference frequency signal. Non-linearities at the HF end of the
audio band are shown up by interactions between the third harmonic of the
8 kHz tone and the second harmonic of the 11.95 kHz one, which would produce
a 100 Hz difference tone.

These high-frequency tests allow performance evaluations under dynamic conditions
as distinct from the more usual single-frequency 'steady-state' tests. This
is important, not only because components behave differently under rapid rates
of change of operating conditions, and can malfunction in ways not found in
steady state, but also because the bulk of the program material on which audio
equipment operates is of a complex electrical form, rich in transients, and
the skilled listener can readily detect alien sounds, within this total, due
to system errors which may not be revealed by simple steady-state tests.

Spectrum analysis

The most powerful - but also the most complex, the most expensive and the
most difficult to interpret - technique for determining system linearity is
that known as 'spectrum analysis'. In the audio field, this is done by feeding
the system with one or more input test tones, and then measuring the amplitude
of all output signal components as a function of their frequency. This will
immediately reveal both harmonics of the signal and their intermodulation
products, usually shown graphically as amplitude on a logarithmic scale (dBV)
as a function of frequency. A good instrument of this kind will allow the
display of signal components as low as -120 dB, mainly limited by the noise
floor.

Early instruments of this type used a layout, shown in simplified form in
FIG. 21, which is somewhat similar to that of a radio receiver. In this, the
input signal from the system under test, after amplification and buffering
by a very low distortion input amplifier, is caused to modulate the amplitude
of a swept frequency sine wave signal, typically in the 100 kHz range, which
is then amplified by a very narrow pass-band HF amplifier.

The output from this amplifier is demodulated and converted into a DC voltage
which is fed to a paper chart recorder to give the type of display sketched
in FIG. 22. The frequency sweep system is coupled both to the chart recorder
paper drive and to the variable frequency oscillator so that there will be
a precise relationship between the frequency being sampled and the paper chart
position. A fundamental problem with this type of instrument is that the response
speed of the IF amplifier/demodulator system is directly related to the bandwidth
chosen for the IF amp., so, if a high degree of spectral resolution is required
(i.e. narrow IF pass-band) the sweep frequency rate and the chart drive speed
(which are coupled together) must both be slow, perhaps requiring a minute
or more for a complete 20 Hz to 20 kHz sweep.

Modern instruments for use in the audio band employ digital signal processing
techniques based on 16 to 20-bit quantization resolution and sampling rates
up to 204.8 kHz, depending on the application. The components of the signal
under analysis are then resolved by fast Fourier transform (FFT) techniques.
Typical contemporary instruments of this type are the Hewlett Packard 3562A,
35665A and 35670A dynamic signal analyzers. Instruments of this type are somewhat
more rapid in response than the earlier swept frequency systems, with a typical
high-resolution frequency spectrum, depending on the input signal frequency
span, requiring less than a minute to display or refresh. A range of outputs
would normally be available, including interface data for use with a computer,
but if the output required is in the form of a graphical display, this would
be provided by a laser or bubble jet printer. A typical numerical analysis
of such an encoded FFT signal would allow, as , for example, in the Bruel
and Kjaer 2012 instrument, the simultaneous graphical display of individual
harmonic distortion components as a function of frequency - a facility which
is very useful for loudspeaker, headphone or microphone design.

FIG. 21 Spectrum analyzer layout.

FIG. 22

FIG. 23 Typical oscilloscope display.

OSCILLOSCOPES

The remaining major class of test instrument in the audio engineer's laboratory
is the 'cathode ray oscilloscope' - normally just called an 'oscilloscope'
or 'scope'. This is an instrument which allows a voltage waveform to be displayed,
visually, as a graph of voltage as a function of time. Normally, the horizontal
axis of the display - the .Y-axis - will represent elapsed time, and the vertical
axis - the Y-axis - will represent the signal voltage being displayed. The
way by which this is done is shown, in a simplified manner, in Figs 23 and
24.

The oscilloscope tube has an electron source at one end, typically an indirectly
heated thermionic 'cathode', from which a stream of electrons is drawn and
accelerated towards a layer of fluorescent material, the 'screen', deposited
on the inner face of the glass envelope of the tube. The internal electrodes
in the tube cause the electron beam to be focused so that its region of impact
on the screen generates a small, bright, point of light. If the target material
is zinc sulphide, which is an efficient phosphor material, the emitted light
will be of a green color. Other colored screens are available, but green is
preferred by most engineers as a restful color for extended viewing. A group
of four deflection plates is interposed between the electron source and the
screen as shown in FIG. 23, and the lateral or vertical position of impact
of the electron beam on the screen will be controlled by the voltages present
on these plates at the instant when the individual electrons forming the stream
pass between them.

FIG. 24 Timebase scan waveform.

In a simple 'scope, the horizontal scan voltage will be a 'sawtooth' waveform
of the kind shown in FIG. 24. In principle, this could be generated by a 'function
generator' IC, but in practice, since it’s essential that the voltage waveform
is extremely linear between points A and B, and occupy as little time as practicable
in returning from point B to point C to begin the cycle again, 'scope manufacturers
will normally provide their own discrete component circuitry for this purpose,
though there will usually be an external electrical input point to allow the
user to supply his or her own horizontal scan waveform, when needed. (Since
it’s desirable to avoid confusion of the display due to the presence of extra
images during the scan return from B to C, the trace is usually blanked out
during this period.) The voltage waveform which it’s desired to display is
then applied, after suitable amplification, to the Y plates, which causes
the electron beam to be deflected vertically, to an extent which is accurately
related to the applied voltage.

The vertical or horizontal position of the display on the tube screen is
controlled by the application of a suitable DC voltage to the plates, normally
called the 'A'-shift' or ?-shift' controls. To preserve the sharpness of the
spot focused on the screen the electronic circuitry of the X and Y amplifiers
is chosen so that the voltages applied to the deflection plates are in 'push-pull'
(i.e. equal in magnitude but opposite in polarity). To allow some fine trimming
of the focus and deflection voltages , for optimum spot sharpness, an 'astigmatism'
control will usually be provided.

The length of time which is occupied in traversing from A to B will determine
the time axis of the display, so that , for example, if the waveform displayed
on FIG. 23 as a result of an alternating voltage applied to the Y plates is
a 1 kHz sine wave (i.e. 1 ms/cycle), then the horizontal (X-axis) time scale
will be 5 ms, which would imply a 200 Hz repetition rate for the X waveform.
The ability of the 'scope to display very high-frequency input waveforms is
limited only by the ability of the 'scope electronics to charge and discharge
the stray capacitances associated with the plate electrodes, through the inevitable
inductances due to the plate connecting leads.

Generally, the 'scope time base generator circuit will allow the user to
choose the X-axis repetition frequency, as a switchable sequence of scan times,
ranging, perhaps, from 10 s/cm to 0.1 µ^?t?. There will also be a control
which will allow some variation in these pre-set speed settings to facilitate
synchronization of the signal being displayed with the horizontal scan speed.
On some 'scopes, there will also be an option which will enable the user to
increase the scan width by a factor of 10. Since the repetition rate remains
the same, the effect of this will be to leave the waveform synchronized with
the horizontal scan, but to spread this out so that the time component of
the waveform can be seen in greater detail, since the X-shift control will
allow the waveform displayed to be moved horizontally. The facility of expanding
a small portion of the X trace, to speed up part of the display, while leaving
the remainder unaffected, is still sometimes offered, as has been the ability
to display different traces at different scan speeds, but these are unusual
facilities.

An interesting curiosity in this field is that the spot of light on a 'scope
screen is able to move at a velocity which is greater than the speed of light
-- because it’s not itself a physical entity, merely a historical record.

The X and Y amplifiers, which amplify the time base and y input signals before
presentation to the deflection plates, are generally similar in sensitivity
and bandwidth, to facilitate the use of either of these inputs for signal
purposes. A normal maximum y input sensitivity will be in the range 1-10 mV/cm,
with a bandwidth of up to 100 MHz, depending on the cost and performance bracket
of the instrument. For audio use a 2 mV/cm sensitivity and a 10-20 MHz bandwidth
would be quite adequate. The need for such a high HF bandwidth arises because,
particularly with MOSFET devices, circuit instabilities may lead to spurious
oscillation in the 5-20 MHz range, and it’s desirable that the 'scope should
be able to show this, when it’s present.

It’s also normally expected that the Y amplifier should be 'direct coupled',
so that a DC voltage can be displayed as a vertical offset of the horizontal
scan position. The 'shift' voltages are added to the y (or X-axis) signals,
after the input attenuation, so that the required scan position can be obtained
under signal input conditions where both AC and DC inputs are present. If
it’s preferred to ignore the DC component of the signal, an input blocking
capacitor can be switched into circuit. The 'direct coupled' input facility
is very useful for measuring small DC offset voltages in sensitive regions
of the circuitry.

Multiple trace 'scope displays

With the exception of small, portable or battery operated instruments, almost
all modern 'scopes offer multiple 'trace' displays, which enable two or more
Y-axis signals to be shown simultaneously, one above the other on the screen.
This facility is exceedingly valuable, in that it allows the comparison of
the input and output waveforms from any system under test, so that changes
in waveform shape, or delays in time between input and output, can be seen
immediately.

When using a simple notch-type THD meter, a dual trace display is also very
helpful in that the input waveform can be shown on, say, the upper trace,
and used for display synchronization, while the noise and distortion residues
are displayed on the other. Not only does this allow the identification of
the position on the waveform at which the defect occurred, but it also prevents
the loss of synchronization which would otherwise take place when the input
waveform was 'notched out'. With systems using digital signals multiple trace
displays allow the timing of a number of output or input pulse streams to
be checked, though this is an area somewhat outside the scope of this book.

In early 'scopes, the provision of multiple trace displays was achieved by
the use of separate electron 'guns', focusing electrodes and X and Y plate
assemblies, but this made for bulky and expensive CR tubes. In modern 'scopes,
the technique employed uses a combination of fast electronic switching applied
to the input signal in synchronism with a superimposed square wave vertical
deflection waveform of adjustable size. This allows one or other of the input
signals to be switched to the Y amplifier, and thence to the Y deflection
plates, at a vertical display position determined by users by that choice
of square wave size. The display sequence may be either 'alternate', in which
sequential '^-axis' sweeps are chosen to display alternate inputs, or 'chopped',
in which the input waveforms are rapidly sampled, during the sweep, and routed,
up or down, to the chosen display positions.

Normally the choice of trace division technique is determined automatically
by the 'scope circuitry according to the A'-axis sweep speed selected. At
low sweep speeds, the input waveforms will be chopped at a frequency which
is high in relation to the signal frequency. This would be preferable , for
example, if it was desired to do time comparisons between waveforms from different
Y inputs. For high-frequency signals, alternate displays are normally selected,
on the grounds that the persistence of vision will make the display seem continuous.
The accuracy of the relative horizontal position of such a display is, in
this case, dependent on the freedom from 'jitter' of the time-base synchronization.

Time-base synchronization:

The ability to 'freeze' a repetitive waveform on the 'scope display is an
essential quality in any oscilloscope, and facilities are invariably provided
to allow trace synchronization with any specified signal input, but also from
an external 'sync' input. Various types of synchronization are also usually
provided by switch selection, in the hope that some of these may be more effective
that others. My choice of words is prompted by the experience, over many years
and with many 'scopes, that some 'scopes and some waveforms can prove very
difficult to lock on the screen. In fact, one of my personal priorities in
the choice of an oscilloscope is ease of synchronization. Sadly, one only
ever discovers whether this is good or poor after the purchase of the instrument
- unless one has had previous experience with that make and model.

External X-axis inputs and Lissajous figures

The input to the X-axis amplifiers can usually be disconnected from the output
of the sawtooth waveform generator and taken from an external 'Time Base Input'
connection. This then allows a single-beam 'scope to be used to display 'Lissajous
figures' of the kind shown in FIG. 25. In the ring-shaped display of FIG.
25(a), the external X input and Y inputs are at the same frequency, though
at phase quadrature. If they are in phase, or in phase opposition, the resultant
display will be a straight line at 45° or 135° to the horizontal axis.

FIG. 25 Typical Lissajous figures.

FIG. 26 Mono/stereo signal display.

In FIG. 25(b), the X input is at twice the frequency of the Y one, while
in FIG. 25(c) the Y input is twice the frequency of the X input. In FIG. 25(d)
the X input is three times the frequency of the Y input -- and so on.

This provides a rough and ready means of assessing the frequency of an input
sinusoidal signal if one has a calibrated sine wave source. However, small
frequency drifts in the signal inputs lead to continuous snake-like inter-twinings
of the display, which makes the precise determination of frequency difficult
to achieve.

With a dual trace oscilloscope, the same result may be obtained more easily
by displaying the signal of unknown frequency, in synchronism on one trace,
and then using the second trace to display the output from a variable, but
calibrated signal generator. Apart from the difficulty of obtaining simultaneous
synchronization, the unknown frequency can be determined from noting when
both traces display the same number of cycles.

An interesting application of the Lissajous principle is shown in FIG. 26,
where the LH and RH channels of a stereo signal, such as , for example, an
FM tuner, are connected to the X and Y inputs of the 'scope.

A mono signal will give the kind of display shown in FIG. 26(a), where the
length of the line display will depend on the magnitude of the input signal.

On a speech or music input, a stereo signal will give the kind of display
shown in FIG. 26(b), where the extent of the 'roundness' of the pattern is
an indication of the extent of the image separation of the two channels.

This allows the adjustment for maximum channel separation of an FM stereo
decoder , for example, without the need to seek special test signals.

'X-axis' signal outputs

Most 'scopes also provide a 'time-base output' socket, at which the 'X-axis'
scan voltage waveform ( FIG. 24) is available. This is required for controlling
the frequency sweep of the voltage controlled RF oscillator in a frequency
modulated oscillator, or 'wobbulator', whose use was described in SECTION
2 (Tuners and Radio Receivers), and for which the output waveforms were shown
in Figs 41-43. A similar system is used for the 'Panoramic' display of radio
signals as a function of frequency.

Digital storage oscilloscopes

Oscilloscopes, like all other electronic test equipment, has become more
versatile, more complicated and more expensive with the passage of time, and
this development is particularly notable in the field of 'storage' oscilloscopes,
which allow the provision of numerically stored displays.

The ability to 'freeze' a transitory waveform, so that it could be measured
or examined at leisure, has been sought for very many years, and early instruments
endeavored to meet this need by the use of long-persistence screen phosphors,
which would continue to display the screen waveform, though at a much reduced
brilliance , for a minute or more following switch-off. An improvement on
the performance of long-persistence screens was obtained by the use of a dual-phosphor
screen coating, in which a normally dormant phosphor layer could be reactivated
to display its last image, by flooding it with infra-red radiation. However,
contemporary digital technology, and fast A/D converters, offer the ability
to capture and store the actual signal in numerical form - information which
is refreshed each scan. In the technique used, the incoming signal is repetitively
sampled, at a rate which is high in relation to the horizontal sweep speed,
and the instantaneous signal voltage level is stored for a time which is long
enough for its value to be noted and transformed (quantized) into a digitally
encoded form.

This instantaneous numerical equivalent to the input signal amplitude is
then stored in a fast random access memory store, while the output from a
digital-to-analogue converter (DAC) is simultaneously displayed on the oscilloscope
screen. The advantage of this type of display - apart from the convenience
of being able to examine a possibly transient waveform at one's leisure -
is that, once the signal is in digital form, a wide range of mathematical
manipulations may be performed on it. An example of this is the Fast Fourier
Transform process noted above in relation to spectrum analysis. The stored
signal can also be fed to a computer or printer for an immediate hard copy.

It’s also common, with modern digital storage oscilloscopes, to use the microprocessor
system within the control electronics to show the X-axis time scale on the
'scope display, as well as the DC and AC parameters, including frequency,
of the input signal(s). Input signal sampling rates of up to 500 Mega-samples/s
are available in top-range instruments - though these are expensive. Vertical
resolution equivalent to '8-bit' encoding (±0.15 mm on a 100x125 mm tube face
size) is typical.

Modern test instrumentation of the various kinds described above allows a
great deal of information to be obtained about the electrical characteristics
of audio equipment. However, as indicated at the beginning of this SECTION,
our understanding of the relationship between the measured performance and
the quality of sound produced by the equipment under test is still less comprehensive
that we would wish, and innovations in technology, particularly in the field
of digital audio, continually pose new questions about the acoustic significance
of those imperfections which remain.