So far, OPT-1A Core = T44 x S62mm, Np = 2,320t, wire =
0.355mm Cu dia, RwP = 117r, Primary
winding loss = 1.44% with nominal RLa-a = 8k0.
Primary turns per layer = 145t for 16 primary layers. Fsat =
14.1Hz at 632Va-a, 1.6Tesla.
Secondary = 4 layers each 51t, 1,06mm Cu dia, one layer = 3 x 17t
wound trifilar.
RwS for 4 // 51t = 0.072r gives Sec loss = 1.8%.
ALTERNATIVE SINGLE SECONDARY FOR ALL SPEAKERS ABOVE 4r0.39.If the amp is to be only used with speaker
loads above 4r0, and there are no demands
for high class A Po, the secondary may be configured to give only
ONE load match, and
it will be for the Middle primary RLa-a 8k0 and secondary between
5r0 and 6r0, say 5r0.
It is usually impossible to get an exact load ratio of exactly 8k0
: 5r0, 1.600 : 1.0.
Turns per layer are determined by layer filling, winding losses,
and wire sizes, but usually a ZR
giving 8k0 : 4r7 to 5r6 will do, so TR may be between 42 and 38 to
1.0. To achieve this,
subdivided sec windings are used, but once links are soldered
there will be no need to ever
alter the OPT link settings.

Table 1. OPT-1A, possible load matches for single sec
links.

Pri 2,320t
RLa-a

Sec
51tpl,
4 // 51t, 3 // 68t

Sec
54tpl,4 // 54t,
3 // 72t

Sec
57tpl,
4 // 57t, 3 // 76t

Sec
60tpl,
4 // 60t, 3 // 80t

8k0

3r9

6r9

4r3

7r7

4r8****

8r6

5r4****

9r5

7k0

3r4

6r0

3r8

6r7

4r2

3r7

4r7****

8r3

6k0

2r9

5r2****

3r2

5r8

3r6

6r5

4r0

7r1

5k0

2r4

4r3

2r7

4r8****

3r0

5r4****

3r4&

5r9

4k0

1r9

3r4

2r2

3r9

2r4

4r3

2r7

4r7****

This table shows **** at each best single sec arrangement to suit
all RL > 4r0.
The tubes used would be 2 x KT88 and have about same idle Pda but
lower Ea for each RLa-a shown.
If you used the original OPT-1A design with 4 x 51tpl secs, then
Ns = 3 // 68t will give 6k0 : 5r2.
Where the tubes have lower Ea to suit 6k0, There is fair
performance for all speakers above 4r0.

Fig 1. OPT-1A for UL or triode or beam tetrode, pentode (
and possible CFB ).
Fig 1 has my preferred 50% UL taps, also found in Leak amps.
Mullard said 43% was the correct %.
RCA used 30%. There is no strict rule for UL tap % which should be
between say 35% and 55% to be
most effective at reducing 6550 beam tetrode Ra from a typical 33k
at Ia 50mAdc, down to about 2k2
at 50% UL taps. I have used 60% UL taps for 6550 which reduces Ra
to about 2k0, and allows 45W
AB of beam tetrode power with RLa-a 8k0.

The B+ = Ea = +500Vdc, and this is what I regard as maximum for
UL with 6550, KT88, KT90, KT120.
Fixed bias is best.

OPT-1A does allow for UL taps at 25%, 37.5%, 50%. The 50% UL THD
is as low as triode for same initial
class A Po.

Table 2. OPT-1A, Triode KT88, 6550 etc Ea +500Vdc, Ia
50mAdc ea.

RLa-a
2,320t

Ns
4 // 51t

Ns
3 // 68t

Ns
2 // 102t

Po
Max

Va-a
Vrms

4k0

1r9

3r4

7r7

40W

410

6k0

2r9

5r1

11r6

34W

466

8k0

3r9

6r9

15r5

29W

494

16k0

7r7

13r7

30r9

21W

580

OPT-1A gives higher amount of class A in all modes with Ea
+400Vdc, and KT120 allows Ia = 87mAdc.
Alternatively, use of 4 x EL34, each with Ia 50mAdc also gives
splendid class A operation.
With lower Ea, the maximum Class AB for UL is about 50W for 4k0.

41.Fig 2. OPT-1A for 12.5% CFB ( and possible UL ).
Fig 2 shows OPT1A with two primary layers 6-7 and 12-13 used for
12.5% CFB windings.
The 12.5% means that 2 primary layers of a total 16 layers will
give Vac at cathode
= 12.5% of total Vac between anode and cathode. This Cathode
Feedback alone will reduce
tetrode tube Ra to less than any use of UL taps. However, the
screens may be connected to
primary taps at 5 and 14 where there is 70Vrms- where Va =
245Vrms-, and Vk-0V = 35Vrms+.
Therefore Vg2-k = 90Vrms-, and the tube has 37.5% UL taps but also
has the 12.5% CFB.
The result is a low gain output stage with very effective Ra but
spectral content that is probably
best. UL taps to OPT will make Eg2 = Ea, and Eg1 needs greater
-Vdc bias.
When I first built the 8585
amp in 1995 with Ea +400V and fixed bias and CFB and
UL taps, and with a
similar OPT, and with 4 x GE6550A, I was able to get 50W AB and
THD = 0.7% without any GNFB.

I later found that Ea could be increased to +480Vdc with screens
taken to a fixed Eg2 rail at +330Vdc.
The increased Ea allowed bigger Va swing and with 4 x KT90 the amp
could easily make 100W class AB.
But the first 20W is all important for hi-fi amps.

I have placed +++ against load matches I might approve of. The
output Po is what you should get
at Sec output, and you can see the Va-a is all above 820Vrms,
severely dangerous.

See OPT-3A at bottom of this page for a slightly different
OPT for high Ea.

It uses same core T44mm x S62mm and same primary winding but
secondary has less turns of thicker
wire to get higher TR and ZR so suite the higher Va swing. I have
never ever wanted to use any octal
based tubes with Ea higher than +500Vdc, and I think its always
better to use a quad of tubes with lower
Ea than a pair of tubes with higher Ea. OPT-3A is for those poor
souls who believe maximum Power is
most important, even though they rarely ever use more than 2W of
average Po from their amps.

Notice OPT-A has 0.38mm Nomex insulation between 8 P-S interfaces
and 2 P-P interfaces
between anode and cathode windings with 500Vdc difference.
Previously, I specified 0.5mm polyester, but nearest Nomex size is
0.51mm and 0.38mm is nearest
size below 0.5mm to allow easy winding without having to cramp the
wound bobbin when
winding is complete to avoid the bulge making it difficult to
insert Es into the wound bobbin.

There is a slight increase of shunt capacitance insulation reduced
from original 0.5mm to
0.38mm, but Nomex data shows dielectric constant is less for
0.38mm than for 0.51mm, so
shunt C increase is not alarming.
----------------------------------------------------------------------------------------------------------------------------43. Fig 3 below was first drawn over 10 years ago. I have
tidied it up and added more useful
information.

44. OPT-1A, Height of bobbin contents of OPT-1A was
calculated in step 42 above.
The total height of all contents in the bobbin should not exceed
0.8 x core H size
= 0.8 x 22mm = 17.6mm. The listed heights above for bobbin content
= 15.8mm, and the
bobbin base is estimated at 2.1mm thick so height of all things in
window = 17.9mm.
But the cheeks of the bobbin extend higher than the bobbin
contents so clearance between
top of wound content and core = 22mm - 17.9mm = 4.1mm. The wire
will bulge at it is
wound around the rectangular core but it should be was to insert
Es to bobbin window
without fowling cover insulation. Once Es and Is are assembled
into wound bobbin and bolts
tightened pieces of phenolic sheet should be pushed into any gaps
to prevent any movement.
Scrap kitchen bench covering 0.6mm thick is ideal, from a kitchen
joiners rubbish bin.

In a number of OPTs I wound, I painted on slow curing epoxy
varnish to every insulation
surface and winding layer liberally. At end of a day's winding I
cramped the windings tight
with G-cramp and plywood squares while bobbin was still clamped
between lathe clamps.
This forced excess epoxy out, or into voids unfilled, and it was
left for a day or two until
the wound bobbin was quite solid.
This method usually gave me plenty of clearance to easily insert
Es.

You need to fill the bobbin window efficiently, but not have wire
and insulation too high
to work with.
-------------------------------------------------------------------------------------------------------------------45. Core Permeability, µ.
µ is just a number, with no units. µ is used often and context
MUST be remembered.
For transformer cores, µ is permeability or the core, but for a
triode, µ is the amplification factor.

For an air cored coil, µ = 1.0, and very little different
to the µ of the coil in a vacuum.
For an air cored coil, the magnetic field strength for a given ac
or dc current below 1kHz is quite
low compared to the same coil with an iron core. So the air cored
coil has much lower inductance
than the same coil, but fitted with an iron core. The iron cored
coil may have many thousands of
times more inductance than an air cored coil. For a toroid core
with a wound strip of thin GOSS
material, µ may be 40,000, ie, there is 40,000 times more
inductance with GOSS core than
without it, roughly speaking. I don't have time for a page full of
hideously complex equations about
each statement I make, but if buy an air cored speaker crossover
coil of say 3mH, and add a bar
core taken from a fused PT, you should find the core measures
12mH, because the bar core aids
the production of stronger magnetic field and increases the
inductance. Adding an iron core which
forms a continuous loop using overlapping laminations, the
inductance may increase to 1,000mH.

GOSS or CRGO E+I, C-cores and other shapes of cores have
maximum µ between about 5,000
and 17,000. The µ for GOSS has increased from 1947 when max µ was
about 5,000 to maybe
17,000 in 2017, all due to changes made to rolling and heat
treatments for the silicon steel.
Thus modern E+I or C-cores made after 1990 will usually have max µ
> 10,000, and this means
the cores of PT do not get as hot as something made in 1937. It
also means there is much less
distortion current produced in OPT transformers which are expected
to work down to 20Hz.

However, there are strict limits for Vac level and Frequency
because all metallic
cores are limited to their maximum field strength, Bac, which can
be between 0.8Tesla and
2.5Tesla for exotic materials. GOSS usually may have max Bac =
1.6Tesla.

Graph 1. Core permeability of GOSS with max µ = 5,000.
Graph 1 gives results of a test to plot the µ vs Bac for an old
sample of GOSS E&I lams which may have
been similar to the core material used for the 1947 Williamson.
The sample I have used here was taken
from a fused PT or OPT and was wasteless pattern E+I with T28mm.
The lam thickness was 0.5mm.

The core sample for my test will not compare well with what you
may find in OPTs made after 1980.
In 1950s, GOSS lams had max µ = 5,000, but I have used E+I lams
made by Sankey in Australia in about
2001 with max µ = 17,000. The E+I power transformers I wound using
this material barely got warm.
But at some time before 2010 Sankey closed their business due to
plummeting demand caused by cheap
imports of transformers mainly from China where wages are 1/10 of
any western nations and with
appalling work conditions.

The reason for drawing these graphs is to show what partial air
gapping can do to reduced the max µ
to make a core for a PP amp less prone to Idc magnetization which
may cause horrible sound is the net
Bdc for a PP core exceeds about 0.1Tesla due to the two output
tubes having Idc difference exceeding
15%. Most GOSS E+I cores and C-cores may have max µ > 10,000
and the technical performance of the
core is better than it has to be, and very prone to Idc non
balance in output tubes unless there is an Idc
balancing circuit with a pair of LED which should glow equally
bright when balance is within +/- 5%.

I found the sample of old core I used would not like working with
maximum intermeshing of lams above
0.7Tesla. Despite the max µ = 5,000, it occurs at 0.5Tesla, and
for a PT this material would be useless
now if used with typical Bac 1.3T at 50Hz. In a PP OPT, this
material would show signs of core saturation
at only 0.7T. It would be impossible to achieve a good OPT with
Fsat 14Hz and you could only ever get
Fsat at about 32Hz with the design formulas at this website which
is based on core material able to have
Bac max = 1.6T.

If you are going to make your own OPTs, and use old second hand
cores, please test the material and
understand what you are doing. Don't ever blame me for the poor
transformer performance if you have not
tested the core and you don't know what you are doing.

For Curve A, the direction of E+I alternates for each
consecutive E; this is known as maximum lamination
intermeshing aka interleaving, and it always gives the maximum
possible µ for E+I lams.

For Curve B, to obtain results for "partial air gap", I
re-arranged the laminations so E+I were divided into 3
sub-stacks each with 7 + 6 + 7 E+I lams, with the Es in each
consecutive sub-stack facing the opposite
direction as the total stack was built up.

The results show the same applied Vac, and records the calculated
Bac, XL, L and µ.

You can see the µ is not constant, and hence the inductance
changes for different levels of Vac.
The L changes during each wave cycle, and the change of XL
reactance causes distortion currents to be
created, mostly 3H, so XL must be kept high and well above the amp
RL value , even at low Vac levels
where XL is lowest, and for low audio F. Most transformer cores
are tested with mains F 50Hz or 60Hz,
and I used a sig gene with 47Hz and an audio amp.
Iron caused distortion is lowest for Bac < 0.32Tesla. µ may be
highest where Bac is between 0.5T and 1.5T.

If Vrms is at maximum for coil, and Bac max = 1.6T at 14Hz, you
have some good core material and the Bac
at 50Hz = 0.45T for the same Vrms and THD in bass music notes will
be quite low.
--------------------------------------------------------------------------------------------------------------DC MAGNETIZATION OF PP OPT CORES
PP OPTs have Idc input to a CT on primary winding and equal Idc
flows in each 1/2 primary to each anode.
The direction of turns for the whole primary is the same but there
is opposite direction of Idc flow direction
in each 1/2 primary so the Idc flows causes magnetizing fields
which oppose one another so that there is
virtually no Idc magnetizing effect.
But where one Idc flow is more or less than the other, there is
unequal Idc and the difference between the
two Idc can cause partial or total Idc magnetization of the core.
In other words, the core becomes magnetized
like any electromagnet which is powered with Idc flow. For PP OPT,
best signal operation with low THD is
achieved with equal Idc flows in each 1/2 primary, but in the real
world there is always some slight inequality
so all PP OPTs do have some slight Idc magnetization, but it is
benign, and won't affect the music at all
if Idc both tubes is high enough at say about 50mAdc, and the Idc
difference is less than + / - 3mAdc.

The Idc magnetization of an iron cored coil = Bdc, and is
calculated
Bdc = 12.6 x Idc x N x µ / ( 10,000 x ML ), where 12.6,
10,000 are constants, N = turns, Idc is in Amps dc,
ML = magnetic path length of iron in mm, and Bdc is in Tesla.
But because the difference of Idc in each 1/2 primary winding
causes the Idc magnetization of a winding
on iron core with CT,

The core Bdc cannot exceed about 1.6T for GOSS when it becomes
fully magnetically saturated, so where
calculated Bdc > 1.6T, the real Bdc < 1.7T.
Where calculated Bdc > 1.6T, small changes to B due to Iac will
make the coil behave as if it has no inductance
and as though has only wire resistance. Large Iac changes with
high Bdc present will reduce total B below 1.6T
so that for part of the wave cycle the coil appears to have some
inductance.

For OPT-1A, if µ = 5,000, the Idc to saturate core = 1.6 x
10,000 x 245mm / ( 12.6 x 0.5 x 2,320t x 5,000 )
= 0.054Adc = 54mAdc. In the real world, if one of the two output
KT88 does not conduct Idc, the core will
saturate fully, and the music will still be heard at low levels
but the tube which is conducting is under duress
and if music level is turned up the sound is just horrible.
If the core was a wound toroid of GOSS, its µ may be 40,000, and
the Idc difference needed to saturate core
= 7mAdc, a small amount of Idc. I have often serviced amps where
Iadc in one was say 45mAdc, and 55mAdc
in the other and this would cause full saturation of a toroid
core, and I am not at all in favour of ever using a
toroid OPT core unless it had an air gap to reduce the max µ.

The core with µ = 5,000 could have Idc imbalance = 10mAdc, which
might cause Bdc = 0.3Tesla, and and
although THD would increase, the amp would still work. I have used
E&I with max µ = 17,000, much higher
than needed, but had active circuit to give Idc balance condition
using two LED at front panel.
To calculate the Idc required for core saturation, the max µ value
is used. But the µ varies with Idc or Iac flow
and Curve A above shows max µ = 5,000 but minimum µ is 1,000, so
with a low Idc difference the Bdc will
be lower than calculation predicts.

I have found µ need not be higher than max 4,000 and if it is only
1,000 at very low Idc and Iac levels then
the Lp is still usually high enough to give XLp = RLa-a at below
20Hz.

If the use of E&I sub-stacks reduces max µ to below
4,000, the the Idc imbalance is much better tolerated.

For best operation without Idc imbalance, there should be an
adjustment pot to alter Eg1 grid bias -Vdc
for each tube and a circuit to drive a pair of LED used to
indicate state of Idc balance, as I show in a
number of tube amp schematics.

With C-cores, the partial air gap is really a normal air gap, but
it would be much smaller than for a Single
Ended OPT with a high Idc flow. In a 50W mosfet powered amp I
built before 2000, I used C-cores for a PP
OPT and the µ max was about 5,000 max with cut surfaces tightly
together. With a single layer of plastic
about 0.02mm thick, the µ max was reduced by 1/2, and the
saturation behavior became much less likely
to cause excessive currents in the mosfets at very low F. Bass
performance is excellent,
see solidstateamps4-50w-mono-mosfet.html

If all Es are just butted to all Is, and there is no real air gap,
the change of grain direction in crystal
structure at each join of E to I acts like an air gap and a core
with max µ of say 5,000 may have µ of only
900 when arranged as for an SE OPT, but with no gap. The low µ
obtained this way is usually too low to
be able to get enough primary inductance. The sub-stack air gap
can give the wanted max µ of say 4,000
by altering the number of Es and Is in each sub-stack. It is a
tedious process to stack up Es and Is and
then change the sub-stacks and re-measure everything; it can take
days. Automatic Idc balancing circuits
may not work well because they often become unstable with LF
oscillations when GNFB is used.

46. OPT-1A. Calculate maximum and minimum primary inductance. The minimum Lp is of greatest interest because its value
affects how the tubes work at low F and the
stability of amp at LB with GNFB, especially where there is no Sec
load connected and tube gain is
highest at low levels.

For most PP amps, if minimum Lp reactance XLp = RLa-a at 20Hz,
then all is well, and as Va-a
increases above very low levels the µ rises to perhaps 3 times the
low level value so the F where
XLp = RLa-a can move down from say 20Hz to 7Hz and inductance
shunting effects become negligible.

The core µ is usually highest at lowest F. So if core µ = 843 at
20Hz, and very low Bac with Va-a = 5Vrms,
it should be found that at highest Va-a for say 50W for 8k0 at
20Hz, the µ will be close to high as it can go
with partial air gapping, say 3 x 843 = 2,529.
Therefore if the core with partial air gap has max µ = 4,000, it
is highly unlikely the minimum µ will be less
than 843, because one effect of partial air gapping is to reduce
the range of variability of µ.

if modern core material with max µ 10,000 is used, partial air
gapping may reduce max µ to 4,000 so
minimum µ = say 1,320.

The core could even be NOSS material, and it usually has max µ
3,500 with max intermeshing so there
is no need for partial air gapping and you may find there is
always enough primary Lp to avoid load shunting
by Lp at low F.
I have used NOSS for 85W amps with 4 x KT88 per channel, with Afe
= 44mm x 62mm and Np = 2,080t and
I had no problems during testing where Lp became too low at LF and
a low level.

However, a LF gain shelving network is essential between V1 and V2input / driver tubes to reduce open loop
gain below 20Hz. This always makes the amp recover quickly after
excessive Vin is applied, and prevents
peaks in F response below 20Hz due to total phase shift becoming
high enough to cause the peaking.

Mr Williamson said way back in 1947 that for RLa-a 10k0, minimum
Lp should be 100H, so there should be
10H for each 1k0 of load, so
Minimum Lp = 10H x RLa-a / 1,000, where RLa-a = Nominal value, in
ohms.For OPT1A, Min Lp = 10H x 8,000 / 1,000 = 80Henry.
This is where the core µ will be minimum at say 50Hz.

This minimum core µ is easily reached with GOSS E+I or C-cores
providing Np and
Afe are both large enough, and Afe > 300 x sq.rt Po.

The measured LF response and LF pole is never equal to where XLp =
RLa-a because the tubes have Ra. The -3dB pole for LF response is where XLp = ( load RLa-a
parallel to tube Ra-a ).

Two KT88 tubes with 40% UL may have Ra-a = 5.6k, so 8k0 // 5k6 =
3k3, and if Lp = 80H,
then -3dB LF pole = 3,300 ( 6.28 x 80H ) = 6.6Hz, and this may be
without any GNFB or gain shelving
network and at low levels to avoid core saturation, and the KT88
have grid Vg-g that has flat F response
with -3dB at below 4Hz. To explore VLF response, the coupling caps
between V1-V2 and V2 to KT88 may
need increasing from 0.47uF to 1uF, and C at input
increased.

If 2 x KT88 were used in pure beam tetrode mode with fixed Eg2 and
no CFB, then Ra-a might be 60k
at idle Idc.
Without any sec load, the -3dB F response pole is
where XLp = Ra-a = 60k, and -3dB is at 60,000 / ( 6.28 x 80H ) =
119Hz.

In triode mode, with Ra-a = 2k0, and no RLa-a, -3dB is at 4Hz.

I conclude the F response for any OPT relies on the "total
terminating resistance"
across the primary or secondary which are nominal loads parallel
to Vac source resistance.

Core saturation occurs as a result of applied Vac and frequency
across the inductance of coil, and the
onset of saturation above 1.2 Tesla is most clearly seen on an
oscilloscope where no load is connected.

For OPT-1A, we would find Fsat 7Hz at Va-a = 316Vrms, and 3.5Hz at
Va-a = 158Vrms. Most music
has very little content below 32Hz and bass signals cannot ever be
at the maximum possible Va-a level
because there would be no headroom for all other musical
frequencies. Therefore core saturation is
not a problem in most hi-fi amps even where the Fsat occurs at max
Va-a at say 32Hz.

I have always preferred all my PP OPTs to saturate at 14Hz if
possible because there is less bass
distortion and the bass is subjectively superior. The amp should
have C+R network at input with pole
at say 7Hz to exclude stray signals with very low F. OPTs in amps
can be tested using a pink noise
source. If the pink noise has bandwidth from 3Hz to 20kHz, and
output Vo increased to where clipping
on peaks begins to occur, there can be an irregular knocking sound
in OPT caused by occasional high
levels of LF causes intermittent core saturation. I found adding
the 7Hz HPF minimised the saturation
at very LF. And the sobering fact is that where pink noise begins
to clip, a Vac meter will show average
Vac level is 1/3 of peak level. Music signals are remarkably
similar to pink noise but has some F present
that are harmonically related to others, and there are high
average level changes, but as soon as
music Vac clips, you have reached maxim possible and average level
is 1/3. This means that a 50W
amp happily makes 5W average, and if speakers are 88dB/W/M, 5W
average gives 95dB SPL and
this is way above the 85dB level most ppl can tolerate for more
than 1/2 an hour.

48.Calculate leakage inductance, LL.
The leakage inductance in written specifications is sometimes
described as being "referred to the primary."
This means it is considered to be an inductance in series with the
primary load looking into one end of
the primary with the other end grounded. But with a PP amp, there
are two opposite phases applied
across Pri so there is 1/2 the total LL in series between each
anode and OPT input. Typical good
values of this inductance would be 10mH for RLa-a = 10ka-a, ie,
1mH per 1k0 of RLa-a.

LL = 0.417 x Np squared x TL x [ ( 2 x n x c ) + a ] / (
1,000,000,000 x n squared x b )
where LL = leakage inductance in Henry, 0.417 is a constant for
all equations to work,
Np = primary turns,
TL = average turn length around bobbin = 2T + 2S + ( H x 22 / 7
)mm.
2 is a constant because there is an area at each end of a layer
where leakage occurs,
n = number of dielectric gaps, ie, the concentric gaps between
layers of P and S windings.
c = the dielectric gap, ie, the averaged distance between the
copper wire surfaces of P and
S windings, a = height of the finished winding in the bobbin, b =
the traverse width of the
winding across the bobbin.
Distances are all in mm!

49.Shunt capacitance of an OPT, C. Capacitance in
a tube amp OPT is like having a perfect
OPT with no C and having a C across the primary. This C acts to
reduce the primary load at HF in a
similar manner to how Lp reduces primary load at LF.

C exists between primary turns and between primary layers which
all sum to what is called
"self capacitance" of a primary winding. For a 2,320t winding with
no P+S interleaving, ie, with all Pri
turns together, possible self C = 500pF.
But with 5 P sections separated by 4 S sections, the self C of
each P section may be < 100pF, but
because 5 pri sections 5 are in series, and well apart from each
other, total self C = 20pF, and is a
negligible quantity.

The sectioned P winding is like an RF choke with 5 separated
windings in series with mutually shared
inductance but winding has low self C so resonance of the RF coil
is above the wanted RF band for
the coil.

But in all OPT there is a much larger C between Pri layers next to
Sec layers and it must be minimized
by using thicker insulation than needed to prevent arcing between
B+ and earthy sec.

The C between an anode connection and 0V depends on the Vac
between P and S layers.
Where 2 wire layers have no Vac level or phase difference, there
is is no capacitance. And where each
layer had the same Vac level but opposite phase, C is twice that
for where one layer has Vac and the
other has none, or a negligible amount such as where one layer has
20 times Vac of other layer, as in
an OPT with primary Vac many times secondary Vac. Therefore Vac of
sec may be ignored and sec
regarded as "earthy" and at 0V for shunt C calculations.

At each anode there is C between anode and 0V, Ca-0V, and it is
considered equal at both anodes.
Because each Va is oppositely phased, the C between anodes is the
two Ca-0V in series, so Ca-a =
= 0.5 x Ca-0V.
For a SE OPT the shunt C is between anode and 0V = Ca-0V.

The capacitance between P and S sections is like capacitance between two metal
plates.
C = ( A x K ) / ( 113.1 x d ) where Capacitance is pF,
A is the area in square millimetres of the plates assumed to be of
equal area and shape,
K is the dielectric constant of the material between the plates,
with air being = 1.0.
113.1 is a constant for all equations to work,
d is the average distance in millimetres between the plates for
the whole area of the plates.

For many OPT, turn length varies depending on where the P-S
interface occurs in bobbin height.
But calculations will be accurate enough if the TL is the average
for all turns, ie, 280mm.
Traverse width across bobbin = 62mm, area of each interface =
280mm x 62mm = 17,360sq.mm.

The calculated C between each P-S interface is transformed by the
position of the P-S along
each 1/2 primary winding.
The first P-S interface down from 'anode 1', or the top of the
wind-up at above the GH-IJ-KL is at
a position of 6.5 layers / 8.0 layers along the P winding from the
CT at 0Vac.
Thus position of this C has turn ratio = 6.5 / 8.0 = 0.81, and ZR
= 0.81 squared = 0.66, so 779pF
is transformed to appear at anode as ZR 0.66 x 779pF = 514pF.

There are 4 x P-S in each side of CT, with a total of 8 x P-S for
the whole Pri. If both ends of
Pri were shunted, and CT disconnected from 0V, C between Pri and
Sec = 8 x 779pF = 6,232pF.

But for where opposite phases of Vac are applied to a winding with
CT, Ca-a was calculated 492pF,
which is less than 1/12 of the C for each P-S measured alone.

This all leads to a formula where if there are 4 or more P-S
interfaces along a winding with Vac at
one end only, the effective C from live end to 0V end may be more
easily calculated as :-
Ca-0V = ( No of P-S x C at each P-S ) / 3.0.

This is slightly higher than calculated step by step using ZR
ratios etc, but there is always slightly
more Ca-a due to primary ends being near the core for part of a
turn and other stray C in anode circuitry.

During class A PP action, the shunt Ca-0V at each anode =
1,038pF, and Ca-a remains a constant
519pF. But for class AB where one tube is cut off, the anode
conducting has Ca-0V = 2 x 1,034pF
= 2,068pF because 1/2 the primary is coupled to the other half
connected to tube with zero current.
But R load at conducting anode reduces to 1/4 x RLa-a so the ratio
of C to RLa of tube does not
change so little change to F response occurs.

For OPT-1A in a real amp with enough local and global NFB, the
Va-a at -6dB max level for 1kHz can
be kept constant and power to RLa-a 8k0 kept constant because
tubes can provide the extra anode
current flow in Ca-a at HF by working more heavily into class AB.
Hence it is not unusual to see a tube amp give flat bandwidth
between 5Hz and 70kHz at the -6dB
level and often the -3dB level and a few at the 0dB level. Because
there is so little audio Vac below
20Hz and above 20kHz tubes will never overload at 0dB level
between 20Hz and 20kHz.

FREQUENCY RESPONSE AT LOW LEVELS.F response at low levels depends on amount of NFB which
effectively reduces Ra-a.
Without GNFB the response is determined by Ra-a, and the RLa-a,
and shunting effects of Lp and Ca-a. For OPT-1A with two KT88, RLa-a 8k0, Min Lp = 80H, Ca-a =
519pF.
If the output tubes are driven by Vac with bandwidth 1.0Hz to
1MHz, the theoretical F response for
various modes are :- Table 4.

Mode, all with
RLa-a 8k0,
Low Va-a < 20Vrms

Ra-a

Ra-a
// Ra-a

F1

F2

Pure Tetrode,
fixed Eg2

64k

7k1

14.1Hz

43kHz

40% UL

5k6

3k3

6.6Hz

92.5kHz

Triode, g2 to
anode

2k2

1k7

3.4Hz

179kHz

20% CFB, fixed
Eg2

1k5

1k3

2.6Hz

466kHz.

This table shows what is possible if HF there is no series
resonance with Ca-a which usually causes
a dip in response at Fo. For LL 4.2mH and Ca-a 519pF, Fo =
108kHz, so the lower two F2 of 179kHz
and 466kHz are very optimistic, but F2 at least 100kHz looks white
possible, but in my 300W amps
with 12 x KT88 and with 40% UL, I did get 270kHz.

PARALLEL L+C RESONANCE AT MID FREQUENCY.Lp and Ca-a both shunt the whole OPT primary, and are in
parallel, and produce an Fo at between say
300Hz 2kHz. The core µ reduces and so does Lp as F rises and the
only way to find out the Fo is by
measuring it in a test circuit. Suppose Fo = 2kHz. At 2kHz, XCa-a
= 153k = XLp, so Lp = 12H. You should
find the OPT causes 0.0degrees of phase shift at 2kHz. And at
2kHz, you should find Primary Zin maybe
4 times higher than 153k because the parallel L+C network always
has higher Z at Fo than either XC
or XL which are always equal at Fo. If Va-a is high, core µ goes
higher, Lp increases and Fo reduces.
I have measured OPT where the Fo is 350Hz, suggesting plenty of
Lp, but Ca-a is too high because of
excessive number of P-S interfaces and with insulation not thick
enough for low C.

--------------------------------------------------------------------------------------------------------------50. OPT-1A LCR MEASUREMENT. I don't have a OPT-1A to test, so I can only predict what you
might find above 1 to 49 steps.

The Theory page explains whatever I may have omitted in this page.
-----------------------------------------------------------------------------------------------------------
But for those who wish to plot the F response, here is a blank
sheet you may download and print.

CATHODE FEEDBACK WINDINGS.
CFB windings will complicate the capacitance calculation because
some of C between anodes appears at
cathodes Ck-k, The exact interaction with the split portions of
Ca-a and also LL is difficult to predict.
But for a well interleaved OPT with say 5S+4P, there is little
sign of any HF oscillation. I found the best
in a wound bobbin for say 2 x CFB layers and say 14 Anode layers
was with CFB layer between an anode
layer and a Sec sec layer. The typical amount of CFB I like to use
is between 12.5% and 20% of the total of all
Pri turns.
Consider 20% CFB with a KT88 with a fixed Eg2. This makes KT88 act
like a triode with µ = 4.3, and Ra = 710r.
But instead of the electrostatic shunt NFB within the triode, the
external loop FB is via linear transformer windings
and is series voltage FB so the THD is less with CFB than a real
triode, the nearest type would be 300B.
But the KT88 acts las though it has 20% UL tap, which would give
Ra 5k2 and UL µ = 31.2, but with ß = 0.2
and the UL µ is reduced to CFB µ = 4.3; the µ reduction by NFB = 1
/ 7.25, or 16dB NFB.
But with a class A load RLa-k = 4k0, the amount of NFB = 12dB
approx, so THD reduction factor = about
0.25. The PP connection with 2 x KT88 PP and CFB for 10W class A
will give THD < 0.5%, and with 10dB
GNFB, THD < 0.16%. it s GF,with 8k0 R, so if there
is 5% THD o,15Fof local series voltage NFB.
Thus the majority of error correction is done in the output stage
but there is a tendency for HF oscillations
above 100kHz where there 2 cascaded applications of NFB, and open
loop gain > 1.0, and phase shift >
180 degrees.
I found that placing Zobel networks using 2n2 + 2k2 across each
anode portion of 1/2 primary helped to
prevent HF oscillations. The use of CFB windings does make the
shelving networks used between input V1
and driver V2 stages more effective and easier to configure. There
is little point to trying to further explain
the increased complexity with CFB, but it works well. Regardless
of CFB windings, the Ca-a should be
minimised as much as possible.

The effect of Ca-a reducing bandwidth is reduced by the CFB so F
response of an OPT with CFB and
no R load may well over 100kHz.

Fig 5. Shelving networks for typical amp, OPT 8k0 : 8r0.
I have had many emails from those who find their amplifier
oscillates badly even with only 6dB GNFB.
This much NFB may make the sound worse, and 20dB should be
possible for amp with 40% UL taps
for 2 x KT88. The values of following R + C must be optimised :-
C1+R1 for HPF pole 5Hz.
C2+R4, for advance of phase of Vac fed back.
C4+R5, for reducing HF gain V1 by -21dB between 32kHz and 340kHz.
C5+R6, for reducing LF gain V1 by -15dB between 25Hz and 2.5Hz.
R12+C10, for providing 5r7 load at above 100kHz to load the KT88
where speaker has
Z value much above its nominal value above 20kHz.

There is much more about gain shelving and F response with and
without GNFB at basic-tube-3.html

Values of primary inductance, LL leakage inductance and C8+C9
shunt C are approximate, and cannot
be changed in any existing OPT. The real model on an OPT such as
OPT-1A has many C and is is difficult
and confusing to try to any more modelling that the simple diagram
shows above.
The shelving of LF and HF gain with shelving networks will allow
GNFB to be applied with poor quality OPT
with a low primary Lp and 10 times the LL shown above. The F
response at 1/2 full Po should have -3dB poles
at 5Hz and 65kHz where output load is correct, in this case 8r0.
The R+C networks achieve
low overshoot and reduce ringing on 10kHz square waves when a 0.22
uF is connected
across ouput without any R load. At 1/10 full Po, uses of any C
load between 2uF and 0.1uF
should be tolerated with response peak less than +6dB, and less
than +3dB above 20kHz
where nominal 8r0 load is also connected.
--------------------------------------------------------------------------------------------------
OPT-3A for high Po.

Fig 6. OPT-3A for 155W for 4k0 : 3r0 and 6r8.OPT-3A is for KT120 only, to give up to 155W class AB1. At
idle, KT120 has 50mAdc for Pda 35W.
With Va-a = 848Vrms for RLa-a 5k0, Fsat = 20Hz, OK because the amp
for high Po will never be
subjected to high bass levels.
Ea at +700Vdc is allowed because RLa-a is fairly high and lowest
RLa = 1k0, Ia pk is less than 580mApk,
and which KT120 can do, providing Eg2 is high enough at about
+440Vdc.
OPT-3A is very similar to OPT-1A, can be arranged with lower Ea =
+500V and a quad of KT88, 6550
for Po = 100W, but much more initial class A Po.
--------------------------------------------------------------------------------------------------------------------------------------

SOME GENERAL NOTES ABOUT OPTs. For those who struggle to
see what is inside a
transformer with layered windings :-Fig 7.
Fig 16 shows a cross section through a hypothetical transformer
with concentric layered windings
neatly wound with an interleaving pattern of 2P + S, aka P-S-P ie,
with two primary sections and
one secondary section located between each primary section. Fig 16
shows 4 layers of primary each
with 14t and 1 layer of secondary with 10t.

TR = 56 / 10 = 5.6. The ZR = TR squared = 31.36.

There is only one section of secondary, but if the bobbin space
permitted, here could have been two
more secondary layers, one above and below existing primary
sections, and interleaving pattern
could be S-P-S-P-S. The HF response will be far better with the
additional P-S interfaces.
The number of LAYERS must NOT be confused with the number of
SECTIONS. A winding "section" is
one or more layers of wires devoted solely to either primary or
secondary current. There is no direct
connection between the sections for primary and sections for
secondary. Because P and S are not
connected, this is an "isolation transformer".

But the primary could be connected in series with secondary which
would give just one winding of 66t.
This makes the transformer into an "auto-transformer." Where this
is done, the TR becomes 66t / 10t
= 6.6 and ZR = 43.56.
There is no primary or secondary, and there is only unshared part
of winding, 56t, and and shared part
of winding, 10t. The current in shared 10t = current in load for
10t - current in unshared. This makes the
auto-transformer more efficient with less copper losses than same
turns used on isolation transformer.

However, isolation transformers are much more common than
auto-trans because the isolation is needed
where windings have different Vdc or where the transformer is for
a PSU powered by mains, which
MUST NOT have direct connection to any secondary or 0V rail etc.

All isolation trans have P and S connected by shared magnetic
field only. Auto transformers rely on
the same magnetic field but allow Vac of two windings to be in
series connection. The auto transformer
needs to have its shared portion of winding well interleaved with
the unshared part or else the HF response
is just as poor an isolation transformer with little
interleaving.

All tube amp OPTs have a primary load between say 100r and
10,000r, and secondary load of say 8r0,
so ZR varies between 12.5 : 1 to 1,250 : 1 to give TR between 3.53
: 1 and 35.3 : 1. To get good F
response there will always be at least 2 sec sections of one layer
of thick wire with at least one section
of primary with many more turns of thin wire. The S-P-S
arrangement may work OK for a small 6W OPT
for a single 6BQ5. But for a 300W amp with many tubes there may be
5 Pri sections each with 2
layers, and 6 Sec sections with one layer of thicker wire.

Each layer of secondary wire may be subdivided into "secondary sub
sections" to allow varied series
and parallel connections to give variable load matches to suit a
wide range of speaker ohm load values,
while keeping the anode load fairly constant and optimal.

There are no designs here which require rectangular section wire.
Bifilar or trifilar windings have the
advantage of not having wires coming from a winding end between
bobbin cheeks.
--------------------------------------------------------------------------------------------------------------------------WINDING AND VARNISHING. Layers of wire and insulation will
never be tight enough to prevent
small movements of wire due to audio signal generating magnetic
forces which move wires to
cause sound to be produced by OPT. I suggest it is important to
apply slow setting epoxy varnish
by brush to all windings and all surfaces of insulation as the
bobbin is wound. All things needed to
wind an OPT should be immediately available, such as cut to size
insulation.
This keeps the time spent winding to a minimum which should be
less than 8 hours. Mobile phones
MUST be switched off, family life abandoned. The epoxy varnish
applied at 9AM will still be soft at
5PM, bulging windings can be clamped tight with G-clamp and neatly
sized wood blocks, and the
windings are flattened, and and allowed to cure for 2 days. Liquid
varnish in bulge will squeeze
around to fill voids and excess will drip over bench to make a
mess.
Clean up the mess up before it goes rock hard.

The end result Calls the Angels to their happy task of delivering
great music to ears of us poor mortals.

The practice of hand winding OPT and varnishing as you wind can be
extremely messy and epoxy
fumes given off by varnish can be quite toxic so a chemical mask
must be worn.
There should be fume extractor fan. Wattyl 7008 two pack epoxy
polyurethane floor varnish is OK,
and has long enough pot life to allow a full day for winding
OPT-1A. This varnish kept getting onto my
hands and I needed a bowl of methylated spirits and roll of
kitchen tissue paper to keep hands clean
as I wound. To gain winding skills, there is no substitute for
practice, and I suggest beginners learn
by winding a perfectly layered and insulated choke before they
even think about an OPT.
----------------------------------------------------------------------------------------STRAPPING PATTERN LABELS. Strapping patterns 4A and 4C have
board terminals labelled with
capital letters. Thus any terminal board can have 26 labels which
is usually enough, without using
double letters, AA BB etc. For my OPTs in 300W amps, there were 12
sec windings needing 24 labels.
A strapping pattern for "16r0" is not shown because it is unlikely
to ever be used. I have always used
numbers for primary connections with high volts, and letters for
secondary connections with low volts.
This avoids confusion when winding or when servicing the amp.
See 300w-monobloc-about.html
and output-trans-winding.html

STRAPPING TERMINALS. Where OPTs are made "open frame" and
not potted, they should have terminal
boards on each side of bobbin with one side for primary
connections with dangerous high volts and the
other side with a board shown above. Open frame transformers MUST be covered with a box screwed to
the chassis.
The best OPTs are potted in their own steel sheet box which helps
screen them from stray magnetic fields
from power transformers.

TRANSFORMER POTTING. Try Googling "potted audio
transformers".
A potted transformer must be bolted to steel sheet pot cover and
have a 3mm thick terminal board well
attached to transformer and with turrets or 2mm brass bolts and
nuts acting as turrets for connection of sleeved
wires from transformer or amp wiring.
The potted transformer should have 4 protruding bolts to fix to a
chassis or have nuts welded to transformer
frame or brackets to allow bolts to screw into nuts when on the
chassis. The chassis must have a pre-cut hole
to allow the terminals to project into sub chassis area to allow
easy wiring. High Vac terminals should be well
positioned at 10mm centres to avoid large Vdc of Vac difference
between any terminals. The low Vac Secondary
winding terminals must be carefully laid out and all labelled
permanently and clearly.

All parts of the cores should be 10mm away from sides and top of
pot, and board made smaller than pot plan
area to allow potting mix to be poured in until it fills the pot
to underside of board. Before potting, the transformer
must have been well varnished and very dry. Once the potting is
done, there is no way future access can be
gained to transformer.

A suitable potting compound is made by an Australian company
Chemtools, and is sold from industrial
suppliers in many locations. For those in Canberra they might
try M&G Industrial Supplies, 02 6280 7517.
The product you need is PTC-7000, Silicon Potting Compound
in packs of 200grams, 1Kg, 5Kg.
This material will probably adhere well to clean bare steel pots.
Mains and rectifier or audio frequency magnetic
fields can vibrate steel pots or boxes covering transformers,
hence the need for the potting mix to adhere to pot.

Chokes should also be potted. Access to windings is only possible
with amp upside down on a carpet lined bench.
The strapping alterations always makes non technical owners
extremely worried, and if there is a way to fuck up
the strapping, and causing a short circuit, amp owners will
definitely find it.

So you should understand that a great deal of thoughtful
hand-crafting and tradework is involved with making
good OPTs and PTs and chokes.
------------------------------------------------------------------------------------------------Forward to PP OPT Calcs Page 4