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Abstract:

For frequency bin error estimation, multiple hypotheses are formed for
different frequency bin errors, pilot offsets, or combinations of
frequency bin error and pilot offset. For each hypothesis, received
symbols are extracted from the proper subbands determined by the
hypothesis. In one scheme, the extracted received symbols for each
hypothesis are despread with a scrambling sequence to obtain despread
symbols for that hypothesis. A metric is derived for each hypothesis
based on the despread symbols, e.g., by deriving a channel impulse
response estimate based on the despread symbols and then deriving the
metric based on the channel impulse response estimate. In another scheme,
the extracted received symbols for each hypothesis are correlated, and a
metric is derived based on the correlation results. For both schemes, the
frequency bin error and/or the pilot offset are determined based on the
metrics for all hypotheses evaluated.

Claims:

1. An apparatus comprising: a wireless receiver, wherein the wireless
receiver comprises: a processor configured to perform despreading of
received symbols with a scrambling sequence for each of a plurality of
hypotheses, to derive a metric for each hypothesis based on despread
symbols for the hypothesis, and to determine a frequency error based on
metrics derived for the plurality of hypotheses; and a memory coupled to
the processor; wherein the processor is configured to derive a channel
impulse response estimate for each hypothesis based on the despread
symbols for the hypothesis, and to derive the metric for each hypothesis
based on the channel impulse response estimate for the hypothesis.

2. The apparatus of claim 1, wherein the processor is configured to form
the plurality of hypotheses for a range of frequency errors, wherein each
hypothesis corresponds to a different hypothesized frequency error.

3. The apparatus of claim 1, wherein the processor is configured to form
the plurality of hypotheses for a range of frequency errors and for
multiple pilot offsets, wherein each hypothesis corresponds to a
different combination of frequency error and pilot offset.

4. The apparatus of claim 1, wherein for each hypothesis the processor is
configured to extract the received symbols for subbands determined by the
hypothesis, and to perform despreading of the extracted received symbols
with the scrambling sequence.

5. The apparatus of claim 4, wherein the extracted received symbols are
hypothesized to be for a scattered pilot sent on different sets of
subbands in different symbol periods.

6. The apparatus of claim 4, wherein the extracted received symbols are
hypothesized to be for a continual pilot sent on a predetermined set of
subbands.

7. The apparatus of claim 1, wherein the processor is configured to
derive the metric for each hypothesis based on energy of a largest
channel tap in the channel impulse response estimate for the hypothesis.

8. The apparatus of claim 1, wherein the processor is configured to
identify large channel taps in the channel impulse response estimate for
each hypothesis based on a threshold, and to derive the metric for each
hypothesis based on energy of the large channel taps for the hypothesis.

9. The apparatus of claim 1, wherein the received symbols are for data
and pilot transmitted using orthogonal frequency division multiplexing
(OFDM).

10. The apparatus of claim 1, wherein the received symbols are for data
and pilot transmitted using single-carrier frequency division multiple
access (SC-FDMA).

11. A method comprising: performing despreading of received symbols with
a scrambling sequence for each of a plurality of hypotheses; deriving a
metric for each hypothesis based on despread symbols for the hypothesis;
and determining a frequency error based on metrics derived for the
plurality of hypotheses. wherein the deriving the metric for each
hypothesis comprises: deriving a channel impulse response estimate for
each hypothesis based on the despread symbols for the hypothesis; and
deriving the metric for each hypothesis based on the channel impulse
response estimate for the hypothesis.

12. The method of claim 11, further comprising: forming the plurality of
hypotheses for a range of frequency errors and for multiple pilot
offsets, wherein each hypothesis corresponds to a different combination
of frequency error and pilot offset.

13. The method of claim 11, further comprising: forming the plurality of
hypotheses for a range of frequency errors, wherein each hypothesis
corresponds to a different hypothesized frequency error.

14. The method of claim 11, further comprising: extracting the received
symbols for subbands determined by the hypothesis; and performing
despreading of the extracted received symbols with the scrambling
sequence.

15. An apparatus comprising: means for performing despreading of received
symbols with a scrambling sequence for each of a plurality of hypotheses;
means for deriving a metric for each hypothesis based on despread symbols
for the hypothesis; and means for determining a frequency error based on
metrics derived for the plurality of hypotheses. wherein the means for
deriving the metric for each hypothesis comprises: means for deriving a
channel impulse response estimate for each hypothesis based on the
despread symbols for the hypothesis; and means for deriving the metric
for each hypothesis based on the channel impulse response estimate for
the hypothesis.

16. The apparatus of claim 15, further comprising: means for forming the
plurality of hypotheses for a range of frequency errors and for multiple
pilot offsets, wherein each hypothesis corresponds to a different
combination of frequency error and pilot offset.

17. The apparatus of claim 15, further comprising: means for forming the
plurality of hypotheses for a range of frequency errors, wherein each
hypothesis corresponds to a different hypothesized frequency error.

18. The apparatus of claim 15, further comprising: means for extracting
the received symbols for subbands determined by the hypothesis; and means
for performing despreading of the extracted received symbols with the
scrambling sequence.

19. A non-transitory computer readable medium containing software that,
when executed, causes the computer to perform the acts of: performing
despreading of received symbols with a scrambling sequence for each of a
plurality of hypotheses; deriving a metric for each hypothesis based on
despread symbols for the hypothesis; and determining a frequency error
based on metrics derived for the plurality of hypotheses. wherein the
deriving the metric for each hypothesis comprises: deriving a channel
impulse response estimate for each hypothesis based on the despread
symbols for the hypothesis; and deriving the metric for each hypothesis
based on the channel impulse response estimate for the hypothesis.

[0003] The present disclosure relates generally to communication, and more
specifically to techniques for performing frequency synchronization in a
communication system.

[0004] II. Background

[0005] Orthogonal frequency division multiplexing (OFDM) is a
multi-carrier modulation technique that can provide good performance for
some wireless environments. OFDM partitions the overall system bandwidth
into multiple (K) orthogonal frequency subbands, which are also called
carriers, subcarriers, tones, and so on. With OFDM, each subband is
associated with a respective carrier that may be modulated with data. In
the following description, "subband" and "carrier" are synonymous terms
and are used interchangeably.

[0006] In an OFDM system, a transmitter processes (e.g., encodes,
interleaves, and modulates) traffic data to generate modulation symbols
and further maps the modulation symbols to the K total subbands. The
transmitter then transforms the modulation symbols for each OFDM symbol
period to the time domain and forms an OFDM symbol. The transmitter
transmits the OFDM symbols to a receiver.

[0007] The receiver performs the complementary processing on the OFDM
symbols received from the transmitter. The receiver transforms each
received OFDM symbol to the frequency domain to obtain K received symbols
for the K subbands. The received symbols are noisy and distorted versions
of the modulation symbols sent by the transmitter. The receiver typically
performs frequency synchronization to determine frequency error at the
receiver. The frequency error may be due to difference in the oscillator
frequencies at the transmitter and the receiver, Doppler shift, and so
on. Frequency synchronization is challenging in certain channel
environments such as low signal-to-noise ratio (SNR) conditions, fast
fading, and so on. Furthermore, it is desirable to perform frequency
synchronization quickly so that the processing overhead is as low as
possible.

[0008] There is therefore a need in the art for techniques to perform
frequency synchronization in a communication system.

SUMMARY

[0009] Techniques for performing frequency synchronization in a
communication system are described herein. The frequency error at a
receiver may be decomposed into a fractional portion and an integer
portion. The fractional portion is less than one bin and may be estimated
and removed in a manner known in the art. A bin is the spacing between
adjacent subbands. The integer portion is also called frequency bin error
and is an integer number of bins. The frequency bin error may be
estimated using the techniques described herein.

[0010] In an embodiment of frequency bin error estimation, multiple
hypotheses are initially formed for different frequency bin errors,
different pilot offsets, or different combinations of frequency bin error
and pilot offset. A pilot may be sent on different sets of subbands, and
each pilot offset corresponds to a different set of subbands on which the
pilot may have been sent. For each hypothesis, received symbols are
extracted from the proper subbands determined by the hypothesis. The
extracted received symbols are hypothesized to be for (1) a scattered
pilot that is sent on different sets of subbands in different symbol
periods and/or (2) a continual pilot that is sent on a fixed set of
subbands in all symbol periods.

[0011] In an embodiment, the extracted received symbols for each
hypothesis are despread with a scrambling sequence to obtain despread
symbols for that hypothesis. The scrambling sequence is used to generate
the scattered and continual pilots at the transmitter. A metric is then
derived for each hypothesis based on the despread symbols for that
hypothesis, e.g., by deriving a channel impulse response estimate based
on the despread symbols and then deriving the metric based on the channel
impulse response estimate. In another embodiment, the extracted received
symbols for each hypothesis are correlated, and a metric is derived for
the hypothesis based on the correlation results. For both embodiments,
the frequency bin error and/or the pilot offset are determined based on
the metrics for all hypotheses evaluated.

[0012] The frequency bin error estimation may also be performed in other
manners, as described below. Various aspects and embodiments of the
invention are described in further detail below.

BRIEF DESCRIPTION OF THE DRAWINGS

[0013] The features and nature of the present invention will become more
apparent from the detailed description set forth below when taken in
conjunction with the drawings in which like reference characters identify
correspondingly throughout.

[0022] The word "exemplary" is used herein to mean "serving as an example,
instance, or illustration." Any embodiment or design described herein as
"exemplary" is not necessarily to be construed as preferred or
advantageous over other embodiments or designs.

[0023] The frequency synchronization techniques described herein may be
used for various communication systems such as an OFDM system, an
orthogonal frequency division multiple access (OFDMA) system, a
single-carrier frequency division multiple access (SC-FDMA) system, and
so on. An OFDMA system utilizes OFDM. An SC-FDMA system may utilize
interleaved FDMA (IFDMA) to transmit on subbands that are distributed
across the system bandwidth, localized FDMA (LFDMA) to transmit on a
block of adjacent subbands, or enhanced FDMA (EFDMA) to transmit on
multiple blocks of adjacent subbands. In general, modulation symbols are
sent in the frequency domain with OFDM and in the time domain with
SC-FDMA.

[0024] For clarity, the techniques are specifically described below for
two exemplary OFDM-based systems that implement Digital Video
Broadcasting for Handhelds (DVB-H) and Integrated Services Digital
Broadcasting for Terrestrial Television Broadcasting (ISDB-T). DVB-H and
ISDB-T support digital transmission of multimedia over a terrestrial
communication network. DVB-H has 3 modes of operation for FFT sizes of
2K, 4K and 8K. ISDB-T has 3 modes of operation for FFT sizes of 256, 512
and 1K. DVB-H is described in document ETSI EN 300 744, entitled "Digital
Video Broadcasting (DVB); Framing structure, channel coding and
modulation for digital terrestrial television," November 2004. ISDB-T is
described in document ARIB STD-B31, entitled "Transmission System for
Digital Terrestrial Television Broadcasting," July 2003. These documents
are publicly available.

[0025] FIG. 1 shows a block diagram of a transmitter 110 and a receiver
150 in an OFDM-based system 100, which may implement DVB-H, ISDB-T,
and/or some other design. At transmitter 110, a transmit (TX) data
processor 120 receives and processes (e.g., formats, encodes,
interleaves, and symbol maps) traffic data to generate data symbols. As
used herein, a data symbol is a modulation symbol for traffic data, a
pilot symbol is a modulation symbol for pilot, which is data that is
known a priori by both the transmitter and receiver, and a zero symbol is
a signal value of zero.

[0026] An OFDM modulator 130 receives and multiplexes the data symbols and
pilot symbols onto data subbands and pilot subbands, respectively. A data
subband is a subband used to send traffic data, and a pilot subband is a
subband used to send pilot. A given subband may serve as a data subband
in one OFDM symbol period and as a pilot subband in another OFDM symbol
period. An OFDM symbol period is the duration of one OFDM symbol and is
also referred to as a symbol period. The pilot symbols may be multiplexed
with the data symbols as described below. OFDM modulator 130 obtains K
transmit symbols for the K total subbands in each OFDM symbol period.
Each transmit symbol may be a data symbol, a pilot symbol, or a zero
symbol. OFDM modulator 130 transforms the K transmit symbols for each
OFDM symbol period with a K-point inverse fast Fourier transform (IFFT)
or inverse discrete Fourier transform (IDFT) to obtain a transformed
symbol that contains K time-domain chips. OFDM modulator 130 then repeats
a portion of the transformed symbol to generate an OFDM symbol. The
repeated portion is often called a cyclic prefix or a guard interval and
is used to combat frequency selective fading, which is a frequency
response that varies across the system bandwidth due to multipath in a
wireless channel. OFDM modulator 130 provides an OFDM symbol for each
OFDM symbol period. A transmitter unit (TMTR) 132 receives and processes
(e.g., converts to analog, amplifies, filters, and frequency upconverts)
the OFDM symbols and generates a modulated signal, which is transmitted
via an antenna 134 to receiver 150.

[0027] At receiver 150, an antenna 152 receives the modulated signal from
transmitter 110 and provides a received signal to a receiver unit (RCVR)
154. Receiver unit 154 conditions (e.g., filters, amplifies, frequency
downconverts, and digitizes) the received signal to obtain input samples.
An OFDM demodulator (Demod) 160 processes the input samples as described
below and obtains K received symbols for the K total subbands in each
OFDM symbol period. The received symbols include received data symbols
for the data subbands and received pilot symbols for the pilot subbands.
OFDM demodulator 160 performs frequency synchronization to estimate and
remove the frequency error at receiver 150. OFDM demodulator 160 also
performs data demodulation/detection on the received data symbols with a
channel estimate to obtain data symbol estimates, which are estimates of
the data symbols sent by transmitter 110. A receive (RX) data processor
170 then processes (e.g., symbol demaps, deinterleaves, and decodes) the
data symbol estimates to obtain decoded data. In general, the processing
by OFDM demodulator 160 and RX data processor 170 is complementary to the
processing by OFDM modulator 130 and TX data processor 120, respectively,
at transmitter 110.

[0028] Controllers/processors 140 and 180 control the operation of various
processing units at transmitter 110 and receiver 150, respectively.
Memories 142 and 182 store data and program codes for transmitter 110 and
receiver 150, respectively.

[0029]FIG. 2 shows an exemplary subband structure 200 for system 100. The
overall system bandwidth of BW MHz is partitioned into multiple (K)
subbands that are given indices of 0 through K-1, where K may be a
configurable value. The spacing between adjacent subbands is BW/K MHz.
For subband structure 200, the K total subbands are arranged into 12
disjoint interlaces. The 12 interlaces are disjoint in that each of the K
subbands belongs in only one interlace. Each interlace contains
approximately K/12 subbands that are uniformly distributed across the K
total subbands such that consecutive subbands in the interlace are spaced
apart by 12 subbands. Thus, interlace u, for uε{0, . . . , 11},
contains subbands u, u+12, u+24, . . . . Index u is the interlace index
as well as a subband offset that indicates the first subband in the
interlace. FIG. 2 only shows four interlaces 0, 3, 6 and 9.

[0030]FIG. 3A shows a pilot structure 300 for DVB-H. Pilot structure 300
includes a continual pilot and a scattered pilot. The continual pilot is
sent on C subbands that are distributed across the system bandwidth,
where C is dependent on the mode. The pilot is continual in that it is
sent on the same C subbands in all OFDM symbol periods. These C subbands
include subbands 0, 48, 54, . . . , K-1 and are given in ETSI EN 300 744.
The scattered pilot is sent on one interlace in each OFDM symbol period.
The transmission timeline for DVB-H is partitioned into frames, with each
frame including 68 OFDM symbols that are given indices of 0 through 67.
The scattered pilot is sent on interlace 0 in OFDM symbol 0, interlace 3
in OFDM symbol 1, interlace 6 in OFDM symbol 2, interlace 9 in OFDM
symbol 4, interlace 0 in OFDM symbol 5, and so on. The scattered pilot is
thus sent on the same four interlaces in each set of 4 OFDM symbols.

[0031]FIG. 3B shows a pilot structure 310 for ISDB-T. Pilot structure 310
includes only a scattered pilot that is sent on interlaces 0, 3, 6 and 9
in each set of 4 OFDM symbols. The transmission timeline for ISDB-T is
also partitioned into frames, with each frame including 204 OFDM symbols
that are given indices of 0 through 203. The scattered pilot is sent on
interlace 0 in OFDM symbol 0 and cycles through interlaces 0, 3, 6 and 9
in the same manner as the scattered pilot for DVB-H.

[0032] For both DVB-H and ISDB-T, the pilot symbols for each OFDM symbol
are generated based on a pseudo-random binary sequence (PBRS) that is
derived from a specific generator polynomial. The PBRS sequence contains
K bits and is given as:

{w}={w0,w1,w.sub.2,w3,w4, . . . ,wK-1}. Eq (1)

PBRS bit Wk, for kε{0, . . . , K-1}, is used to generate a
BPSK modulation symbol that is used as a pilot symbol for subband k. The
pilot symbols for interlace u, for uε{0, 3, 6, 9}, are generated
with PBRS bits {wu, wu+12, wu+24, wu+36, . . . }.

[0033] Table 1 lists the values for some parameters for the three modes in
DVB-H and ISDB-T. In Table 1, parameters N, K, C and S are given for one
OFDM symbol. The number of scattered pilot subbands (S) for both DVB-H
and ISDB-T and the number of continual pilot subbands (C) for DVB-H are
dependent on the mode. For ISDB-T, K is an integer multiple of 12, and
interlaces 0, 3, 6 and 9 contain the same number of pilot subbands. For
DVB-H, K is not an integer multiple of 12, and interlace 0 contains one
more pilot subband than interlaces 3, 6 and 9. For simplicity, the
following description assumes that the interlaces contain the same number
of (S) pilot subbands.

[0036] Δf is the fractional portion of the frequency error, which is
less than one bin;

[0037] fbin is one bin, which is the spacing between adjacent
subbands; and

[0038] m is the integer portion of the frequency error, which is an
integer number of bins.

The integer portion of the frequency error is also called frequency bin
error or coarse bin frequency error.

[0039] A coarse frequency estimator 412 estimates the fractional frequency
error Δf based on the pre-processed samples and in a manner known
in the art. A rotator 414 receives the estimated fractional frequency
error Δ{circumflex over (f)} from estimator 412 and the estimated
frequency bin error {circumflex over (m)} from a frequency bin error
estimator 420, removes the estimated total frequency error from the
pre-processed samples, and provides frequency-corrected samples. A cyclic
prefix removal unit 416 removes the cyclic prefix appended to each OFDM
symbol and provides received samples.

[0040] An FFT/DFT unit 418 performs a fast Fourier transform (FFT) or
discrete Fourier transform (DFT) on the received samples for each OFDM
symbol period and provides frequency-domain received symbols for the K
total subbands. Frequency bin error estimator 420 estimates the frequency
bin error based on the received pilot symbols and provides the estimated
frequency bin error, as described below. Rotator 414 may remove the
estimated frequency bin error from the pre-processed samples, as shown in
FIG. 4. Alternatively, a frequency bin correction unit can remove the
estimated frequency bin error from the received data symbols (not shown
in FIG. 4). A channel estimator 422 derives a channel estimate based on
the received pilot symbols. The channel estimate may be a time-domain
channel impulse response estimate or a frequency-domain channel frequency
response estimate. A data demodulator 424 performs data
demodulation/detection on the received data symbols with the channel
estimate and provides data symbol estimates.

[0041] Although not shown in FIG. 4 for simplicity, OFDM demodulator 160
may include processing units for fine frequency tracking, fine time
tracking, frame synchronization, and/or other functions.

[0042] Frequency bin error estimator 420 estimates the frequency bin error
and further determines the scattered pilot offset, which indicates the
specific interlace used for the scattered pilot in each OFDM symbol
period. The maximum frequency bin error is determined by the accuracy of
the reference oscillator at receiver 150, the center frequency of the
modulated signal being received, and the mode used by the system. For
example, if the reference oscillator has a maximum error of 5 parts per
million (ppm) and the center frequency is 800 MHz, then the maximum
frequency error is ±4 KHz. This ±4 KHz frequency error corresponds
to ±4 bins for a subband spacing of 1116 Hz for mode 3 in ISDB-T and
to ±6 bins for a subband spacing of 697 Hz for mode 3 in DVB-H. For
ISDB-T, there is an ambiguity of ±4 bins. Hence, the correct frequency
bin error is one of 9 "frequency" hypotheses for -4, -3, -2, -1, 0, +1,
+2, +3 and +4 bin errors.

[0043] Receiver 150 typically does not have frame timing when first tuned
to transmitter 110. In this case, for a given OFDM symbol, receiver 150
does not know whether the scattered pilot is being sent on interlace 0,
3, 6 or 9. As shown in FIG. 2, a pilot offset of 0 corresponds to the
scattered pilot being sent on interlace 0, a pilot offset of 1
corresponds to the scattered pilot being sent on interlace 3, a pilot
offset of 2 corresponds to the scattered pilot being sent on interlace 6,
and a pilot offset of 3 corresponds to the scattered pilot being sent on
interlace 9. There is thus an ambiguity of 4 pilot offsets. Hence, the
correct pilot offset is one of 4 "time" hypotheses for pilot offsets of
0, 1, 2 and 3.

[0044] The frequency bin error estimation may be performed in various
manners. In an embodiment, the estimation is performed based on an
assumption that both frequency bin error and pilot offset are unknown.
For this embodiment, multiple hypotheses are formed jointly for frequency
and time. In another embodiment, the estimation is performed in two
steps, with the first step determining the frequency bin error and the
second step determining the pilot offset. For this embodiment, multiple
hypotheses are formed separately for frequency and time. The frequency
bin error estimation may also be performed based on various metrics. In
an embodiment, the estimation is performed based on metrics derived from
despreading the received symbols. In another embodiment, the estimation
is performed based on metrics derived from correlating the received
symbols.

[0045] Table 2 lists four exemplary frequency bin error estimation
schemes, the hypotheses and metrics for each scheme, and the system(s)
for which each scheme is applicable. For clarity, schemes 1 and 4 are
specifically described below.

[0046] For frequency bin error estimation scheme 1 in Table 2, multiple
frequency/time hypotheses are formed for different combinations of
frequency bin error and pilot offset. The total number of frequency/time
hypotheses to evaluate is equal to the product of the number of
hypotheses for frequency bin error (for frequency uncertainty) and the
number of hypotheses for pilot offset (for time uncertainty), which is
9×4=36 frequency/time hypotheses for the example described above
for ISDB-T. One frequency/time hypothesis is the correct hypothesis for
both frequency bin error and pilot offset, and the remaining
frequency/time hypotheses are incorrect.

[0047] The received symbols at receiver 150, without any frequency error,
may be expressed as:

Zk(l)=Hk(l)Sk(l)+Nk(l), Eq (3)

where

[0048] Sk(l) is a modulation symbol sent on subband k in OFDM symbol
period 1;

[0049] Hk(l) is a channel gain for subband k in OFDM symbol period 1;

[0050] Zk(l) is a received symbol for subband k in OFDM symbol period
1; and

[0051] Nk(l) is the noise for subband k in OFDM symbol period 1.

Sk(l) may be a data symbol or a pilot symbol. The pilot symbols are
generated based on the PBRS sequence, and the pilot symbol for subband k
may be given as Sk(l)=(4/3)wk, where 4/3 is a scaling factor
for pilot relative to data.

[0052] If the frequency error is x bins, and assuming that the fraction
frequency error Δf has been removed by rotator 414, then the
received symbols for OFDM symbol periods l and l+1 may be expressed as:

Zk+x(l)=Hk(l)Sk(l)+Nk+x(l), and Eq (4)

Zk+x(l+1)=e.sup.j2πxGHk(l+1)Sk(l+1)+Nk+x(l+1),
Eq (5)

where G is a guard interval ratio. As shown in equations (4) and (5), a
frequency error of x bins results in the modulation symbol sent on
subband k being received on subband k+x at the receiver. The factor
e.sup.j2πxG is due to phase rotation in the received symbols for OFDM
symbol l+1 relative to the phase of the received symbols for OFDM symbol
l with a frequency error of x bins.

[0053] In an embodiment, each frequency/time hypothesis covers a set of
four consecutive OFDM symbols l through l+3. A given frequency/time
hypothesis Hx,y, which corresponds to a hypothesized frequency error
of x bins and a hypothesized pilot offset of y, may be evaluated as
follows. First, the received symbols are extracted from pilot subbands
corresponding to frequency bin error x and pilot offset y. In particular,
received symbols are extracted from interlaces x, x+3, x+6 and x+9 in the
four OFDM symbols for y=0, from interlaces x+3, x+6, x+9 and x in the
four OFDM symbols for y=1, from interlaces x+6, x+9, x and x+3 in the
four OFDM symbols for y=2, and from interlaces x+9, x, x+3 and x+6 in the
four OFDM symbols for y=3. The extracted received symbols for each OFDM
symbol are then despread with the corresponding bits of the PBRS sequence
to obtain despread symbols. The despread symbols for OFDM symbols l+1,
l+2 and l+3 are multiplied with e-j2πxG, e-j4πxG and
e-j6πxG, respectively, to account for the phase rotation across
OFDM symbols due to the frequency error of x bins. The results of the
processing are estimated channel gains (or simply, channel gains) for the
pilot subbands. The channel gains for hypothesis Hx,y for pilot
offsets of y=0, 1, 2 and 3 are given below:

[0054] Each hypothesis Hx,y in equation (6) includes four rows of
channel gains, one row for each OFDM symbol. Each row includes S channel
gains for S pilot subbands in one OFDM symbol. The channel gains are
derived from the received symbols that are extracted from different
subbands depending on the frequency bin error x and the pilot offset y.

[0055] In an embodiment, a metric is derived for frequency/time hypothesis
Hx,y based on a channel impulse response estimate. For this
embodiment, the channel gains from the four OFDM symbols for hypothesis
Hx,y are first sorted based on subband indices. As an example, for
hypothesis Hx,0, the sorted channel gains may be given as:

[0056] An FFT/DFT may then be performed on the 4S sorted channel gains
{Hx,y} to obtain a time-domain channel impulse response estimate
with 4S channel taps, which may be given as:

{hx,y}={hx,y(0),hx,y(1),hx,y(2), . . .
,hx,y(4S-1)}. Eq (8)

Since 4S is not a power of 2, the channel gains {Hx,y} may be
zero-filled to a power two, and an FFT may then be performed on the
zero-filled channel gains.

[0057] In general, channel gains may be obtained for any number of
interlaces and any number of subbands in each interlace. The length of
the channel impulse response estimate is dependent on the number of
channel gains and may be shorter than 4S. As shown in Table 1, DVB-H has
many more scattered pilot subbands than ISDB-T. To reduce computational
complexity, a subset of the scattered pilot subbands may be used to
derive the metric for each hypothesis. For example, the first 16, 32 and
64 pilot subbands in each OFDM symbol may be used for modes 1, 2 and 3,
respectively, in DVB-H. The channel impulse response estimate for each
hypothesis may then be derived using 64-, 128- and 256-point FFTs for
modes 1, 2 and 3, respectively. The FFT size is four times the number of
pilot subbands selected for use.

[0058] If hypothesis Hx,y is a wrong hypothesis, then one or both of
the following apply: [0059] 1. The extracted received symbols are
received data symbols having random complex values. After the PRBS
despreading, the despread symbols remain random complex values. [0060] 2.
The extracted received symbols are received pilot symbols that are
shifted from their correct frequency alignment by a multiple of 3
subbands. When these received pilot symbols are despread with the PRBS
sequence, the resultant despread symbols are random scrambled values. In
either of the two cases above, the despread symbols are noisy and are not
representative of the channel gains. The channel impulse response
estimate derived from these noisy despread symbols would then contain
mostly noise.

[0061] Conversely, if hypothesis Hx,y is the correct hypothesis, then
the extracted received symbols are the received pilot symbols properly
aligned in both time and frequency. When these received pilot symbols are
despread with the PRBS sequence, the resultant despread symbols are good
estimates of the channel gains. A channel impulse response estimate may
then be derived based on these channel gains. This channel impulse
response estimate includes a signal component that is above the noise
floor.

[0062] A metric may be defined based on the channel impulse response
estimate in various manners. In an embodiment, a metric Mx,ya
is set to the energy of the largest tap in the channel impulse response
estimate, which may be expressed as:

M x , y a = max n h x , y ( n ) 2 .
Eq ( 9 ) ##EQU00003##

[0063] In another embodiment, a metric Mx,yb is set to the total
energy of all taps in the channel impulse response estimate, which may be
expressed as:

M x , y b = n = 0 4 S - 1 h x , y ( n )
2 . Eq ( 10 ) ##EQU00004##

[0064] In yet another embodiment, the metric Mx,yc is set to the
energy of large taps in the channel impulse response estimate, which may
be expressed as:

where Eth is a threshold used to determine whether a given tap is
large. Eth may be set to a fixed value or to a predetermined
percentage (e.g., 10%) of the total energy of all taps.

[0065] In yet another embodiment, a metric Mx,yd is set to a
non-coherent sum of metrics obtained for multiple (L) sets of OFDM
symbols, as follows:

M x , y d = i | M x , y ( i ) , Eq
( 12 ) ##EQU00006##

where Mx,y(i) is a metric obtained for OFDM symbol set i.
Mx,y(i) may be obtained based on equation (9), (10) or (11). The L
OFDM symbol sets may be adjacent to one another or spread out over time.

[0066] In general, a metric Mx,y may be derived for hypothesis
Hx,y based on equation (9), (10), (11), (12) or some other equation.
For the embodiments described above, the FFT operation coherently sums
the channel gains {Hx,y} and provides the channel taps {hx,y}.
This coherent sum provides high processing gain and yields good detection
performance even in low SNR conditions. In some other embodiments, metric
Mx,y may be derived based on the channel gains {Hx,y} in other
manners, e.g., by summing the energies of the channel gains.

[0067] In any case, a metric Mx,y is obtained for each frequency/time
hypothesis. The metrics for all frequency/time hypotheses may be
compared, and the hypothesis with the largest metric may be provided as
the correct hypothesis. The frequency bin error for the correct
hypothesis may be provided to rotator 414, as shown in FIG. 4. The pilot
offset for the correct hypothesis may be provided to channel estimator
422 and possibly other processing units within receiver 150.

[0068]FIG. 5 shows a block diagram of a frequency bin error estimator
420a, which is an embodiment of estimator 420 within OFDM demodulator 160
in FIG. 4. Within estimator 420a, a control unit 510 receives inputs
indicative of the range of frequency errors (e.g., ±4 bins) and
whether the pilot offset is known. Control unit 510 forms hypotheses
covering all frequency and/or time uncertainty. A despreading unit 512
obtains received symbols for the K total subbands, extracts the received
symbols from the proper subbands for the hypothesis Hx,y being
evaluated, performs despreading of the extracted received symbols with
the PBRS sequence, rotates the despread symbols for each OFDM symbol by
e-j2πxG to obtain the channel gains {Hx,y}, where v is 0, 1,
2 and 3 for the four OFDM symbols in a set being evaluated.

[0069] Channel estimator 422 receives the channel gains for each
hypothesis Hx,y and derives a channel impulse response estimate
{hx,y} for that hypothesis. A metric computation unit 514 derives a
metric Mx,y for each hypothesis based on the channel impulse
response estimate, e.g., using any of the embodiments described above.
Unit 514 may non-coherently sum multiple metrics obtained for different
OFDM symbol sets as shown in equation (12) or may omit this non-coherent
sum, e.g., for a fast fading channel. A detection unit 516 receives the
metrics for all hypotheses, identifies the largest metric, and provides
the hypothesis with the largest metric as the correct hypothesis.

[0070] For frequency bin error estimation scheme 4 in Table 2, the
frequency bin error may be determined based on the continual pilot that
is sent on the same interlace in all OFDM symbol periods so that there is
no ambiguity as to the pilot subbands. Once the frequency bin error has
been determined, the pilot offset may be ascertained based on the
scattered pilot. By decoupling the frequency bin error and the pilot
offset, the frequency bin error may be determined with 13 frequency
hypotheses for a frequency error range of ±6 bins, and the pilot
offset may be determined with 4 time hypotheses.

[0071] A frequency hypothesis Hx corresponds to a hypothesized
frequency error of x bins. The number of frequency hypotheses to evaluate
is dependent on the frequency error range. Each frequency hypothesis may
be evaluated as follows.

[0072] If hypothesis Hx is correct, then continual pilot symbols are
received on subbands k+x, for kεCP, where CP denotes the set of
continual pilot subbands to be considered. CP may contain all or a subset
of the continual pilot subbands. Equations (4) and (5) may then be
expressed as:

are pilot symbols sent on subband k. Since the same PBRS sequence is used
for all OFDM symbols, the pilot symbols are not a function of OFDM symbol
index l.

[0073] If the wireless channel is relatively static over two consecutive
OFDM symbol periods, then Hk(l+1)≈Hk(l) for all
subbands. In this case, the correlation between two received symbols in
two OFDM symbols l and l+1 for each pilot subband may be expressed as:

Each correlation interval corresponds to a different pair of OFDM
symbols. For example, a first accumulated result may be obtained for OFDM
symbols l and l+1 as shown in equation (16), a second accumulated result
may be obtained for OFDM symbols l+1 and l+2, a third accumulated result
may be obtained for OFDM symbols l+2 and l+3, and a fourth accumulated
result may be obtained for OFDM symbols l+3 and l+4. The four accumulated
results may then be summed to obtain the overall result shown in equation
(17). In general, the correlation results may be accumulated across any
number of subbands and any number of OFDM symbols.

[0074] If hypothesis Hx is not correct because the hypothesized
frequency bin error x is not equal to the actual frequency bin error m,
or x≠m, then received data symbols are extracted from subbands k+x,
for kεCP. Equations (4) and (5) may then be expressed as:

Zk+x(l)=Hk+x-m(l)Dk+x-m(l)+Nk+x-m(l), and Eq (18)

Zk+x(l+1)=e-j2πxGHk+x-m(l+1)Dk+x-m(l+1)+Nk+x-
(l+1), Eq (19)

where Dk+x-m(l) and Dk+x-m(l+1) are data symbols sent on
subband k+x-m in OFDM symbols l and l+1, respectively. The extracted
received symbols may be correlated and accumulated across pilot subbands,
as follows:

Equation (20) indicates that the magnitude squares of the channel gains
do not sum up coherently due to the random nature of the data symbols
Dk+x-m(l) and Dk+x-m(l+1). If the data symbols are
independently and identically distributed (i.i.d.) with zero mean, which
is typically the case, then the accumulated result may be given as:

In equation (22), the correlation results Zk+x(l)Zk+x*(l+1) are
coherently summed over both frequency and time, the accumulated result is
rotated by e-j2πxG, and the real part of the rotated result is
provided as metric Qxa. If hypothesis Hx is correct, then
the rotated result would have a large positive real part, and metric
Qxa is a large value. Conversely, if hypothesis Hx is
incorrect, then the rotated result is a small value, and metric
Qxa is likewise a small value.

[0076] The description above assumes that the wireless channel is
relatively static over the correlation interval. This assumption may not
be true for a fast fading channel, and the correlation between the
received symbols may then be expressed as:

where θk(l) is a random variable for the phase difference in
the wireless channel observed by subband k between OFDM symbol periods l
and l+1. A computer simulation was performed for different channel
realizations and for a number of OFDM symbol periods. For each OFDM
symbol period, the phase difference was determined for each pilot
subband, and the phase differences for all pilot subbands were plotted as
a histogram. This histogram typically has a single nodal peak.

[0077] If θk(l) is centered near 90°, 180° or
270°, then the following metric Qxs provides good
performance:

In equation (24), the correlation results are coherently summed over both
frequency and time, and the squared magnitude of the accumulated result
is provided as metric Qxs.

[0078] In a fast fading channel, the single nodal peak may shift rapidly
from one correlation interval to the next. For example, the peak may be
centered near 0° in one correlation interval and may shift to
180° in the next correlation interval. θk(l) may thus
be nearly out of phase in consecutive correlation intervals. In this
case, a metric Qxf may be defined as:

In equation (25), the correlation results are (1) coherently summed over
frequency to take advantage of the single nodal distribution of
θk(l) for the pilot subbands and (2) non-coherently summed
over time to account for fast and random changes in θk(l).
Metric Qxf may provide better performance for a fast fading
channel.

[0079] In general, metric Qxs is better for static and slow
fading channels, and metric Qxf is better for a fast fading
channel. A metric Qxc may be defined based on both
Qxs and Qxf, as follows:

Qxc=αQxs+(1-α)Qxf, Eq (26)

where α is a weighting factor that determines the weights to be
given to Qxs and Qxf. Qxc is equal to
Qxs for α=1, is equal to Qxf for α=0, and
is equal to a weighted sum of Qxs and Qxf for
0<α<1. Computer simulation shows that α=0.2 provides
good performance for both slow and fast fading channels. α may also
be a configurable value.

[0080] In general, a metric Q may be derived for hypothesis Hx based
on equation (22), (24), (25), (26) or some other equation. Metric Qx
may be computed for each frequency hypothesis, and the metrics for all
hypotheses may be compared. The hypothesis with the largest metric may be
provided as the correct hypothesis, as follows:

m ^ = arg x max { Q x } . Eq ( 27 )
##EQU00017##

[0081] The pilot offset may be determined based on the scattered pilot
once the frequency bin error has been determined based on the continual
pilot. A time hypothesis Hy corresponds to a hypothesized pilot
offset of y, which means that the scattered pilot is being sent on
interlace 3y in OFDM symbol period l. Four time hypotheses are formed for
y=0, 1, 2 and 3, and each hypothesis may be evaluated as follows. For
hypothesis Hy, the scattered pilot is hypothesized to have been sent
on subbands {circumflex over (m)}+3y+12j, for j=0, 1, 2 . . . , in OFDM
symbol periods l and l+4. The correlation between two received symbols in
OFDM symbol periods l and l+4 for each pilot symbol may then be expressed
as:

A metric Qy may be derived for hypothesis Hy by substituting
Z.sub.{circumflex over (m)}+3y+12j(l)Z.sub.{circumflex over
(m)}+3y+12j*(l+4) for Zk+x(l)Zk+x*(l+1) in the equations
described above. Four metrics are obtained for four time hypotheses. The
time hypothesis with the largest metric may be provided as the correct
hypothesis.

[0082]FIG. 6 shows a block diagram of a frequency bin error estimator
420b, which is another embodiment of estimator 420 within OFDM
demodulator 160 in FIG. 4. Within estimator 420b, a control unit 610
receives inputs indicative of the range of frequency errors (e.g., ±4
bins) and whether the pilot offset is known. Control unit 610 forms a set
of frequency hypotheses covering all frequency uncertainty and a set of
time hypotheses covering all time uncertainty. A correlation unit 612
obtains received symbols for the K total subbands, extracts the received
symbols from the proper subbands for hypothesis Hx or Hy being
evaluated, performs correlation on the extracted received symbols, and
provides correlation results for different subbands and correlation
intervals.

[0083] A metric computation unit 614 derives a metric Qx or Qy
for each hypothesis based on the correlation results for that hypothesis,
e.g., using any of the embodiments described above. Unit 614 may
coherently sum the correlation results across subbands and may coherently
or non-coherently sum across correlation intervals. A detection unit 616
receives the metrics for all frequency hypotheses, identifies the largest
metric, and provides the frequency bin error for the frequency hypothesis
with the largest metric as the estimated frequency bin error. Detection
unit 616 also receives the metrics for all time hypotheses, identifies
the largest metric, and provides the pilot offset for the time hypothesis
with the largest metric as the correct pilot offset.

[0084] For scheme 2 in Table 2, hypotheses are formed jointly for
frequency bin error and pilot offset, and each hypothesis is evaluated
using correlation-based metrics, e.g., the metrics shown in equations
(22), (24), (25) and/or (26). For scheme 3 in Table 2, hypotheses are
formed separately for frequency bin error and pilot offset, and each
hypothesis is evaluated using despreading-based metrics, e.g., the
metrics shown in equations (9), (10), (11) and/or (12). A scheme may also
use a combination of despreading-based metric and correlation-based
metric. For example, a despreading-based metric may be used for frequency
hypotheses, and a correlation-based metric may be used for time
hypotheses. Other metrics defined in other manners may also be used to
evaluate the hypotheses.

[0085] FIG. 7 shows an embodiment of a process 700 for performing
frequency error estimation by despreading the received symbols.
Time-domain input samples are processed to obtain frequency-domain
received symbols for the K total subbands (block 710). Multiple
hypotheses are formed for different frequency bin errors (or bin
offsets), different pilot offsets, or different combinations of frequency
bin error and pilot offset (block 712). For each hypothesis, received
symbols are extracted from the proper subbands determined by the
hypothesis (block 714). The extracted received symbols are hypothesized
to be for (1) a scattered pilot that is sent on different sets of
subbands in different symbol periods and/or (2) a continual pilot that is
sent on the same set of subbands in all symbol periods. The extracted
received symbols for each hypothesis are despread with a scrambling
sequence, e.g., the PBRS sequence, to obtain despread symbols for that
hypothesis (block 716). A metric is then derived for each hypothesis
based on the despread symbols for that hypothesis (block 718). For block
718, a channel impulse response estimate may be derived for each
hypothesis based on the despread symbols for the hypothesis. The metric
for each hypothesis may then be derived based on the channel impulse
response estimate for the hypothesis, as described above. In any case,
the frequency bin error and/or the pilot offset are determined based on
the metrics for all hypotheses evaluated (block 720).

[0086]FIG. 8 shows an embodiment of a process 800 for performing
frequency error estimation by correlating the received symbols.
Time-domain input samples are processed to obtain frequency-domain
received symbols for the K total subbands (block 810). Multiple
hypotheses are formed for different frequency bin errors, different pilot
offsets, or different combinations of frequency bin error and pilot
offset (block 812). For each hypothesis, received symbols in multiple
symbol periods are extracted from the proper subbands determined by the
hypothesis (block 814). The extracted received symbols are hypothesized
to be for a scattered pilot and/or a continual pilot. For each
hypothesis, correlation is performed on the extracted received symbols
for each subband to obtain correlation results for that hypothesis (block
816). A metric is then derived for each hypothesis based on the
correlation results for all subbands and correlation intervals for that
hypothesis (block 818). For example, the metric for each hypothesis may
be derived by coherently summing the correlation results across subbands
and coherently or non-coherently summing the correlation results across
correlation intervals. The metric may also be derived based on a weighted
sum of metrics obtained with different accumulation schemes, e.g., as
shown in equation (26). In any case, the frequency bin error and/or the
pilot phase is determined based on the metrics for all hypotheses
evaluated (block 820).

[0087]FIG. 9 shows an embodiment of a process 900 for performing
frequency error estimation in multiple stages. A frequency error is
determined based on a first pilot (e.g., a continual pilot) by evaluating
a first set of hypotheses for a range of frequency errors (block 912). A
pilot offset is determined based on a second pilot (e.g., a scattered
pilot) by evaluating a second set of hypotheses for a set of pilot
offsets and with the frequency error determined from the first pilot
(block 914). The two sets of hypotheses may be evaluated using the same
or different metrics.

[0089] The techniques described herein may be implemented by various
means. For example, these techniques may be implemented in hardware,
firmware, software, or a combination thereof. For a hardware
implementation, the processing units used to perform frequency error
estimation may be implemented within one or more application specific
integrated circuits (ASICs), digital signal processors (DSPs), digital
signal processing devices (DSPDs), programmable logic devices (PLDs),
field programmable gate arrays (FPGAs), processors, controllers,
micro-controllers, microprocessors, electronic devices, other electronic
units designed to perform the functions described herein, or a
combination thereof.

[0090] For a firmware and/or software implementation, the techniques may
be implemented with modules (e.g., procedures, functions, and so on) that
perform the functions described herein. The software codes may be stored
in a memory (e.g., memory 182 in FIG. 1) and executed by a processor
(e.g., processor 180). The memory may be implemented within the processor
or external to the processor.

[0091] The previous description of the disclosed embodiments is provided
to enable any person skilled in the art to make or use the present
invention. Various modifications to these embodiments will be readily
apparent to those skilled in the art, and the generic principles defined
herein may be applied to other embodiments without departing from the
spirit or scope of the invention. Thus, the present invention is not
intended to be limited to the embodiments shown herein but is to be
accorded the widest scope consistent with the principles and novel
features disclosed herein.

Patent applications by Shimman Patel, San Diego, CA US

Patent applications in class Particular pulse demodulator or detector

Patent applications in all subclasses Particular pulse demodulator or detector