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To My Wife

PARKASH

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Preface
Out of the various television systems in use in different countries, India adopted the 625-B monochrome (black and white) and the compatible PAL-B colour systems. Most European and many other countries are also using these standards. However, majority of the available books on television deal with the American 525 line monochrome and NTSC colour systems. Moreover, in the latest editions of the popular books on this subject, emphasis has totally shifted to colour television because of its wide acceptance in all the advanced countries. Since colour television has just been introduced in India, both monochrome and colour transmissions will co-exist for a long time in this country. Keeping these facts in view, this text has been prepared to provide an integrated approach with equal emphasis on both monochrome and colour systems. The book has been designed to meet the requirements of a modern text book on ‘Television Engineering’ for Electrical and Electronics Engineering students at the degree level. It will also meet the needs of a comprehensive course on TV Engineering in Polytechnics and Technical Schools. In addition the book will be of immense value to practising engineers and technicians. Students engaged in self study will also benefit very much from this text. The matter has been so presented that any Engineering student with a basic knowledge of the various electronic building blocks and fundamentals of communication systems will have no difficulty in understanding the subject. Comprehensive design criteria for various sections of the receiver have been given in each chapter without going into rigorous mathematical details. Due emphasis has also been laid on TV receiver servicing and servicing equipment. Detailed charts for locating faults and trouble shootting together with alignment procedures for the various sections of the receiver have also been included. Early TV receivers manufactured in India and other countries used vacuum tube circuitry. However, with rapid advances in technology, hybrid circuitry soon came into use and transistors replaced most vacuum tubes. With the widespread development of integrated circuits, special ICs are now available and are fast replacing discrete circuitry employing transistors. Since these developments have been very fast, sets employing tubes, only transistors and ICs are in use simultaneously. In view of this fact discussion of TV circuits using tubes, hybrid circuitry and ICs has been included in the chapter devoted to receiver circuits. The stress, however, is more on solid state receiver circuits and design. Because of the importance of colour transmission and reception, two comprehensive chapters have been exclusively devoted to the techniques of colour television and various colour television systems. All modern colour receivers use solid-state devices in most sections of the receiver. It is natural that receivers manufactured in India will also be of this type with specially

designed ICs for performing almost all the functions in the receiver. Therefore, the description of colour receivers using vacuum tubes has been totally omitted. However, the functioning and use of special instruments necessary for the manufacture, testing and servicing all types of colour receivers has been included in corresponding chapters. Chapter 1 gives basic principles of TV transmission and reception. Chapters 2, 3 and 4 deal with analysis and synthesis of TV pictures, composite video signal and channel bandwidth requirements. Chapters 5 and 6 include discussion on receiver picture tubes and television camera tubes. Chapter 7 is on TV studio equipment and transmission principles. Chapter 8 gives a block schematic approach to monochrome TV receivers. Chapter 9 explains the propagation phenomena and antenna systems with special reference to TV transmission and reception. Chapter 10 is devoted to various applications of television. In Chapters 11 to 24 detailed circuit analysis and design principles of the various sections of the receiver are given. In Chapters 25 and 26 fundamentals of colour television and various colour TV systems have been fully described, naturally with a greater emphasis on the PAL-B & G systems. Chapter 27 is exclusively devoted to special circuits like remote control tuning, automatic fine tuning etc. Chapter 28 deals with all types of equipment needed for testing, alignment and servicing monochrome and colour receivers. Chapter 29 discusses in detail procedures for alignment of various sections of the receiver. Comprehensive details for trouble shooting and servicing are presented in Chapter 30. Diagnostic test charts are also included. The manuscript and its various pre-drafts have been used successfully by the author for a selective one semester course on TV Engineering at the final degree level at BITS, Pilani. About twice the material necessary for a one semester course is included in the book. By a judicious choice of chapters and sequencing the instructor can prepare suitable presentations for a two semester sequence for both degree and diploma students emphasising basic principles, overall systems or technological details. To assist the student and the instructor a set of review questions are included at the end of each chapter. R.R. Gulati

Suggested Further Reading
This book was published in 1983 and ever since its popularity has been growing. Its relevance as a comprehensive course in Television Engineering lies in its excellent presentation of the fundamentals of television transmission and reception. In it, analysis and synthesis of TV pictures, generation of composite video and audio signals, channel bandwidth requirements and design factors for various sections of the receiver have been evolved from first principles and supported with mathematical derivations where necessary. However, the author wishes to point out that during the past two decades or so, television receiver designs have gradually changed because of rapid technological advances in the field of entertainment electronics. Therefore, the students are advised to reinforce their learning by further reading books which describe latest techniques and circuits of modern television receivers. This author has contributed in this direction by writing the following three books now published by NEW AGE INTERNATIONAL PUBLISHERS. 1. COLOUR TELEVISION–PRINCIPLES AND PRACTICE. This book describes colour TV principles in depth and gives detailed insight of colour TV systems and standards, frequency synthesized tuning and channel selection, chroma processing sub-systems and matrixing, modern receiver circuits employing latest ICs and also colour receiver alignment and servicing. 2. MODERN TELEVISION PRACTICE–PRINCIPLES, TECHNOLOGY AND SERVICING. The main feature of this book is the side by side coverage of B&W and colour TV transmission and reception techniques for a better grasp of the entire field of television engineering. The 2nd edition of this book published in 2002 also contains chapters on Satellite Television Technology, Cable Television, VCR and Video Disc Recording and Playback, Teletext Broadcast Service and TV Games, Digital Television and Advanced Television Systems. 3. COMPOSITE SATELLITE AND CABLE TELEVISION. This book presents basics and systematic exposition to various equipments, devices, and circuit formulations involved in Satellite and Cable Television. Its 2nd edition contains extended coverage on Signal Encoding & Compression Techniques, Digital Satellite Transmission and Reception, Conditional Access (CAS) System, Direct-to-Home Satellite Broadcasts, High Definition TV (HDTV) and TV Home Entertainment Theatres. R. R. Gulati

Introduction
Development of Television Television* means ‘to see from a distance’. The desire in man to do so has been there for ages. In the early years of the twentieth century many scientists experimented with the idea of using selenium photosensitive cells for converting light from pictures into electrical signals and transmitting them through wires. The first demonstration of actual television was given by J.L. Baird in UK and C.F. Jenkins in USA around 1927 by using the technique of mechanical scanning employing rotating discs. However, the real breakthrough occurred with the invention of the cathode ray tube and the success of V.K. Zworykin of the USA in perfecting the first camera tube (the iconoscope) based on the storage principle. By 1930 electromagnetic scanning of both camera and picture tubes and other ancillary circuits such as for beam deflection, video amplification, etc. were developed. Though television broadcast started in 1935, world political developments and the second world war slowed down the progress of television. With the end of the war, television rapidly grew into a popular medium for dispersion of news and mass entertainment. Television Systems At the outset, in the absence of any international standards, three monochrome (i.e. black and white) systems grew independently. These are the 525 line American, the 625 line European and the 819 line French systems. This naturally prevents direct exchange of programme between countries using different television standards. Later, efforts by the all world committee on radio and television (CCIR) for changing to a common 625 line system by all concerned proved ineffective and thus all the three systems have apparently come to stay. The inability to change over to a common system is mainly due to the high cost of replacing both the transmitting equipment and the millions of receivers already in use. However the UK, where initially a 415 line monochrome system was in use, has changed to the 625 line system with some modification in the channel bandwidth. In India, where television transmission started in 1959, the 625-B monochrome system has been adopted. The three different standards of black and white television have resulted in the development of three different systems of colour television, respectively compatible with the three monochrome systems. The original colour system was that adopted by the USA in 1953 on the recommendations of its National Television Systems Committee and hence called the NTSC system. The other two colour systems–PAL and SECAM are later modifications of the NTSC system, with minor improvements, to conform to the other two monochrome standards.
*From the Greek tele (= far) and the Latin visionis (from videre = to see).

2

INTRODUCTION

3

Regular colour transmission started in the USA in 1954. In 1960, Japan adopted the NTSC system, followed by Canada and several other countries. The PAL colour system which is compatible with the 625 line monochrome European system, and is a variant of the NTSC system, was developed at the Telefunken Laboratories in the Federal Republic of Germany (FRG). This system incorporates certain features that tend to reduce colour display errors that occur in the NTSC system during transmission. The PAL system was adopted by FRG and UK in 1967. Subsequently Australia, Spain, Iran and several other countries in West and South Asia have opted for the PAL system. Since this system is compatible with the 625-B monochrome system, India also decided to adopt the PAL system. The third colour TV system in use is the SECAM system. This was initially developed and adopted in France in 1967. Later versions, known as SECAM IV and SECAM V were developed at the Russian National Institute of Research (NIR) and are sometimes referred to as the NIR-SECAM systems. This system has been adopted by the USSR, German Democratic Republic, Hungary, some other East European countries and Algeria. When both the quality of reproduction and the cost of equipment are taken into account, it is difficult to definitely establish the superiority of any one of these systems over the other two. All three systems have found acceptance in their respective countries. The deciding factor for adoption was compatibility with the already existing monochrome system. Applications of Television Impact of television is far and wide, and has opened new avenues in diverse fields like public entertainment, social education, mass communication, newscasts, weather reports, political organization and campaigns, announcements and guidance at public places like airport terminals, sales promotion and many others. Though the capital cost and operational expenses in the production and broadcasting of TV programmes are high compared to other media, its importance for mass communication and propagation of social objectives like education are well recognized and TV broadcasts are widely used for such purposes. Closed Circuit Television (CCTV) is a special application in which the camera signals are made available over cable circuits only to specified destinations. This has important applications where viewers need to see an area to which they may not go for reasons of safety or convenience. Group demonstrations of surgical operations or scientific experiments, inspection of noxious or dangerous industrial or scientific processes (e.g. nuclear fuel processing) or of underwater operations and surveillance of areas for security purposes are some typical examples. A special type of CCTV is what might be called wired community TV. Small communities that fall in the ‘shadow’ of tall geographical features like hills can jointly put up an antenna at a suitable altitude and distribute the programme to the subscribers’ premises through cable circuits. Another potential use of CCTV that can become popular and is already technically feasible is a video-telephone or ‘visiphone’. Equipment Television broadcasting requires a collection of sophisticated equipment, instruments and components that require well trained personnel. Television studios employ extensive lighting facilities, cameras, microphones, and control equipment. Transmitting equipment for

4

MONOCHROME AND COLOUR TELEVISION

modulation, amplification and radiation of the signals at the high frequencies used for television broadcast are complex and expensive. A wide variety of support equipment essential in broadcast studios, control rooms and outside includes video tape recorders, telecine machines, special effects equipment plus all the apparatus for high quality sound broadcast. Coverage Most programmes are produced live in the studio but recorded on video tape at a convenient time to be broadcast later. Of course, provision for live broadcast also has to be there for VIP interviews, sports events and the like. For remote pick-ups the signal is relayed by cable or RF link to the studio for broadcasting in the assigned channel. Each television broadcast station is assigned a channel bandwidth of 7 MHz (6 MHz in the American, 8 MHz in the British and 14 MHz in the French systems). In the earlier days TV broadcast was confined to assigned VHF bands of 41 to 68 MHz and 174 to 230 MHz. Later additional channel allocations have been made in the UHF band between 470 and 890 MHz. Because of the use of VHF-UHF frequencies for television broadcast, reception of TV signals is limited to roughly the line of sight distance. This usually varies between 75 and 140 km depending on the topography and radiated power. Area of TV broadcast coverage can be extended by means of relay stations that rebroadcast signals received via microwave links or coaxial cables. A matrix of such relay stations can be used to provide complete national coverage. With the rapid strides made in the technology of space and satellite communication it has now become possible to have global coverage by linking national TV systems through satellites. Besides their use for international TV networks, large countries can use satellites for distributing national programmes over the whole area. One method for such national coverage is to set up a network of sensitive ground stations for receiving signals relayed by a satellite and retelecasting them to the surrounding area. Another method is to employ somewhat higher transmitter power on the satellite and receive the down transmissions directly through larger dish antenna on conventional television receivers fitted with an extra front-end converter. A combination of both the methods was successfully tested in India where NASA’s ATS-6 satellite was used for the SITE programme trials in 1975-76. This resulted in the launching of INSAT 1-A in April 1982. Recent Trends In the last decade, transistors and integrated circuits have greatly improved the quality of performance of TV broadcasting and reception. Modern camera tubes like vidicon and plumbicon have made TV broadcast of even dimly lit scenes possible. Special camera tubes are now used for different specific applications. The most sensitive camera tubes available today can produce usable signals even from the scenes where the human eye sees total darkness. With rapid advances in solid state technology, rugged solid state image scanners may conceivably replace the fragile camera tubes in the not-too-distant future. Experimental solid state cameras are already in use for some special applications. Solid state ‘picture-plates’ for use in receivers are under active development. Before long the highly vulnerable high vacuum glass envelope of the picture tube may be a thing of the past. Since solid state charge coupled devices are scanned by digital addressing, the camera scanner and picture plate can work in exact synchronism with no non-linear distortions of the reconstructed picture. An important recent technological advance is the use of pseudo-random scan. The signal so generated requires much less bandwidth than the one for conventional method of scanning.

INTRODUCTION

5

Besides all this, wider use of composite devices, made by integrated solid state technology, for television studio and transmitter equipment as well as for receivers will result in higher quality of reproduction, lower costs and power consumptions with increased reliability and compactness. Special mention may be made of the surface acoustic wave filter to replace the clumsy and expensive IF transformer. Further, large screen TV reception systems based on projection techniques now under development will make it possible to show TV programmes to large audience as in a theatre. With the rapid development of large scale integrated (LSI) electronics in the last decade, digital communication by pulse code modulation (PCM) has made immense progress. The advantage gained is, that virtual freedom from all noise and interference is obtained by using a somewhat larger bandwidth and a specially coded signal. Even if the final transmission in TV is retained in its present form, so that all previous receivers remain usable, the processing of pictures from the camera to the transmitter input is likely to change over to PCM techniques. Unlike the case of monochrome TV standards, the International Telecommunication Union (ITU), a UN special agency, has already adopted a single set of standards accepted by all member countries for the production and processing of picture signals by digital methods. Digital TV has become all the more attractive since solid state cameras compatible with digital signal processing and deflection circuitry have also been developed and are at present in the field testing stage.

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1
Elements of a Television System

1
Elements of a Television System
The fundamental aim of a television system is to extend the sense of sight beyond its natural limits, along with the sound associated with the scene being televised. Essentially then, a TV system is an extension of the science of radio communication with the additional complexity that besides sound the picture details are also to be transmitted. In most television systems, as also in the C.C.I.R. 625 line monochrome system adopted by India, the picture signal is amplitude modulated and sound signal frequency modulated before transmission. The carrier frequencies are suitably spaced and the modulated outputs radiated through a common antenna. Thus each broadcasting station can have its own carrier frequency and the receiver can then be tuned to select any desired station. Figure 1.1 shows a simplified block representation of a TV transmitter and receiver.

1.1

PICTURE TRANSMISSION

The picture information is optical in character and may be thought of as an assemblage of a large number of bright and dark areas representing picture details. These elementary areas into which the picture details may be broken up are known as ‘picture elements’, which when viewed together, represent the visual information of the scene. Thus the problem of picture transimission is fundamentally much more complex, because, at any instant there are almost an infinite number of pieces of information, existing simultaneously, each representing the level of brightness of the scene to the reproduced. In other words the information is a function of two variables, time and space. Ideally then, it would need an infinite number of channels to transmit optical information corresponding to all the picture elements simultaneously. Presently the practical difficulties of transmitting all the information simultaneously and decoding it at the receiving end seem insurmountable and so a method known as scanning is used instead. Here the conversion of optical information to electrical form and its transmission are carried out element by element, one at a time and in a sequential manner to cover the entire scene which is to be televised. Scanning of the elements is done at a very fast rate and this process is repeated a large number of times per second to create an illusion of simultaneous pick-up and transmission of picture details. A TV camera, the heart of which is a camera tube, is used to convert the optical information into a corresponding electrical signal, the amplitude of which varies in accordance with the variations of brightness. Fig. 1.2 (a) shows very elementary details of one type of camera tube (vidicon) to illustrate this principle. An optical image of the scene to be transmitted is focused by a lens assembly on the rectangular glass face-plate of the camera tube. The inner 8

side of the glass face-plate has a transparent conductive coating on which is laid a very thin layer of photoconductive material. The photolayer has a very high resistance when no light falls on it, but decreases depending on the intensity of light falling on it. Thus depending on the light intensity variations in the focused optical image, the conductivity of each element of the photolayer changes accordingly. An electron beam is used to pick-up the picture information now available on the target plate in terms of varying resistance at each point. The beam is formed by an electron gun in the TV camera tube. On its way to the inner side of the glass faceplate it is deflected by a pair of deflecting coils mounted on the glass envelope and kept mutually perpendicular to each other to achieve scanning of the entire target area. Scanning is done in the same way as one reads a written page to cover all the words in one line and all the lines on the page (see Fig. 1.2 (b)). To achieve this the deflecting coils are fed separately from two sweep oscillators which continuously generate saw-tooth waveforms, each operating at a different desired frequency. The magnetic deflection caused by the current in one coil gives horizontal motion to the beam from left to right at a uniform rate and then brings it quickly to

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MONOCHROME AND COLOUR TELEVISION

the left side to commence the trace of next line. The other coil is used to deflect the beam from top to bottom at a uniform rate and for its quick retrace back to the top of the plate to start this process all over again. Two simultaneous motions are thus given to the beam, one from left to right across the target plate and the other from top to bottom thereby covering the entire area on which the electrical image of the picture is available. As the beam moves from element to element, it encounters a different resistance across the target-plate, depending on the resistance of the photoconductive coating. The result is a flow of current which varies in magnitude as the elements are scanned. This current passes through a load resistance RL, connected to the conductive coating on one side and to a dc supply source on the other. Depending on the magnitude of the current a varying voltage appears across the resistance RL and this corresponds to the optical information of the picture.
Glass plate Focusing lens Photoconductive surface Light Object to be televised Video signal output RL Power supply Electron beam Electron gun Cathode Conductive coating

If the scanning beam moves at such a rate that any portion of the scene content does not have time to move perceptibly in the time required for one complete scan of the image, the resultant electrical signal contains the true information existing in the picture during the time of the scan. The desired information is now in the form of a signal varying with time and scanning may thus be identified as a particular process which permits the conversion of information existing in space and time coordinates into time variations only. The electrical information obtained from the TV camera tube is generally referred to as video signal (video is Latin for ‘see’). This signal is amplified and then amplitude modulated with the channel picture carrier frequency. The modulated output is fed to the transmitter antenna for radiation along with the sound signal.

1.2

SOUND TRANSMISSION

The microphone converts the sound associated with the picture being televised into proportionate electrical signal, which is normally a voltage. This electrical output, regardless of the complexity of its waveform, is a single valued function of time and so needs a single channel for its transmission. The audio signal from the microphone after amplification is frequency modulated, employing the assigned carrier frequency. In FM, the amplitude of the carrier signal is held constant, whereas its frequency is varied in accordance with amplitude variations of the modulating signal. As shown in Fig. 1.1 (a), output of the sound FM transmitter is finally combined with the AM picture transmitter output, through a combining network, and fed to a common antenna for radiation of energy in the form of electromagnetic waves.

1.3

PICTURE RECEPTION

The receiving antenna intercepts the radiated picture and sound carrier signals and feeds them to the RF tuner (see Fig. 1.1 (b)). The receiver is of the heterodyne type and employs two or three stages of intermediate frequency (IF) amplification. The output from the last IF stage
Control grid Cathode Heater Accelerating anode Focusing anode Final anode

Screen Electron beam

Electron gun Base Deflection coils EHT

Phosphor coating

Fig. 1.3 Elements of a picture tube.

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MONOCHROME AND COLOUR TELEVISION

is demodulated to recover the video signal. This signal that carries the picture information is amplified and coupled to the picture tube which converts the electrical signal back into picture elements of the same degree of black and white. The picture tube shown in Fig. 1.3 is very similar to the cathode-ray tube used in an oscilloscope. The glass envelope contains an electrongun structure that produces a beam of electrons aimed at the fluorescent screen. When the electron beam strikes the screen, light is emitted. The beam is deflected by a pair of deflecting coils mounted on the neck of the picture tube in the same way and rate as the beam scans the target in the camera tube. The amplitudes of the currents in the horizontal and vertical deflecting coils are so adjusted that the entire screen, called raster, gets illuminated because of the fast rate of scanning. The video signal is fed to the grid or cathode of the picture tube. When the varying signal voltage makes the control grid less negative, the beam current is increased, making the spot of light on the screen brighter. More negative grid voltage reduces the brightness. if the grid voltages is negative enough to cut-off the electron beam current at the picture tube there will be no light. This state corresponds to black. Thus the video signal illuminates the fluorescent screen from white to black through various shades of grey depending on its amplitude at any instant. This corresponds to the brightness changes encountered by the electron beam of the camera tube while scanning the picture details element by element. The rate at which the spot of light moves is so fast that the eye is unable to follow it and so a complete picture is seen because of the storage capability of the human eye.

1.4

SOUND RECEPTION

The path of the sound signal is common with the picture signal from antenna to the video detector section of the receiver. Here the two signals are separated and fed to their respective channels. The frequency modulated audio signal is demodulated after at least one stage of amplification. The audio output from the FM detector is given due amplification before feeding it to the loudspeaker.

1.5

SYNCHRONIZATION

It is essential that the same coordinates be scanned at any instant both at the camera tube target plate and at the raster of the picture tube, otherwise, the picture details would split and get distorted. To ensure perfect synchronization between the scene being televised and the picture produced on the raster, synchronizing pulses are transmitted during the retrace, i.e., fly-back intervals of horizontal and vertical motions of the camera scanning beam. Thus, in addition to carrying picture detail, the radiated signal at the transmitter also contains synchronizing pulses. These pulses which are distinct for horizontal and vertical motion control, are processed at the receiver and fed to the picture tube sweep circuitry thus ensuring that the receiver picture tube beam is in step with the transmitter camera tube beam.

ELEMENTS OF A TELEVISION SYSTEM

13

1.6

RECEIVER CONTROLS

The front view of a typical monochrome TV receiver, having various controls is shown in Fig. 1.4. The channel selector switch is used for selecting the desired channel. The fine tuning control is provided for obtaining best picture details in the selected channel. The hold control is used to get a steady picture in case it rolls up or down. The brightness control varies the beam intensity of the picture tube and is set for optimum average brightness of the picture. The contrast control is actually the gain control of the video amplifier. This can be varied to obtain the desired contrast between the white and black contents of the reproduced picture. The volume and tone controls form part of the audio amplifier in the sound section, and are used for setting the volume and tonal quality of the sound output from the loudspeaker.

Vertical hold Channel selector Fine tuning Contrast

Tone

Brightness Volume and On - off

Fig. 1.4 Television receiver controls

1.7

COLOUR TELEVISION

Colour television is based on the theory of additive colour mixing, where all colours including white can be created by mixing red, green, and blue lights. The colour camera provides video signals for the red, green, and blue information. These are combined and transmitted along with the brightness (monochrome) signal. Each colour TV system* is compatible with the corresponding monochrome system. Compatibility means that colour broadcasts can be received as black and white on monochrome receivers. Conversely colour receivers are able to receive black and white TV broadcasts. This is illustrated in Fig. 1.5 where the transmission paths from the colour and monochrome cameras are shown to both colour and monochrome receivers. At the receiver, the three colour signals are separated and fed to the three electron guns of colour picture tube. The screen of the picture tube has red, green, and blue phosphors arranged in alternate dots. Each gun produces an electron beam to illuminate the three colour phosphors separately on the fluorescent screen. The eye then integrates the red, green and blue colour information and their luminance to perceive the actual colour and brightness of the picture being televised.
* The three compatible colour television systems are NTSC, PAL and SECAM.

14 Colour Receiver Controls

MONOCHROME AND COLOUR TELEVISION

NTSC colour television receivers have two additional controls, known as Colour and Hue controls. These are provided at the front panel along with other controls. The colour or saturation control varies the intensity or amount of colour in the reproduced picture. For example, this control determines whether the leaves of a tree in the picture are dark green or light green, and whether the sky in the picture is dark blue or light blue. The tint or hue control selects the correct colour to be displayed. This is primarily used to set the correct skin colour, since when flesh tones are correct, all other colours are correctly reproduced. It may be noted that PAL colour receivers do not need any tint control while in SECAM colour receivers, both tint and saturation controls are not necessary. The reasons for such differences are explained in chapters exclusively devoted to colour television.
Optical filters R Object G B Combining matrix vR R1 vG R2 vB R3 R v Transmission path Red, green and blue guns R Colour G receiver B Colour picture tube Red, green and blue phosphors

Review Questions
1. Why is scanning necessary in TV transmission ? Why is it carried out at a fast rate ? 2. What is the basic principle of operation of a television camera tube ? 3. What is a raster and how is it produced on the picture tube screen ? 4. Why are synchronizing pulses transmitted along with the picture signal ? 5. Why is FM preferred to AM for sound signal transmission ? 6. Describe briefly the functions of various controls provided on the front panel of a TV receiver. 7. Describe the basic principle of colour television transmission and reception. 8. Describe the function of saturation and hue controls in a NTSC colour TV receiver.

2
Analysis and Synthesis of Television Pictures

2
Analysis and Synthesis of Television Pictures
The basic factors with which the television system must deal for successful transmission and reception of pictures are: (a) Gross Structure: Geometric form and aspect ratio of the picture. (b) Image Continuity: Scanning and its sequence. (c) Number of Scanning Lines: Resolution of picture details. (d) Flicker: Interlaced scanning. (e) Fine Structure: Vertical and horizontal resolution. (f) Tonal Gradation: Picture brightness transfer characteristics of the system.

2.1

GROSS STRUCTURE

The frame adopted in all television systems is rectangular with width/height ratio, i.e., aspect ratio = 4/3. There are many reasons for this choice. In human affairs most of the motion occurs in the horizontal plane and so a larger width is desirable. The eyes can view with more ease and comfort when the width of a picture is more than its height. The usage of rectangular frame in motion pictures with a width/height ratio of 4/3 is another important reason for adopting this shape and aspect ratio. This enables direct television transmission of film programmes without wastage of any film area. It is not necessary that the size of the picture produced on the receiver screen be same as that being televised but it is essential that the aspect ratio of the two be same, otherwise the scene details would look too thin or too wide. This is achieved by setting the magnitudes of the current in the deflection coils to correct values, both at the TV camera and receiving picture tube. Another important requirement is that the same coordinates should be scanned at any instant both by the camera tube beam and the picture tube beam in the receiver. Synchronizing pulses are transmitted along with the picture information to achieve exact congruence between transmitter and receiver scanning systems.

2.2

IMAGE CONTINUITY

While televising picture elements of the frame by means of the scanning process, it is necessary to present the picture to the eye in such a way that an illusion of continuity is created and any 16

ANALYSIS AND SYNTHESIS OF TELEVISION PICTURES

17

motion in the scene appears on the picture tube screen as a smooth and continuous change. To achieve this, advantage is taken of ‘persistence of vision’ or storage characteristics of the human eye. This arises from the fact that the sensation produced when nerves of the eye’s retina are stimulated by incident light does not cease immediately after the light is removed but persists for about 1/16th of a second. Thus if the scanning rate per second is made greater than sixteen, or the number of pictures shown per second is more than sixteen, the eye is able to integrate the changing levels of brightness in the scene. So when the picture elements are scanned rapidly enough, they appear to the eye as a complete picture unit, with none of the individual elements visible separately. In present day motion pictures twenty-four still pictures of the scene are taken per second and later projected on the screen at the same rate. Each picture or frame is projected individually as a still picture, but they are shown one after the other in rapid succession to produce the illusion of continuous motion of the scene being shown. A shutter in the projector rotates in front of the light source and allows the film to be projected on the screen when the film frame is still, but blanks out any light from the screen during the time when the next film frame is being moved into position. As a result, a rapid succession of still-film frames is seen on the screen. With all light removed during the change from one frame to the next, the eye sees a rapid sequence of still pictures that provides the illusion of continuous motion. Scanning. A similar process is carried out in the television system. The scene is scanned rapidly both in the horizontal and vertical directions simultaneously to provide sufficient number of complete pictures or frames per second to give the illusion of continuous motion. Instead of the 24 as in commercial motion picture practice, the frame repetition rate is 25 per second in most television systems. Horizontal scanning. Fig. 2.1 (a) shows the trace and retrace of several horizontal lines. The linear rise of current in the horizontal deflection coils (Fig. 2.1 (b)) deflects the beam across the screen with a continuous, uniform motion for the trace from left to right. At the peak of the rise, the sawtooth wave reverses direction and decreases rapidly to its initial value. This fast reversal produces the retrace or flyback. The start of the horizontal trace is at the left
W Start of a line

End of a line

Trace

Retrace

H

Raster

Fig. 2.1 (a) Path of scanning beam in covering picture area (Raster).

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MONOCHROME AND COLOUR TELEVISION

edge of raster. The finish is at the right edge, where the flyback produces retrace back to the left edge.
i(H) i(H) max

Trace

Retrace

Raster of 625 lines Right t Trace Retrace

1st line

2nd line

3rd line Left

Trace period Retrace period One cycle of deflection current

Fig. 2.1 (b) Waveform of current in the horizontal deflection coils producing linear (constant velocity) scanning in the horizontal direction.

Note, that ‘up’ on the sawtooth wave corresponds to horizontal deflection to the right. The heavy lines in Fig. 2.1 (a) indicate the useful scanning time and the dashed lines correspond to the retrace time. Vertical scanning. The sawtooth current in the vertical deflection coils (see Fig. 2.2) moves the electron beam from top to bottom of the raster at a uniform speed while the electron beam is being deflected horizontally. Thus the beem produces complete horizontal lines one below the other while moving from top to bottom. As shown in Fig. 2.2 (c), the trace part of the sawtooth wave for vertical scanning deflects the beam to the bottom of the raster. Then the rapid vertical retrace returns the beam to the top. Note that the maximum amplitude of the vertical sweep current brings the beam to the bottom of the raster. As shown in Fig. 2.2 (b) during vertical retrace the horizontal scanning continues and several lines get scanned during this period. Because of motion in the scene being televised, the information or brightness at the top of the target plate or picture tube screen normally changes by the time the beam returns to the top to recommence the whole process. This information is picked up during the next scanning cycle and the whole process is repeated 25 times to cause an illusion of continuity. The actual scanning sequence is however a little more complex than that just described and is explained in a later section of this chapter. It must however be noted, that both during horizontal retrace and vertical retrace intervals the scanning beams at the camera tube and picture tube are blanked and no picture information is either picked up or reproduced. Instead, on a time division basis, these short retrace intervals are utilized for transmitting distinct narrow pulses to keep the sweep oscillators of the picture tube deflection circuits of the receiver in synchronism with those of the camera at the transmitter. This ensures exact correspondence in scanning at the two ends and results in distortionless reproduction of the picture details.

Most scenes have brightness gradations in the vertical direction. The ability of the scanning beam to allow reproduction of electrical signals according to these variations and the capability of the human eye to resolve these distinctly, while viewing the reproduced picture, depends on the total number of lines employed for scanning. It is possible to arrive at some estimates of the number of lines necessary by considering the bar pattern shown in Fig. 2.3 (a), where alternate lines are black and white. If the thickness of the scanning beam is equal to the width of each white and black bar, and the number of scanning lines is chosen equal to the number of bars, the electrical information corresponding to the brightness of each bar will be correctly reproduced during the scanning process. Obviously the greater the number of lines into which the picture is divided in the vertical plane, the better will be the resolution.However, the total number of lines that need be employed is limited by the resolving capability of the human eye at the minimum viewing distance.
Beam spot Dark White D W D W D W

The maximum number of alternate light and dark elements (lines) which can be resolved by the eye is given by
1 Nv = αρ

where Nv = total number of lines (elements) to be resolved in the vertical direction, α = minimum resolving angle of the eye expressed in radians, and ρ = D/H = viewing-distance/picture height. For the eye this resolution is determined by the structure of the retina, and the brightness level of the picture. it has been determined experimently that with reasonable brightness variations and a minimum viewing distance of four times the picture height (D/H = 4), the angle that any two adjacent elements must subtend at the eye for distinct resolution is approximately one minute (1/60 degree). This is illustrated in Fig. 2.3 (b). Substituting these values of α and ρ we get

1 ≈ 860 (π / 180 × 1 / 60) × 4 Thus if the total number of scanning lines is chosen close to 860 and the scanning beam as illustrated in Fig. 2.3 (a) just passes over each bar (line) separately while scanning all the lines from top to bottom of the picture frame, a distinct pick up of the picture information results and this is the best that can be expected from the system. This perhaps explains the use of 819 lines in the original French TV system.
Nv =

Beam path

ANALYSIS AND SYNTHESIS OF TELEVISION PICTURES

21

In practice however, the picture elements are not arranged as equally spaced segments but have random distribution of black, grey and white depending on the nature of the picture details or the scene under consideration. Statistical analysis and subjective tests carried out to determine the average number of effective lines suggest that about 70 per cent of the total lines or segments get separately scanned in the vertical direction and the remaining 30 per cent get merged with other elements due to the beam spot falling equally on two consecutive lines. This is illustrated in Fig. 2.3 (c). Thus the effective number of lines distinctly resolved, i.e., Nr = Nv × k, where k is the resolution factor whose value lies between 0.65 to 0.75. Assuming the value of k = 0.7 we get, Nr = Nv × k = 860 × 0.7 = 602. However, there are other factors which also influence the choice of total number of lines in a TV system. Tests conducted with many observers have shown that though the eye can detect the effective sharpness provided by about 800 scanning lines, but the improvement is not very significant with line numbers greater than 500 while viewing pictures having motion. Also the channel bandwidth increases with increase in number of lines and this not only adds to the cost of the system but also reduces the number of television channels that can be provided in a given VHF or UHF transmission band. Thus as a compromise between quality and cost, the total number of lines inclusive of those lost during vertical retrace has been chosen to be 625 in the 625-B monochrome TV system. In the 525 line American system, the total number of lines has been fixed at 525 because of a somewhat higher scanning rate employed in this system.

2.4

FLICKER

Although the rate of 24 pictures per second in motion pictures and that of scanning 25 frames per second in television pictures is enough to cause an illusion of continuity, they are not rapid enough to allow the birghtness of one picture or frame to blend smoothly into the next through the time when the screen is blanked between successive frames. This results in a definite flicker of light that is very annoying to the observer when the screen is made alternately bright and dark. This problem is solved in motion pictures by showing each picture twice, so that 48 views of the scene are shown per second although there are still the same 24 picture frames per second. As a result of the increased blanking rate, flicker is eliminated. Interlaced scanning. In television pictures an effective rate of 50 vertical scans per second is utilized to reduce flicker. This is accomplished by increasing the downward rate of travel of the scanning electron beam, so that every alternate line gets scanned instead of every successive line. Then, when the beam reaches the bottom of the picture frame, it quickly returns to the top to scan those lines that were missed in the previous scanning. Thus the total number of lines are divided into two groups called ‘fields’. Each field is scanned alternately. This method of scanning is known as interlaced scanning and is illustrated in Fig. 2.4. It reduces flicker to an acceptable level since the area of the screen is covered at twice the rate. This is like reading alternate lines of a page from top to bottom once and then going back to read the remaining lines down to the bottom.

Fig. 2.4 Principle of interlaced scanning. Note that the vertical retrace time has been assumed to be zero.

In the 625 lime monochrome system, for successful interlaced scanning, the 625 lines of each frame or picture are divided into sets of 312.5 lines and each set is scanned alternately to cover the entire picture area. To achieve this the horizontal sweep oscillator is made to work at a frequency of 15625 Hz (312.5 × 50 = 15625) to scan the same number of lines per frame (15625/25 = 625 lines), but the vertical sweep circuit is run at a frequency of 50 instead of 25 Hz. Note that since the beam is now deflected from top to bottom in half the time and the horizontal oscillator is still operating at 15625 Hz, only half the total lines, i.e., 312.5 (625/2 = 312.5) get scanned during each vertical sweep. Since the first field ends in a half line and the second field commences at middle of the line on the top of the target plate or screen (see Fig. 2.4), the beam is able to scan the remaining 312.5 alternate lines during its downward journey. In all then, the beam scans 625 lines (312.5 × 2 = 625) per frame at the same rate of 15625 lines (312.5 × 50 = 15625) per second. Therefore, with interlaced scanning the flicker effect is eliminated without increasing the speed of scanning, which in turn does not need any increase in the channel bandwidth. It may be noted that the frame repetition rate of 25 (rather than 24 as used in motion pictures) was chosen to make the field frequency equal to the power line frequency of 50 Hz. This helps in reducing the undesired effects of hum due to pickup from the mains, because then such effects in the picture stay still, instead of drifting up or down on the screen. In the American TV system, a field frequency of 60 was adopted because the supply frequency is 60 Hz in USA. This brings the total number of lines scanned per second ((525/2) × 60 = 15750) lines to practically the same as in the 625 line system. Scanning periods. The waveshapes of both horizontal and vertical sweep currents are shown in Fig. 2.5. As shown there the retrace times involved (both horizontal and vertical) are due to physical limitations of practical scanning systems and are not utilized for transmitting or receiving any video signal. The nominal duration of the horizontal line as shown in Fig. 2.5 (a) is 64 µs (106/15625 = 64 µs), out of which the active line period is 52 µs and the remaining 12 µs

ANALYSIS AND SYNTHESIS OF TELEVISION PICTURES

23

is the line blanking period. The beam returns during this short interval to the extreme left side of the frame to start tracing the next line. Similarly with the field frequency set at 50 Hz, the nominal duration of the vertical trace (see Fig. 2.5(b)) is 20 ms (1/50 = 20 ms). Out of this period of 20 ms, 18.720 ms are spent in bringing the beam from top to bottom and the remaining 1.280 ms is taken by the beam to return back to the top to commence the next cycle. Since the horizontal and vertical sweep oscillators operate continuously to achieve the fast sequence of interlaced scanning, 20 horizontal lines

lines are lost per frame, as blanked lines during the retrace interval of two fields. This leaves the active number of lines, Na, for scanning the picture details equal to 625 – 40 = 585, instead of the 625 lines actually scanned per frame.
1(H) Trace Retrace f = 15625 Hz

t

52 ms

64 ms 12 ms

Fig. 2.5 (a) Horizontal deflection current.
i(v) f = 50 Hz

Trace

Retrace

18.720 ms

20 ms

t

1.280 ms

Fig. 2.5 (b) Vertical deflection current.

Scanning sequence. The complete geometry of the standard interlaced scanning pattern is illustrated in Fig. 2.6. Note that the lines are numbered in the sequence in which these are actually scanned. During the first vertical trace actually 292.5 lines are scanned. The beam starts at A, and sweeps across the frame with uniform velocity to cover all the picture elements in one horizontal line. At the end of this trace the beam then retraces rapidly to the left side of the frame as shown by the dashed line in the illustration to begin the next horizontal line. Note that the horizontal lines slope downwards in the direction of scanning because the vertical deflecting current simultaneously produces a vertical scanning motion, which is very slow

24

MONOCHROME AND COLOUR TELEVISION

compared with horizontal scanning. The slope of the horizontal trace from left to right is greater than during retrace from right to left. The reason is that the faster retrace does not allow the beam so much time to be deflected vertically. After line one, the beam is at the left side ready to scan line 3, omitting the second line. However, as mentioned earlier it is convenient to number the lines as they are scanned and so the next scanned line skipping one line, is numbered two and not three. This process continues till the last line gets scanned half when the vertical motion reaches the bottom of the raster or frame. As explained earlier skipping of lines is accomplished by doubling the vertical scanning frequency from the frame or picture repetition rate of 25 to the field frequency of 50 Hz. With the field frequency of 50 Hz the height of the raster is so set that 292.5 lines get scanned as the beam travels from top to bottom and reaches
A C C A

point B. Now the retrace starts and takes a period equal to 20 horizontal line periods to reach the top marked C. These 20 lines are known as inactive lines, as the scanning beam is cut-off during this period. Thus the second field starts at the middle of the raster and the first line scanned is the 2nd half of line number 313. The scanning of second field, starting at the middle of the raster automatically enables the beam to scan the alternative lines left unscanned during the first field. The vertical scanning motion otherwise is exactly the same as in the previous field giving all the horizontal lines the same slope downwards in the direction of scanning. As a result 292.5 lines again get scanned and the beam reaches the bottom of the frame when it has completed full scanning of line number 605. The inactive vertical retrace again begins and brings the beam back to the top at point A in a period during which 20 blanked horizontal lines (605 to 625) get scanned. Back at point A, the scanning beam has just completed two fields or one frame and is ready to start the third field covering the same area (no. of lines) as scanned during the first field. This process (of scanning fields) is continued at a fast rate of 50 times a second, which not only creates an illusion of continuity but also solves the problem of flicker satisfactorily.

ANALYSIS AND SYNTHESIS OF TELEVISION PICTURES

25

2.5

FINE STRUCTURE

The ability of the image reproducing system to represent the fine structure of an object is known as its resolving power or resolution. It is necessary to consider this aspect separately in the vertical and horizontal planes of the picture. Vertical resolution. The extent to which the scanning system is capable of resolving picture details in the vertical direction is referred to as its vertical resolution. It has already been explained that the vertical resolution is a function of the scanning lines into which the picture is divided in the vertical plane. Based on that discussion the vertical resolution in the 625 lines system can then be expressed as Vr = Na × k where Vr is the vertical resolution expressed in number of lines, Na is the active number of lines and k is the resolution factor (also known as Kell factor). Assuming a reasonable value of k = 0.69, Vr = 585 × 0.69 = 400 lines It is of interest to note that the corresponding resolution of 35 mm motion pictures is about 515 lines and thus produces greater details as compared to television pictures.
Alternate black and white lines of resolution

Horizontal resolution. The capability of the system to resolve maximum number of picture elements along the scanning lines determines horizontal resolution. This can be evaluated by considering a vertical bar pattern as shown in Fig. 2.7(a). It would be realistic to aim at equal vertical and horizontal resolution and as such the number of alternate black and white bars that should be considered is equal to Na × aspect ratio = 585 × 4/3 = 780 Before proceeding further it must be recognised that as all lines in the vertical plane are not fully effective, in a similar way all parts of an individual line are not fully effective all the time. As explained earlier, it ultimately depends on the random distribution of black and white areas in the picture. Thus for equal vertical and horizontal resolution, the same resolution factor may be used while determining the effective number of distinct picture elements in a horizontal line. Therefore, the effective number of alternate black and white segments in one horizontal line for equal vertical and horizontal resolution are : N = Na × aspect ratio × k = 585 × 4/3 × 0.69 = 533 To resolve these 533 squares or picture elements the scanning spot must develop a video signal of square wave nature switching continuously along the line between voltage levels corresponding to black and peak white. This is shown along the bar pattern drawn in Fig. 2.7(a). Since along one line there are 533/2 ≈ 267 complete cyclic changes, 267 complete square wave cycles get generated during the time the beam takes to travel along the width of the pattern. Thus the time duration th of one square wave cycle is equal to

th =
=

active period of each horizontal line number of cycles

∴

52 × 10 −6 seconds 267 thg frequency of the periodic wave

fh =

1 267 × 10 6 = = 5 MHz th 52

Since the consideration of both vertical and horizontal resolutions is based on identical black and white bars in the horizontal and vertical planes of the picture frame, it amounts to considering a chessboard pattern as the most stringent case and is illustrated in Fig. 2.7(b). Here each alternate black and white square element takes the place of bars for determining the capability of the scanning system to reproduce the fine structure of the object being televised. The actual size of each square element in the chess pattern is very small and is equal to thickness of the scanning beam. It would be instructive to know as an illustration that the size of such a square element on the screen of a 51 cm picture tube is about 0.5 mm2 only. Since the spacing of these small elements in the above consideration corresponds to the limiting resolution of the eye, it will distinguish only the alternate light and dark areas but not the shape of the variations along the scanning line. Thus the eye will fail to distinguish the difference between a square wave of brightness variation and a sine wave of brightness variation in the reproduced picture. Therefore, if the amplifier for the square-wave signal is capable of reproducing a sine-wave of frequency equal to the repetition frequency of the rectangular wave, it is satisfactory for the purpose of TV signals. It may be mentioned that even otherwise

ANALYSIS AND SYNTHESIS OF TELEVISION PICTURES

27

to handle a 5 MHz square wave would necessitate reproduction up to 11th harmonic of a periodic sinusoidal wave of 5 MHz by the associated electronic circuitry. This would mean a bandwidth of atleast up to 5 × 11 = 55 MHz which is excessive and almost impossible to provide in practice. Another justification for restricting the bandwidth up to 5 MHz is that in practice it is rare when alternate picture elements are black and white throughout the picture width and height, and a bandwidth up to 5 MHz has been found to be quite adequate to produce most details of the scene being televised. Therefore, the highest approximate modulating frequency ‘fh’ that the 625 line television system must be capable of handling for successful transmission and reception of picture details is fh =

In the second (525 line) widely used television system, where the active number of lines is 485 and the duration of one active line is 57 µs, the highest modulating frequency fh ≈ 4 MHz. This explains the allocation of 6 MHz as the channel bandwidth in USA and other countries employing the 525 line system in comparison to a channel bandwidth allocation of 7 MHz in countries that have adopted the 625 line system. Similarly in the French 819 TV system where the highest modulating frequency comes to 10.4 MHz a channel bandwidth of 14 MHz is allowed. Colour resolution and bandwidth. As explained above a bandwidth of 5 MHz (4 MHz in the American system) is needed for transmission of maximum horizontal detail in monochrome. However, this bandwidth is not necessary for the colour video signals. The reason is that the human eye’s colour response changes with the size of the object. For very small objects the eye can perceive only the brightness rather than the colours in the scene. Perception of colours by the eye is limited to objects which result in a video frequency output up to about 1.5 MHz. Thus the colour information needs much less bandwidth than monochrome details and can be easily accommodated in the channel bandwidth allotted for monochrome transmission. Low-frequency requirements. The analysis of the signals produced by the bar pattern gives no information regarding the low-frequency requirement of a video amplifier used to handle such signals. This requirement may be determined from consideration of a pattern shown in Fig. 2.8(a). The signal output during vertical excursions of the beam would be a square wave (see Fig. 2.8(b)) at vertical field frequency. It is apparent then, that any amplifier capable of reproducing this waveform would be required to have good square-wave response at 50 Hz. Any degradation in response as shown in Fig. 2.8(c) would result in brightness distortion. In order to have satisfactory square-wave response at field frequency, an amplifier must have good sine-wave response with negligible phase distortion down to a much lower frequency than the field frequency. In addition, to correct phase and amplitude response at the field frequency, it is necessary to preserve the dc component of the brightness signal. Thus a good frequency response from dc to about 5 MHz becomes necessary for true reproduction of the brightness variations and find details of any scene.

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MONOCHROME AND COLOUR TELEVISION

e(t)
V

t t(v)

Fig. 2.8(a) Single bar pattern.

Fig. 2.8(b) Ideal response to scanning of single bar pattern.

e(t)

t

Fig. 2.8(c) Distorted response due to poor low frequency response of the system.

Influence of number of lines on bandwidth. As the number of lines employed in a television picture is increased, the bandwidth necessary for a given quality of definition also increases. This is due to the fact that increasing the number of lines per picture decreases the time duration of each line. This means that the spot travels across the screen at a higher velocity and results in increase of the highest modulating frequency. For example doubling the number of lines per frame would very much improve the vertical resolution, infact it would get doubled but would need increasing the bandwidth in the same ratio. If now, it is required to increase the horizontal resolution so that it again equals the vertical resolution it would be necessary to scan double the number of alternate black and white signal elements in a line, and this would necessitate multiplying the original highest video frequency by a factor of four. The conclusion is that, if the number of lines employed in a television system is increased, it is necessary to increase the video frequency bandwidth in direct proportion to the increase in number of lines to maintain the same degree of vertical definition (as before), and in order to increase horizontal definition in the same proportion as the increase in vertical resolution the video frequency bandwidth must increase as the square of the increase in number of lines. Effect of interlaced scanning on bandwidth. As already explained, interlaced scanning reduces flicker. However, scanning 50 complete frames of 625 lines in a progressive manner would also eliminate flicker in the picture but this would need double the scanning speed which in turn would double the video frequencies corresponding to the picture elements in a line. This would necessitate double the channel bandwidth of that required with interlaced scanning. It should be noted that by employing interlaced scanning, the basic concept of interchangeability of time and bandwidth is not violated, because more time in allowed for transmission and this results in decrease of bandwidth needed for each TV channel. Thus interlaced scanning reduces flicker and conserves bandwidth. Effect of field frequency on bandwidth. With increase in field frequency the time available for each field decreases and this results in a proportionate decrease of the active line period.

ANALYSIS AND SYNTHESIS OF TELEVISION PICTURES

29

Hence, bandwidth increases in direct proportion to the increase in the field frequency. Bandwidth requirement for transmission of synchronising pulses. The equalizing pulses to be discussed later have a pulse width of 2.3 µs with an allowed rise time of 0.2 µs. The highest sinusoidal frequency which must lie in the pass band of the system for effective transmission of these pulses is given by the expression :

10 6 1 = = 2.5 MHz 2 × 0.2 2 × allowed rise time It is then clear that all sync pulses are safely preserved in the video circuitry where, as has been shown, a frequency bandwidth considerably in excess of this figure has to be maintained in order to preserve the required picture definition.
Highest necessary frequency = Interlace error. As explained earlier interlaced scanning provides a means of decreasing the effect of flicker in the TV picture without increasing the system bandwidth. The selection of 2 : 1 as the interlace ratio is the simplest with least circuit complications. Here, by selecting an odd number of lines, the symmetry in frame blanking pulses is achieved and this enables perfect interlaced scanning. Any error in scanning timings and sequence would leave a large number of picture elements unresolved and thus the quality of the reproduced picture gets impaired. Fig. 2.9 shows various cases of interlace error. For convenience of explanation the retrace time has been assumed to be zero. Interlace error occurs due to the time difference in starting the second field. For perfect interlace the second field should start from point ‘b’ (see Fig. 2.9 (a)), i.e., 32 µs away from ‘a’, the starting point of the first field. If it starts early or late interlace error will be there. For a 16 µs delay in the start of the second field (Fig. 2.9 (b)), starting points of the two fields will be 48 µs apart instead of the desired 32 µs. Then the percentage interlace error

48 − 32 × 100 = 50% 32 if the second field starts 16 µs early even then the error would be 50%. For a delay of 32 µs the two fields will overlap (Fig. 2.9 (c)) and the interlace error would be 100%, i.e., half the picture area will go unscanned.
=
Assumed starting point of one scanning field Start of alternate field a b a
16 ms

The two fields overlap a, b

b

32 ms

32 ms

32 ms

(a) Perfect interlace

(b) 50% interlace error

(c) 100% interlace error

Fig. 2.9 Examples of interlace error.

The above examples demonstrate that incorrect start of any field produces vertical displacement between the lines of the two fields. This brings these lines closer leaving gaps

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MONOCHROME AND COLOUR TELEVISION

between the pairs thus formed. The result is a deterioration of the picture’s vertical resolution because certain areas do not get scanned at all. For perfect interlaced scanning it is essential that the starting points at the top of the frame is separated exactly one half line between first and second fields. To achieve this it is necessary to feed two regularly spaced synchronising pulses to the field time base during each frame period. One of these pulses must arrive in the middle of a line and the next at the end of a line. This is shown in Fig. 2.10. Thus the vertical time base must be triggered 50 times per second in the manner explained above. For half line separation between the two fields only the topmost and the extreme bottom lines are then half lines whereas the remaining lines are all full lines. If there are x number of full lines per field, where x may be even or odd, the total number of full lines per frame is then 2x, an even number. To this, when the two half lines get added the total number of lines per frame becomes odd. Thus for interlaced scanning the total number of lines in any TV system must be odd. With an even number of lines the two fields are bound to fall on each other and interlaced scanning would not take place.
W

Further for correct interlacing it becomes necessary that at the transmitter automatic frequency control must be utilized to maintain a horizontal scanning frequency that is exactly 312.5 times as great as the field frequency, i.e., 50 Hz. This is accomplished by generating a stable frequency at 15625 Hz by crystal controlled oscillator circuits. A frequency doubling circuit produces a frequency of 31250 Hz, which is utilized to control the correct generation of equalizing and vertical sync pulses. Four frequency division circuits each with a ratio of 5 : 1 are employed to derive 50 Hz, the vertical scanning frequency (31250 = 5 × 5 × 5 × 5 × 50). Thus all the required frequencies are derived from a common stable source and they automatically remain interlocked in the correct ratios. To achieve this, i.e., frequency division, the total number of lines per frame must be a product of small whole numbers. The frame frequency of 625 satisfies all the above requirements. Similarly 525 lines in the American system and 819 lines in the French system also meet these requirements.

ANALYSIS AND SYNTHESIS OF TELEVISION PICTURES

31

Comparison of various TV systems. Picture and sound signal standards for the principal monochrome television systems are given at the end of chapter 4. The CCIR 625-B monochrome system used in most parts of Europe and adopted by India has a video bandwidth of 5 MHz, whereas the British 625 line system has a video bandwidth of 5.5 MHz. Obviously, here 0.73 has been used as the resolution factor instead of the 0.69 used in our system. So the British system is marginally better than the European system. The French TV system employs 819 lines with a video bandwidth of 10.4 MHz. This system therefore has both much improved vertical resolution and a better horizontal resolution. The American 525 line system employs a frame frequency of 30 as compared to 25 in the CCIR 625-B monochrome system. Thus, the line frequency in this system is 15750, which compares very closely to our system where the line frequency is 15625. However, the American system employs a bandwidth of 4 MHz which suggests that the horizontal resolution of this system is less than all other systems in use. It must be noted that the number of lines employed by a given TV system is not in itself, a guide to the quality of resolution available from the system. It is true that greater the number of lines the better the vertical resolution, but an assessment of the horizontal resolution, i.e., the bandwidth employed by the system is a better overall guide to the quality of definition.

2.6

TONAL GRADATION

In addition to proper bandwidth required to produce the details allowed by the scanning system at the transmitting end and the picture tube at the receiving end, the signal-transmission system should have proper transfer characteristics to preserve same brightness gradation as the eye would perceive when viewing the scene directly. Any non-linearity in the pick-up and picture tube should also be corrected by providing inverse nonlinearities in the channel circuitry to obtain overall linear characteristics. Note that the sensation in the eye to detect changes or brightness is logarithmic in nature and this must be taken into account while designing the overall channel. Various other factors that influence the tonal quality of the reproduced picture are : (a) Contrast. This is the difference in intensity between black and white parts of the picture over and above the brightness level. (b) Contrast ratio. The ratio of maximum to minimum brightness relative to the original picture is called contrast ratio. In broad daylight the variations in brightness are very wide with ratio as high as 10000 : 1, whereas the picture tube, because of certain limitations, cannot produce a contrast with variations more than 50 : 1 or atmost 100 : 1. Ratio of brightness variations in the reproduced picture on the screen of the picture tube, to the brightness variations in the original scene is known as Gamma of the picture. Its value is close to 0.5. In studios, under controlled conditions of light, the variations are less wide than outside and so the brightness variations that can be reproduced by the picture tube are not very much different than that of the scene. Realism is still maintained because the viewer does not actually see the scene being televised. Another factor which makes stringent demands from the system unnecessary is the fact that our eye can accommodate not more than 10 : 1 variations of light intensity at any time. Too bright a representation of the bright areas in a picture would make

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MONOCHROME AND COLOUR TELEVISION

grey areas appear as dark in comparison. This is true at all levels of light intensity with brightness variations in relative ratios of 10 : 1. When a TV receiver is off, there is no beam impinging on the fluorescent screen of the picture tube and no light gets emitted. Then with normal light in the room the screen appears as dull white. But when the receiver is no, and a TV programme is being received the bright portions of the scene appear quite bright because the corresponding amplitude of the video signal makes the control-grid of the picture tube much less negative and the consequent increased beam current causes more light on the screen. However, for a very dark portion of the scene the corresponding video signal makes the grid highly negative with respect to the cathode and thus cuts-off the beam current and no light is emitted on the corresponding portions on the screen. These areas appear to the eye as dark in comparison with the high light areas of the screen, whereas the same area in the absence of beam current when the set was off appeared close to a white shade. This as explained earlier is due to the logarithmic response of the human eye and its inability to accommodate light intensity variations greater than 10 : 1. (c) Viewing distance. The viewing distance from the screen of the TV receiver should not be so large that the eye cannot resolve details of the picture. The distance should also not be so small that picture elements become separately visible. The above conditions are met when the vertical picture size subtends an angle of approximately 15° at the eye. The distance also depends on habit, varies from person to person, and lies between 3 to 8 times the picture height. Most people prefer a distance close to five times the picture height. While viewing TV, a small light should be kept on in the room to reduce contrast. This does not strain the eyes and there is less fatigue.

Review Questions
1. 2. 3. 4. Justify the choice of rectangular frame with width to height ratio = 4/3 for television transmission and reception. How is the illusion of continuity created in television pictures ? Why has the frame reception rate been chosen to be 25 and not 24 as in motion pictures ? What do you understand by interlaced scanning ? Show that it reduces flicker and conserve bandwidth. What do you understand by active and blanking periods in horizontal and vertical scanning ? Give the periods of nominal, active and retrace intervals of horizontal and vertical scanning as used in the 625 line system. How many horizontal lines get traced during each vertical retrace ? What is the active number of lines that are actually used for picture information pick up and reception ? Draw a picture frame chart showing the total number of active and inactive lines during each field and establish the need for terminating the first field in a half line and the beginning the second at the middle of a line at the top. Justify the choice of 625 lines for TV transmission. Why is the total number of lines kept odd in all television systems ? What is the significance of choosing the number of lines as 625 and not 623 or 627 ? What do you understand by resolution or Kell-factor ? How does it affect the vertical resolution of a television picture ? Show that the vertical resolution increases with increase in number of scanning lines.

5. 6.

7.

8.

ANALYSIS AND SYNTHESIS OF TELEVISION PICTURES

33

9.

What is meant by equal vertical and horizontal ‘resolution ?’ Derive an expression for the highest modulating frequency in a television system and show that it is nearly 5 MHz. in the 625-B monochrome system.

10. Show that if the number of lines employed in a TV system is increased then the highest video frequency must increase as the square of the increase in number of lines for equal improvement in vertical and horizontal resolution. 11. Show that the 625-B TV system is only marginally superior to the 525 line American system. 12. What do you understand by interlace error and how does it affect the quality of the picture ? Calculate the percentage interlace error when the second field is delayed by 8 µs. Retrace time may be assumed to be negligible. 13. In the British 625 lines system the resolution factor employed is 0.73 instead of 0.69 as used in the 625-B monochrome system. All other scanning details remaining the same, calculate the highest modulating frequency used in the British system. 14. Explain the need for providing very good low frequency response and phase characteristics in amplifiers used in any TV link, for proper reproduction of brightness variations. 15. The relevant data for a closed circuit TV system is given below. Calculate the highest modulating frequency that will be generated while scanning the most stringent case of alternate black and white dots for equal vertical and horizontal resolution. No. of lines Interlace ratio Picture repetition rate Aspect ratio Vertical retrace time Horizontal retrace time Ans = 250 =1:1 = 50/sec = 4/3 = 10% of the picture frame time = 20% of the total line time ≈ 2 MHz

Assume resolution factor = 0.8 16. Explain the meaning of terms-tonal gradation, contrast, contrast ratio and gamma of the picture. When a TV receiver is off, no electron beam strikes the picture tube screen and the screen face looks a dull white. With the set on and a black and white picture showing on the screen, no electron beam impinges on the darker area of the reproduced picture. But these areas now appear quite black instead of the dull white of the switched-off set. Explain the reason for this difference in appearance.

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3
Composite Video Signal

3
Composite Video Signal
Composite video signal consists of a camera signal corresponding to the desired picture information, blanking pulses to make the retrace invisible, and synchronizing pulses to synchronize the transmitter and receiver scanning. A horizontal synchronizing (sync) pulse is needed at the end of each active line period whereas a vertical sync pulse is required after each field is scanned. The amplitude of both horizontal and vertical sync pulses is kept the same to obtain higher efficiency of picture signal transmission but their duration (width) is chosen to be different for separating them at the receiver. Since sync pulses are needed consecutively and not simultaneously with the picture signal, these are sent on a time division basis and thus form a part of the composite video signal.

3.1

VIDEO SIGNAL DIMENSIONS

Figure 3.1 shows the composite video signal details of three different lines each corresponding to a different brightness level of the scene. As illustrated there, the video signal is constrained to vary between certain amplitude limits. The level of the video signal when the picture detail being transmitted corresponds to the maximum whiteness to be handled, is referred to as peak-white level. This is fixed at 10 to 12.5 percent of the maximum value of the signal while the black level corresponds to approximately 72 percent. The sync pulses are added at 75 percent level called the blanking level. The difference between the black level and blanking level is known as the ‘Pedestal’. However, in actual practice, these two levels, being very close, tend to merge with each other as shown in the figure. Thus the picture information may vary between 10 percent to about 75 percent of the composite video signal depending on the relative brightness of the picture at any instant. The darker the picture the higher will be the voltage within those limits. Note that the lowest 10 percent of the voltage range (whiter than white range) is not used to minimize noise effects. This also ensures enough margin for excessive bright spots to be accommodated without causing amplitude distortion at the modulator. At the receiver the picture tube is biased to ensure that a received video voltage corresponding to about 10 percent modulation yields complete whiteness at that particular point on the screen, and an analogous arrangement is made for the black level. Besides this, the television receivers are provided with ‘brightness’ and ‘contrast’ controls to enable the viewer to make final adjustments as he thinks fit. D.C. component of the video signal. In addition to continuous amplitude variations for individual picture elements, the video signal has an average value or dc component 36

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37

corresponding to the average brightness of the scene. In the absence of dc component the receiver cannot follow changes in brightness, as the ac camera signal, say for grey picture elements on a black background will then be the same as a signal for white area on a grey back-ground. In Fig. 3.1, dc components of the signal for three lines have been identified, each representing a different level of average brightness in the scene. It may be noted that the break shown in the illustration after each line signal is to emphasize that dc component of the video signal is the average value for complete frames rather than lines since the background information of the picture indicates the brightness of the scene. Thus Fig. 3.1 illustrates the concept of change in the average brightness of the scene with the help of three lines in separate frames because the average brightness can change only from frame to frame and not from line to line.
v/v max%

Pedestal height. As noted in Fig. 3.1 the pedestal height is the distance between the pedestal level and the average value (dc level) axis of the video signal. This indicates average brightness since it measures how much the average value differs from the black level. Even when the signal loses its dc value when passed through a capacitor-coupled circuit the distance between the pedestal and the dc level stays the same and thus it is convenient to use the pedestal level as the reference level to indicate average brightness of the scene. Setting the pedestal level. The output signal from the TV camera is of very small amplitude and is passed through several stages of ac coupled high gain amplifiers before being coupled to a control amplifier. Here sync pulses and blanking pulses are added and then clipped at the correct level to form the pedestals. Since the pedestal height determines the average brightness of the scene, any smaller value than the correct one will make the scene darker while a larger pedestal height will result in higher average brightness. The video control operator who observes the scene at the studio sets the level for the desired brightness in the reproduced picture which

D.C. level

Pedestal height

Dark level

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MONOCHROME AND COLOUR TELEVISION

he is viewing on a monitor receiver. This is known as dc insertion because this amounts to adding a dc component to the ac signal. Once the dc insertion has been acomplished the pedestal level becomes the black reference and the pedestal height indicates correct relative brightness for the reproduced picture. However, the dc level inserted in the control amplifier is usually lost in succeeding stages because of capacitive coupling, but still the correct dc component can be reinserted when necessary because the pedestal height remains the same. The blanking pulses. The composite video signal contains blanking pulses to make the retrace lines invisible by raising the signal amplitude slightly above the black level (75 per cent) during the time the scanning circuits produce retraces. As illustrated in Fig. 3.2, the composite video signal contains horizontal and vertical blanking pulses to blank the corresponding retrace intervals. The repetition rate of horizontal blanking pulses is therefore equal to the line scanning frequency of 15625 Hz. Similarly the frequency of the vertical blanking pulses is equal to the field-scanning frequency of 50 Hz. It may be noted that though the level of the blanking pulses is distinctly above the picture signal information, these are not used as sync pulses. The reason is that any occasional signal corresponding to any extreme black portion in the picture may rise above the blanking level and might conceivably interfere with the synchronization of the scanning generators. Therefore, the sync pulses, specially designed for triggering the sweep oscillators are placed in the upper 25 per cent (75 per cent to 100 per cent of the carrier amplitude) of the video signal, and are transmitted along with the picture signal.
100 90 80 72% 75 70 60
Amplitude %

Sync pulse and video signal amplitude ratio. The overall arrangement of combining the picture signal and sync pulses may be thought of as a kind of voltage division multiplexing where about 65 per cent of the carrier amplitude is occupied by the video signal and the upper

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25 per cent by the sync pulses. Thus, as shown in Fig. 3.1, the final radiated signal has a picture to sync signal ratio (P/S) equal to 10/4. This ratio has been found most satisfactory because if the picture signal amplitude is increased at the expense of sync pulses, then when the signal to noise ratio of the received signal falls, a point is reached when the sync pulse amplitude becomes insufficient to keep the picture locked even though the picture voltage is still of adequate amplitude to yield an acceptable picture. On the other hand if sync pulse height is increased at the expense of the picture detail, then under similar conditions the raster remains locked but the picture content is of too low an amplitude to set up a worthwhile picture. A ratio of P/S = 10/4, or thereabout, results in a situation such that when the signal to noise ratio reaches a certain low level, the sync amplitude becomes insufficient, i.e., the sync fails at the same time as the picture ceases to be of entertainment value. This represents the most efficient use of the television system.

3.2

HORIZONTAL SYNC DETAILS

The horizontal blanking period and sync pulse details are illustrated in Fig. 3.3. The interval between horizontal scanning lines is indicated by H. As explained earlier, out of a total line

Fig. 3.3 Horz line and sync details compared to horizontal deflection sawtooth and picture space on the raster.

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MONOCHROME AND COLOUR TELEVISION

period of 64 µs, the line blanking period is 12 µs. During this interval a line synchronizing pulse is inserted. The pulses corresponding to the differentiated leading edges of the sync pulses are actually used to synchronize the horizontal scanning oscillator. This is the reason why in Fig. 3.3 and other figures to follow, all time intervals are shown between sync pulse leading edges. The line blanking period is divided into three sections. These are the ‘front porch’, the ‘line sync’ pulse and the ‘back porch’. The time intervals allowed to each part are summarized below and their location and effect on the raster is illustrated in Fig. 3.3. Details of Horizontal Scanning
Period Total line (H) Horz blanking Horz sync pulse Front porch Back porch Visible line time Time (µs) 64 12 ± .3 4.7 ± 0.2 1.5 ± .3 5.8 ± .3 52

Front porch. This is a brief cushioning period of 1.5 µs inserted between the end of the picture detail for that line and the leading edge of the line sync pulse. This interval allows the receiver video circuit to settle down from whatever picture voltage level exists at the end of the picture line to the blanking level before the sync pulse occurs. Thus sync circuits at the receiver are isolated from the influence of end of the line picture details. The most stringent demand is made on the video circuits when peak white detail occurs at the end of a line. Despite the existence of the front porch when the line ends in an extreme white detail, and the signal amplitude touches almost zero level, the video voltage level fails to decay to the blanking level before the leading-edge of the line sync pulse occurs. This results in late triggering of the time base circuit thus upsetting the ‘horz’ line sync circuit. As a result the spot (beam) is late in arriving at the left of the screen and picture information on the next line is displaced to the left. This effect is known as ‘pulling-on-whites’. Line sync pulse. After the front proch of blanking, horizontal retrace is produced when the sync pulse starts. The flyback is definitely blanked out because the sync level is blacker than black. Line sync pulses are separated at the receiver and utilized to keep the receiver line time base in precise synchronism with the distant transmitter. The nominal time duration for the line sync pulses is 4.7 µs. During this period the beam on the raster almost completes its back stroke (retrace) and arrives at the extreme left end of the raster. Back porch. This period of 5.8 µs at the blanking level allows plenty of time for line flyback to be completed. It also permits time for the horizontal time-base circuit to reverse direction of current for the initiation of the scanning of next line. Infact, the relative timings are so set that small black bars (see Fig. 3.3) are formed at both the ends of the raster in the horizontal plane. These blanked bars at the sides have no effect on the picture details reproduced during the active line period.

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The back porch* also provides the necessary amplitude equal to the blanking level (reference level) and enables to preserve the dc content of the picture information at the transmitter. At the receiver this level which is independent of the picture details is utilized in the AGC (automatic gain control) circuits to develop true AGC voltage proportional to the signal strength picked up at the antenna.

3.3

VERTICAL SYNC DETAILS

The vertical sync pulse train added after each field is somewhat complex in nature. The reason for this stems from the fact that it has to meet several exacting requirements. Therefore, in order to fully appreciate the various constituents of the pulse train, the vertical sync details are explored step by step while explaining the need for its various components. The basic vertical sync added at the end of both even add odd fields is shown in Fig. 3.4. Its width has to be kept much larger than the horizontal sync pulse, in order to derive a suitable field sync pulse at the receiver to trigger the field sweep oscillator. The standards specify that the vertical sync period should be 2.5 to 3 times the horizontal line period. If the width is less than this, it becomes difficult to distinguish between horizontal and vertical pulses at the receiver.
End of second (even) field 623 H 624 H 625 H Beginning of first (odd) field Lines 1, 2, 3rd 1st half

Fig. 3.4 Composite video waveforms showing horizontal and basic vertical sync pulses at the end of (a) second (even) field, (b) first (odd) field. Note, the widths of horizontal blanking intervals and sync pulses are exaggerated. *In colour TV transmission a short sample (8 to 10 cycles) of the colour subcarrier oscillator output is sent to the receiver for proper detection of colour signal sidebands. This is known as colour burst and is located at the back porch of the horizontal blanking pedestal.

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MONOCHROME AND COLOUR TELEVISION

If the width is greater than this, the transmitter must operate at peak power for an unnecessarily long interval of time. In the 625 line system 2.5 line period (2.5 × 64 = 160 µs) has been allotted for the vertical sync pulses. Thus a vertical sync pulse commences at the end of 1st half of 313th line (end of first field) and terminates at the end fo 315th line. Similarly after an exact interval of 20 ms (one field period) the next sync pulse occupies line numbers— 1st, 2nd and 1st half of third, just after the second field is over. Note that the beginning of these pulses has been aligned in the figure to signify that these must occur after the end of vertical stroke of the beam in each field, i.e., after each 1/50th of a second. This alignment of vertical sync pulses, one at the end of a half-line period and the other after a full line period (see Fig. 3.4), results in a relative misalignment of the horizontal sync pulses and they do not appear one above the other but occur at half-line intervals with respect to each other. However, a detailed examination of the pulse trains in the two fields would show that horizontal sync pulses continue to occur exactly at 64 µs intervals (except during the vertical sync pulse periods) throughout the scanning period from frame to frame and the apparent shift of 32 µs is only due to the alignment of vertical sync instances in the figure. As already mentioned the horizontal sync information is extracted from the sync pulse train by differentiation, i.e., by passing the pulse train through a high-pass filter. Indeed pulses corresponding to the differentiated leading edges of sync pulses are used to synchronise the horizontal scanning oscillator. The process of deriving these pulses is illustrated in Fig. 3.5. Furthermore, receivers often use monostable multivibrators to generate horizontal scan, and so a pulse is required to initiate each and every cycle of the horizontal oscillator in the receiver.
Sync pulses

This brings out the first and most obvious shortcoming of the waveforms shown in Fig. 3.4. The horizontal sync pulses are available both during the active and blanked line periods but there are no sync pulses (leading edges) available during the 2.5 line vertical sync period. Thus the horizontal sweep oscillator that operates at 15625 Hz, would tend to step out of synchronism during each vertical sync period. The situation after an odd field is even worse. As shown in Fig. 3.4, the vertical blanking period at the end of an odd field begins midway through a horizontal line. Consequently, looking further along this waveform, we see that the leading edge of the vertical sync pulse comes at the wrong time to provide synchronization for the horizontal oscillator. Therefore, it becomes necessary to cut slots in the vertical sync pulse at half-line-intervals to provide horizontal sync pulses at the correct instances both after even and odd fields. The technique is to take the video signal amplitude back to the blanking level

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4.7 µs before the line pulses are needed. The waveform is then returned back to the maximum level at the moment the line sweep circuit needs synchronization. Thus five narrow slots of 4.7 µs width get formed in each vertical sync pulse at intervals of 32 µs. The trailing but rising edges of these pulses are actually used to trigger the horizontal oscillator. The resulting waveforms together with line numbers and the differentiated output of both the field trains is illustrated in Fig. 3.6. This insertion of short pulses is known as notching or serration of the broad field pulses. Note that though the vertical pulse has been broken to yield horizontal sync pulses, the effect on the vertical pulse is substantially unchanged. It still remains above the blanking voltage level all of the time it is acting. The pulse width is still much wider than the horizontal pulse width and thus can be easily separated at the receiver. Returning to Fig. 3.6 it is seen that each horizontal sync pulse yields a positive spiked output from its leading edge and a negative spiked pulse from its trailing edge. Time-constant of the differentiating circuit is so chosen, that by the time a trailing edge arrives, the pulse due to the leading edge has just about decayed. The negative-going triggering pulses may be removed with a diode since only the positive going pulses are effective in locking the horizontal oscillator.
End of 2nd field 623 624 625 1 4.7 ms 623 624 625 1 2 1 2 3 2 4 5 3

27.3 ms 3 4 5 6

(a) End of 1st field 311 312 313 1 2 314 3 4 315 5 316

311

312

313

314

315

316

317

318

(b)

Fig. 3.6 Differentiating waveforms (a) pulses at the end of even (2nd) field and the corresponding output of the differentiator (H.P.F.) (b) pulses at the end of odd (1st) field and the corresponding output of the differentiator (H.P.F.) Note, the differentiated pulses bearing line numbers are the only ones needed at the end of each field.

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MONOCHROME AND COLOUR TELEVISION

However, the pulses actually utilized are the ones that occur sequentially at 64 µs intervals. Such pulses are marked with line numbers for both the fields. Note that during the intervals of serrated vertical pulse trains, alternate vertical spikes are utilized. The pulses not used in one field are the ones utilized during the second field. This happens because of the half-line difference at the commencement of each field and the fact that notched vertical sync pulses occur at intervals of 32 µs and not 64 µs as required by the horizontal sweep oscillator. The pulses that come at a time when they cannot trigger the oscillator are ignored. Thus the requirement of keeping the horizontal sweep circuit locked despite insertion of vertical sync pulses is realized. Now we turn to the second shortcoming of the waveform of Fig. 3.4. First it must be mentioned that synchronization of the vertical sweep oscillator in the receiver is obtained from vertical sync pulses by integration. This is illustrated in Fig. 3.5 where the time-constant R2C2 is chosen to be large compared to the duration of horizontal pulses but not with respect to width of the vertical sync pulses. The integrating circuit may equally be looked upon as a lowpass filter, with a cuit-off frequency such that the horizontal sync pulses produce very little output, while the vertical pulses have a frequency that falls in the pass-band of the filter. The voltage built across the capacitor of the low-pass filter (integrating circuit) corresponding to the sync pulse trains of both the fields is shown in Fig. 3.7. Note that each horizontal pulse causes a slight rise in voltage across the capacitor but this is reduced to zero by the time the next pulse arrives. This is so, because the charging period for the capacitor is only 4.7 µs and the voltage at the input to the integrator remains at zero for the rest of the period of 59.3 µs. Hence there is no residual voltage across the vertical filter (L.P. filter) due to horizontal syncpulses. Once the broad serrated vertical pulse arrives the voltage across the output of the filter starts increasing. However, the built up voltage differs for each field. The reason is not difficult to find. At the beginning of the first field (odd field) the last horz sync pulse corresponding to the beginning of 625th line is separated from the 1st vertical pulse by full one line and any voltage developed across the filter will have enough time to return to zero before the arrival of the first vertical pulse, and thus the filter output voltage builds up from zero in response to the five successive broad vertical sync pulses. The voltage builds up because the capacitor has more time to charge and only 4.7 µs to discharge. The situation, however, is not the same for the beginning of the 2nd (even) field. Here the last horizontal pulse corresponding to the beginning of 313th line is separated from the first vertical pulse by only half-a-line. The voltage developed across the vertical filter will thus not have enough time to reach zero before the arrival of the first vertical pulse, which means that the voltage build-up does not start from zero, as in the case of the 1st field. The residual voltage on account of the half line discrepancy gets added to the voltage developed on account of the broad vertical pulses and thus the voltage developed across the output filter is some what higher at each instant as compared to the voltage developed at the beginning of the first-field. This is shown in dotted chain line in Fig. 3.7. The vertical oscillator trigger potential level marked as trigger level in the diagram (Fig. 3.7) intersects the two filter output profiles at different points which indicates that in the case of second field the oscillator will get triggered a fraction of a second too soon as compared

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to the first field. Note that this inequlity in potential levels for the two fields continues during the period of discharge of the capacitor once the vertical sync pulses are over and the horizontal sync pulses take-over. Though the actual time difference is quite short it does prove sufficient to upset the desired interlacing sequence.
End of 2nd field 625 (a) End of 1st field 312 (b) After end of 2nd field v0 Trigger level After end of 1st field 2nd field 313 1 1st field

Time error Trigger pulse for the vertical oscillator (c) 0 t

Fig. 3.7 Integrating waveforms (a) pulses at the end of 2nd (even) field (b) pulses at the end of 1st (odd) field (c) integrator output. Note the above sync pulses have purposely been drawn without equalizing pulses.

Equalizing pulses. To take care of this drawback which occurs on account of the halfline discrepancy five narrow pulses are added on either side of the vertical sync pulses. These are known as pre-equalizing and post-equalizing pulses. Each set consists of five narrow pulses occupying 2.5 lines period on either side of the vertical sync pulses. Pre-equalizing and postequalizing pulse details with line numbers occupied by them in each field are given in Fig. 3.8. The effect of these pulses is to shift the half-line discrepancy away both from the beginning and end of vertical sync pulses. Pre-equalizing pulses being of 2.3 µs duration result in the discharge of the capacitor to essentially zero voltage in both the fields, despite the half-line discrepancy before the voltage build-up starts with the arrival of vertical sync pulses. This is illustrated in Fig. 3.9. Post-equalizing pulses are necessary for a fast discharge of the capacitor to ensure triggering of the vertical oscillator at proper time. If the decay of voltage across the capacitor is slow as would happen in the absence of post-equalizing pulses, the oscillator may trigger at the trailing edge which may be far-away from the leading edge and this could lead to an error in triggering. Thus with the insertion of narrow pre and post equalizing pulses, the voltage rise and fall profile is essentially the same for both the field sequences (see Fig. 3.9) and the vertical oscillator is triggered at the proper instants, i.e., exactly at an interval of 1/50th of a second.

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MONOCHROME AND COLOUR TELEVISION

This problem could possibly also be solved by using an integrating circuit with a much larger time constant, to ensure that the capacitor remains virtually uncharged by the horizontal pulses. However, this would have the effect of significantly reducing the integrator output for vertical pulses so that a vertical sync amplifier would have to be used. In a broadcasting situation, there are thousands of receivers for every transmitter. Consequently it is much more efficient and economical to cure this problem in one transmitter than in thousands of receivers. This, as explained above, is achieved by the use of pre and post equalizing pulses. The complete pulse trains for both the fields incorporating equalizing pulses are shown in Fig. 3.10.
29.7 ms 2.3 + 0.1 ms

1st 2nd field ending 1st field ending 2nd half of 623 311

2nd 624

3rd

4th 625 312

5th

1st half of 313

(a) Pre-sync equalizing pulses (five)

1st 2nd half of 3rd 316

2nd 4

3rd 1st field

4th 5

5th

2nd field 317 1st half of 318 (b) Post-sync equalizing pulses (five)

Fig. 3.8 Pre-sync equalizing and Post-sync equalizing pulses.

From the comparison of the horizontal and vertical output pulse forms shown in Figs. 3.7 and 3.9 it appears that the vertical trigger pulse (output of the low-pass filter) is not very sharp but actually it is not so. The scale chosen exaggerates the extent of the vertical pulses. The voltage build-up period is only 160 µs and so far as the vertical synchronizing oscillator is concerned this pulse occurs rapidly and represents a sudden change in voltage which decays very fast. The polarity of the pulses as obtained at the outputs of their respective fields may not be suitable for direct application in the controlled synchronizing oscillator and might need inversion depending on the type of oscillator used. This aspect will be fully developed in the chapter devoted to vertical and horizontal oscillators.

Approximate location of line numbers. The serrated vertical sync pulse forces the vertical deflection circuity to start the flyback. However, the flyback generally does not begin with the start of vertical sync because the sync pulse must build up a minimum voltage across the capacitor to trigger the scanning oscillator. If it is assumed that vertical flyback starts with the leading edge of the fourth serration, a time of 1.5 lines passes during vertical sync before vertical flyback starts. Also five equalizing pulses occur before vertical sync pulse train starts. Then four lines (2.5 + 1.5 = 4) are blanked at the bottom of the pricture before vertical retrace begins. A typical vertical retrace time is five lines. Thus the remaining eleven (20 – (4 + 5) = 11) lines are blanked at the top of the raster. These lines provide the sweep oscillator enough time to adjust to a linear rise for uniform pick-up and reproduction of the picture.

3.5

FUNCTIONS OF VERTICAL PULSE TRAIN

By serrating the vertical sync pulses and the providing pre- and post-equalizing pulses the following basic requirements necessary for successful interlaced scanning are ensured. (a) A suitable field sync pulse is derived for triggering the field oscillator. (b) The line oscillator continues to receive triggering pulses at correct intervals while the process of initiation and completion of the field time-base stroke is going on. (c) It becomes possible to insert vertical sync pulses at the end of a line after the 2nd field and at the middle of a line at the end of the 1st field without causing any interlace error. (d) The vertical sync build up at the receiver has precisely the same shape and timing on odd and even fields.

3.6

SYNC DETAILS OF THE 525 LINE SYSTEM

In the 525 line American TV system where the total number of lines scanned per second is 15750, the sync pulse details are as under :
Details of Horz Blanking Period Field line (H) Horz blanking Horz sync pulse Front porch Back porch Visible line Details of vertical Blanking Period Total field (V) period Visible field time Vertical blanking Time = 1/60 sec. = 16.7 ms = 15 to 16 ms = 0.8 to 1.3 ms Time (µs) 63.5 9.5 to 11.5 4.75 ± 0.5 1.26 (minimum) 3.81 (minimum) 52 to 54

50
Total duration of six (serrated) vertical sync pulses Each serrated pulse Each equalizing pulse (Six pre- and six post-equailzing pulses are provided at H/2 intervals)

Fig. P3.2 4. Sketch the details of horizontal blanking and sync pulses. Label on it (i) front porch, (ii) horizontal sync pulse, (iii) back porch and (iv) active line periods. Why are the front porch and back porch intervals provided before and after the horizontal sync pulse ? Explain why the blanking pulses are not used as sync pulses. Enumerate the basic requriments that must be satisfied by the pulse train added after each field. Why is it necessary to serrate the broad vertical sync pulse ? Sketch the pulse trains that follow after the second and first field of active scanning. Why are the vertical sync pulses notched at 32 µs interval and not at 64 µs interval to provide horizontal sync pulses ?

5. 6.

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7.

Explain how the horizontal and vertical sync pulses are separated and shaped at the receiver. For a time constant of 5 µs for the differentiating circuit, and 100 µs for the integrating circuit, plot the output waveforms from both the circuits for the entire vertical period. Calculate the error in timing for successive vertical fields in the absence of equalizing pulses. Sketch the complete pulse trains that follow at the end of both odd and even fields. Fully label them and explain how the half line discrepancy is removed by insertion of pre-equalizing pulses. Justify the need for pre and post equalizing pulses. Why it is necessary to keep their duration equal to the half-line period ?

8. 9.

10. Justify the need for a blanking period corresponding to 20 complete lines after each active field of scanning. Why does the vertical retrace not begin with the incoming of the first serrated vertical sync pulse ? 11. Sketch the complete pulse trains that follow at the end of odd and even fields in the 525 line television system. Justify the need for six instead of five pre and post equalizing pulses. 12. Show by any suitable means approximate correspondence between line numbers and the location of the electron beam on the screen, both for odd and even fields.

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4
Signal TransmiIIion and Channel Bandwidth

4
Signal Transmission and Channel Bandwidth
In most television systems as also in the C.C.I.R 625 line, the picture signal is amplitude modulated and sound signal frequency modulated before transmission. The channel bandwidth is determined by the highest video frequency required for proper picture reception and the maximum sound carrier frequency deviation permitted in a TV system. Need for modulation. The need for modulation stems from the fact that it is impossible to transmit a signal by itself. The greatest difficulty in the use of unmodulated wave is the need for long antennas for efficient radiation and reception. For example, a quarter-wavelength antenna for the transmitting frequency of 15 kHz would be 5000 meters long. A vertical antenna of this size is unthinkable and in fact impracticable. Another important reason for not transmitting signal frequencies directly is that both picture and sound signals from different stations are concentrated within the same range of frequencies. Therefore, radiation from different stations would be hopelessly and inextricably mixed up and it would be impossible to separate one from the other at the receiving end. Thus in order to be able to separate the intelligence from different stations, it is necessary to translate them all to different portions of the electromagnetic spectrum depending on the carrier frequency assigned to each station. This also overcomes the difficulties of poor radiation at low frequencies. Once signals are translated before transmission, a tuned circuit provided in the RF section of the receiver can be used to select the desired station.

4.1 AMPLITUDE MODULATION
In amplitude modulation the intelligence to be conveyed is used to vary the amplitude of the carrier wave. As an illustration, an amplitude modulated signal is shown in Fig. 4.1 (a) where ec = Ec cos ωct is the carrier wave and em = Em cos ωmt is the modulating signal. Note that the camera signal is actually complex in nature but a single modulating frequency has been chosen for convenience of analysis. The equation of the modulated wave is : e = A cos ωct where A = (Ec + kEm cos ωmt) when k is a constant of the modulator. 54

It may be noted that at kEm = Ec, m = 1 and the corresponding depth of modulation is then termed as 100%. Equation (4.1) may be expanded by the use of trigonometrical identities and expressed as :
mEc mEc cos (ωc – ωm) t – cos (ωc + ωm)t ...(4.2) 2 2 This result shows that if a carrier wave having frequency equal to fc is amplitude modulated with a single frequency fm, the resultant wave consists of the carrier (fc) and the sum and difference components (fc ± fm) of the carrier frequency and the modulating frequency. However, if the modulating signal consists of more than a single frequency, as it would be for a video signal, the equation can be extended to include the sum and difference of the carrier and all frequency components of the modulating signal. This is illustrated in Fig. 4.1 (b) where fm has been shown to be the highest modulating frequency. The region between fc and (fc + fm) is called the upper sideband (USB) and that between fc and (fc – fm) the lower sideband (LSB).

e = Ec cos ωct +

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Therefore if the modulated wave is to be transmitted without distortion by this method, the transmission channel must be atleast of width 2fm centred on fc.

4.2 CHANNEL BANDWIDTH
In the 625 line TV system where the frequency components present in the video signal extend from dc (zero Hz) to 5MHz, a double sideband AM transmission would occupy a total bandwidth of 10 MHz. The actual band space allocated to the television channel would have to be still greater, because with practical filter characteristics it is not possible to terminate the bandwidth of a signal abruptly at the edges of the sidebands. Therefore, an attenuation slope of 0.5 MHz is provided at each edge of the two sidebands. This adds 1 MHz to the required total band space. In addition to this, each television channel has its associated FM (frequency modulated) sound signal, the carrier frequency of which is situated just outside the upper limit of 5.5 MHz of the picture signal. This, together with a small guard band, adds another 0.25 MHz to the channel width, so that a practical figure for the channel bandwidth would be 11.25 MHz. This is illustrated in Fig. 4.2.
Total channel width = 11.25 MHz P 5.5 MHz 5.5 MHz Picture carrier Lower sideband (LSB) 5.5 5 4 3 2 1 0 Upper sideband (USB) 1 2 3 4 5 5.5

Such a bandwidth is too large, and if used, would limit the number of channels in a given high frequency spectrum allocated for TV transmission. Therefore, to ensure spectrum conservation, some saving in the bandwidth allotted to each channel is desirable. Single sideband transmission (SSB). A careful look at eqn. (4.2) reveals that the carrier component conveys no information because its amplitude and frequency remain constant no matter what the amplitude of the modulating voltage is. However, the presence of the carrier frequency is necessary at the receiver for recovering the modulating frequency fm, from the upper sideband by taking (fc + fm) – fc or from the lower sideband by taking fc – (fc – fm). Therefore, though superfluous from the point of view of transmission of intelligence, the carrier frequency is radiated along with the sideband components in all radio-broadcast and TV systems. Such an arrangement results in simpler transmitting equipment and needs a very simple and inexpensive diode detector at the receiver for recovering the modulation components without undue distortion. From eqn. (4.2) it is also obvious that the two sidebands are images of each other, since each is equally affected by changes in the modulating voltage amplitude via the component

SIGNAL TRANSMISSION AND CHANNEL BANDWIDTH

57

mEc . Also, any change in the frequency of the modulating signal results in identical changes 2
in the band spread of the two sidebands. It is seen, therefore, that all the information can be conveyed by the use of one sideband only and this results in a saving of 5 MHz per channel. It may, however, be noted that the magnitude of the detected signal in the receiver will be just half of that obtained when both the sidebands are transmitted. This is no serious drawback because the IF (intermediate frequency) amplifier stages of the receiver provide enough gain to develop reasonable amplitude of the video signal at the output of video detector.

4.3 VESTIGIAL SIDEBAND TRANSMISSION
In the video signal very low frequency modulating components exist along with the rest of the signal. These components give rise to sidebands very close to the carrier frequency which are difficult to remove by physically realizable filters. Thus it is not possible to go to the extreme and fully suppress one complete sideband in the case of television signals. The low video frequencies contain the most important information of the picture and any effort to completely suppress the lower sideband would result in objectionable phase distortion at these frequencies. This distortion will be seen by the eye as ‘smear’ in the reproduced picture. Therefore, as a compromise, only a part of the lower sideband, is suppressed, and the radiated signal then consists of a full upper sideband together with the carrier, and the vestige (remaining part) of the partially suppressed lower sideband. This pattern of transmission of the modulated signal is known as vestigial sideband or A5C transmission. In the 625 line system, frquencies up to 0.75 MHz in the lower sideband are fully radiated. The net result is a normal double sideband transmission for the lower video frequencies corresponding to the main body of picture information. As stated earlier, because of fillter design difficulties it is not possible to terminate the bandwidth of a signal abruptly at the edges of the sidebands. Therefore, an attenuation slope covering approximately 0.5 MHz is allowed at either end. Any distortion at the higher frequency end, if attenuation slope were not allowed, would mean a serious loss in horizontal detail, since the high frequency components of the video modulation determine the amount of horizontal detail in the picture. Fig. 4.3 illustrates the saving of band space which results from vestigial sideband transmission. The picture signal is seen to occupy a bandwidth of 6.75 MHz instead to 11 MHz.
4.25 MHz Saving in band space P 1.25 MHz

Total channel width = 7 MHz 5.5 MHz 0.5 MHz Guard edge S 0.25 MHz

Amplitude

Part of LSB removed by filter 5.5 5 4 2 1.25

LSB .75 0

0.75 MHz

Full USB f(MHz) 5.75

2

4

5.5

Fig. 4.3 Total channel bandwidth using vestigial’ lower sideband.

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MONOCHROME AND COLOUR TELEVISION

4.4 TRASMISSION EFFICIENCY
Though the total power that is developed and radiated at the transmitter has no direct bearing on bandwidth requirements, the saving in power that can be effected by suppressing the carrier and one of the sidebands cannot be totally ignored. This can be demonstrated by considering the power relations in the modulated wave. Based on eqn. (4.2) the total power Pt, in the modulated wave is the sum of the carrier power Pc, and the power in the two sidebands. This can be expressed as Pt = Pc + PUSB + PLSB = where
2 2 2 Ec m2 Ec m 2 Ec + + 2R 8R 8R

...(4.3)

Ec 2

is the r.m.s. value of the sinusoidal carrier wave, and R is the resistance in which

the power is dissipated. Equation (4.3) can be simplified to read as Pt = Pc +

m2 m2 m2 Pc = Pc 1 + Pc + 4 4 2

F GH

I JK

...(4.4)

Note from the above expression that Pc remains constant but Pt depends on the value of the modulation index m. Also note that when several frequency components of different amplitudes modulate the carrier wave, which in fact is the rule rather than an exception, the carrier power Pt is unaffected but the total sideband power gets distributed in the individual sideband component powers. This is so because the total modulating voltage is equal to the square root of the sum of the squares of individual modulating voltages. It can be seen from eqn. (4.4), that at 100% modulation (m = 1) the transmitted power attains its maximum possible value. Pt(max) = 1.5 Pc, where the power contained in the two sidebands has a maximum value of 50% of the carrier power. It is clear then, that the carrier component that is redundent, so far as the transmission of intelligence is concerned, constitutes about 72% of the total power that is radiated in the double sideband, full carrier (better known as A3 modulation) AM system. Therefore, a lot of economy can be effected if the carrier power is suppressed and not transmitted. Furthermore, suppression of one sideband results in more economy and also halves the bandwidth requirements for transmission as compared to A3. In practice SSB is used to save power and bandwidth in mobile communication systems, telemetry, radio navigation, military and several other such applications. However, such a system needs the generation of carrier frequency at the receiver for detection and this necessitates the transmission of a low level pilot carrier along with either of the two sidebands. In addition to this, a single sideband with suppressed carrier requires excellent frequency stability on the part of both transmitter and receiver. Any deviation in frequency and phase of the generated carrier at the receiver would severely impair the quality of the picture when used for television signal transmission. Such difficulties are not unsurmountable, but this tends to make the receiver circuitry more complicated, which in turn adds to the cost of the receiver. In point to point communication systems, where only one receiver is necessary, the additional expense is justifiable and infact SSB is now the accepted mode of communication for such applications. However, in television and radio broadcast systems, where a very large number of receivers simultaneously receive programme from one transmitter, additional cost of receivers is not justified and as such SSB cannot be recmmended. Therefore, as stated earlier, in all TV systems, full carrier is radiated and vestigial sideband transmission is used. In radio broadcast where the channel bandwidth is only 100 kHz, both the sidebands are transmitted along with full carrier.

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59

4.5 COMPLETE CHANNEL BANDWIDTH
The sound carrier is always positioned at the extremity of the fully radiated upper sideband and hence is 5.5 MHz away from the picture carrier. This is its logical place since it makes for minimum interference between the two signals. The FM sound signal occupies a frequency spectrum of about ± 75 KHz around the sound carrier. However, a guard band of 0.25 MHz is allowed on the sound carrier side of the television channel to allow for adequate inter-channel separation. The total channel bandwidth thus occupies 7 MHz and this represents a bandspace saving of 4.25 MHz per channel, when compared with the 11.25 MHz space, which would be required by the corresponding double sideband signal. Figure 4.4 show the complete channel. The frequency axis is scaled ralative to the picture carrier, which is marked as 0 MHz. This makes the diagram very informative, since details such as the widths of the upper and lower sidebands and the relative position of the sound carrier are easily read off.
Total channel width = 7 MHz 5.5 MHz
Amplitude

is allowed to be radiated. The total bandwidth per channel is 8 MHz. Fig. 4.5 (b) illustrates channel details of 525 line American system, where the highest allowed modulating frequency is 4 MHz with a total bandwidth of 6 MHz. In the French 819 line system where the highest modulating frequency is 10.4 MHz a channel bandwidth equal to 14 MHz is allowed. The diagram in Fig. 4.6 shows how two adjacent C.C.I.R. 625 line channels in the VHF Band-I are disposed one after the other.
P = 55.25 MHz S = 60.75 P = 62.25 MHz MHz 5.5 MHz Band I Channel III 54 55 56 57 58 59 60 61 62 63 5.5 MHz Band I Channel IV 64 65 66 67 68 f(MHz) S = 67.75 MHz

Amplitude

54 to 61 (7 MHz)

61 to 68 (7 MHz)

Fig. 4.6 Sideband spectrum of two adjacent channels of the lower VHF band of television station allocations.

4.6 RECEPTION OF VESTIGIAL SIDEBAND SIGNALS
In principle an SSB signal with carrier cannot be demodulated by an envelope detector. Either synchronous demodulation or a square law device to produce effective multiplication of the carrier with the sideband is required. However, it can be shown that if the sideband amplitude is small compared to the carrier, then the envelope of the SSB with carrier signal nearly corresponds to the modulating signal. In that case, envelope detection can be used and is the normal practice in television receivers. With vestigial sideband however, the relative amplitude of the frequencies for which both sidebands exist is double that of the true SSB component at the envelope detector output. In the video signal it would be so for the low frequency content of the picture signal, and in effect, amounts to distortion in terms of relative amplitudes for different frequencies and needs correction at the receiver. This when expressed in another way means that if the picture carrier were successively modulated to an equal depth by a series of frequencies throughout the video frequency range

SIGNAL TRANSMISSION AND CHANNEL BANDWIDTH

61

employed by the system, and the resulting voltage output from the detector recorded, the output voltage against input frequency characteristic obtained would have the form shown in Fig. 4.7. The vestigial sideband extends to 0.75 MHz below the carrier and thereafter this sideband is linearly attenuated down to zero at 1.25 MHz. The detector output voltage would thus be twice as great between 0 Hz and 0.75 MHz than between 1.25 MHz to 5 MHz.

Between 0.75 MHz and 1.25 MHz the output voltage would fall linearly following the sideband attenuation slope of the transmitter. To correct this discrepancy, it is necessary to so shape the receiver response curve, that the frequencies present ‘twice’ are afforded less amplification than those occurring in one sideband only. The desired response is shown in Fig. 4.8. The response curve is shaped to place the picture carrier half-way down the side corresponding to the suppressed sideband. The width of the sloping edge on which the carrier is positioned is twice the width of the vestigial sideband. To understand how this achieves the desired result, refer to Fig. 4.8 and consider the treatment afforded to various frequencies within the video bandwidth. Frequencies between 5 MHz and 0.75 MHz i.e., those present in the upper sideband only, are seen to give unit output. Next, consider a frequency component at 0.5 MHz. This is present in both the sidebands. The total detector output is again unity. The component in the lower sideband gives rise to an output of a volts, while that in the upper sideband gives rise to b volts. From the geometry of the figure we see that (a + b) = 1. As a further example consider the response at 0.7 MHz. This component in the vestigial sideband gives rise to an output = 0.08 V, whilst in the upper sideband, it gives rise to 0.92 V. Again the sum of the two is unity, so that the same output is achieved for frequencies between 0.75 MHz and 5 MHz. Note that at 0.75 MHz the output in the vestigial sideband is zero, and that in the upper sideband it is equal to one. The necessary correction detailed above is carried out at the Intermediate Frequency (IF) amplifier stages of the television receiver by suitably shaping the passband characteristics of the tuned amplifiers. This matter is fully dealt with in Chapter 8. Demerits of Vestigial Sideband Transmission (a) A small portion of the transmitter power is wasted in the vestigial sideband filters which remove the remaining lower sideband. (b) The attenuation slope of the receiver to correct the boost at lower video frequencies places the carrier at 50 per cent output voltage which amounts to introducing a loss of about 6 db in the signal to noise voltage ratio relative to what be available if double sideband transmission is used. (c) Some phase and amplitude distortion of the picture signal occurs despite careful filter design at the transmitter. Also, it is very difficult to tune IF stages of the receiver to correspond exactly with the ideal desired response as shown in Fig. 4.8 and this too introduces some phase and amplitude distortion. (d) More critical tuning at the receiver becomes necessary because for a given amount of local oscillator mismatch or drift after initial tuning, the degeneration of picture quality is less with wider lower sideband than with narrow lower sideband. In this respect the British 625 line system is superior because it allows 1.25 MHz unattenuated lower sideband transmission as compared to 0.75 MHz in most other systems. Despite these demerits of vestigial sideband transmission it is used in all television systems because of the large saving it effects in the bandwidth required for each channel.

4.7 FREQUENCY MODULATION
The sound signal is frequency modulated because of its inherent merits of interference-free reception. Here the amplitude of the modulated carrier remains constant, whereas its frequency is varied in accordance with variations in the modulating signal. The variation in carrier

SIGNAL TRANSMISSION AND CHANNEL BANDWIDTH

63

frequency is made proportional to the instantaneous value of the modulating voltage. The rate at which this frequency variation takes place is equal to the modulating frequency. It is assumed that the phase relations of a complex modulating signal will be preserved. However, for simplicity, it is again assumed that the modulating signal is sinusoidal. The situation is illustrated in Fig. 4.9 which shown the modulating voltage, and the resulting frequency modulated wave. Fig. 4.9 also shows the frequency variation with time, which is seen to be identical to the variations with time of the modulating voltage.
+v 0 –v Modulating signal + v¢ 0 – v¢ Frequency modulation (deviation in frequency greatly exaggerated) t t

+f d 0 –f Frequency vs time in FM t

Fig. 4.9 Basic FM modulation waveforms.

Analysis of Frequency-Modulated (FM) Wave. In order to understand clearly the meaning of instantaneous frequency fi and the associated instan-taneous angular velocity ωi = 2πfωi, the equation of an ac wave in the generalized form may first be written as : e = A sin φ(t) where e = instantaneous amplitude A = peak amplitude φ(t) = total angular displacement at time t. The instantaneous angular velocity ωt is, by definition, the instantaneous rate of change

dφ(t) of angular displacement φ(t). dt
Thus

ωt =

dφ(t) dt

...(4.5)

A sinusoidal wave of constant frequency say fc(ωc = 2πfc) is a special case of eqn. (4.5) and then φ(t) = ωct + θ where θ is the angular position at t = 0. Application of eqn. (4.5) yields the result

64 ωt =

MONOCHROME AND COLOUR TELEVISION

dφ(t) = ωc dt
...(4.6)

A frequency modulated wave with sinusoidal modulation can now be expressed as: ωi = ωc + 2π∆f cos ωmt where ωi = instantaneous angular velocity ωc = angular velocity of carrier wave (average angular velocity). ωm = 2π times the modulating frequency fm. ∆f = maximum deviation of instantaneous frequency from the average value. It may be emphasized that the frequency deviation ∆f is proportional to the peak amplitude (cos ωm t = ± 1) of the modulating signal and is independent of the modulating frequency. The equation of the FM wave can now be obtained by combining eqn. (4.5) and (4.6) to give the value of φ(t). The steps involved are as follows : ωi = Integration gives : φ(t) = ωc t +

dφ(t) = ωc + 2π∆f cos ωm t dt

FG 2π∆f IJ sin ω Hω K
m

mt

+θ

where the constant of integration θ defines the angular position at time t = 0. Substituting the above value of φ(t) into the generalized form e = A sin φ(t) yields : e = A sin ω c t +

FG H

2π∆f sin ω m t ωm

IJ K

...(4.7)

where for the sake of simplicity angle θ has been assumed to be equal to zero. Equation (4.7) is commonly written in the form e = A sin (ωct + mf sin ωmt) where mf is termed the ‘modulation index’ of the FM wave and is defined as : mf = modulation index = ...(4.8)

∆f frequency deviation = fm modulating frequency

It may be noted that for a given frequency deviation, the modulation index varies inversely as the modulating frequency. Also mf is defined only for sinusoidal modulation unlike m of AM which is defined for any modulating signal. Frequency Spectrum of the FM Wave. The eqn. (4.8) is of the form, sine of a sine, and can be expressed as : e = A [(sin ωc t cos (mf sin ωm t) + cos ωc t sin (mf sin ωm t)] The term cos (mf sin ωm t) can be expanded into J0(mf) + 2J2(mf) cos 2ωm t + 2J4(mf) cos 4ωm t + ...... ...(4.9)

*Theory of Bessel’s functions is not necessary for us. Tabulated values of Bessel’s function are widely available.

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MONOCHROME AND COLOUR TELEVISION

(f) As seen in the table when the modulation index (mf) is less than 0.5, i.e., when the frequency deviation is less than half the mdoulating frequency, the second and higherorder sideband components are relatively small and the frequency band required to acommodate the essential part of the signal is the same as in amplitude modulation. On the other hand when mf exceeds unity, there are important higher-order sideband components contained in the wave and this results in increased bandwidth requirements. (g) The modulation index actually depends on both the amplitude and frequency of the modulating tone. It is higher in FM systems that permit large frequency deviation for a maximum amplitude of the modulating tone. This is turn results in higher order significant J coefficients and a larger bandwidth is required for reasonably distortion free transmission. (h) Since a lot of the higher sidebands have insignificant relative amplitudes, their exclusion will not distort the modulated wave unduly, and while calculating channel bandwidth J coefficients having values less than 0.05 for a calculated value of mf can be neglected.

4.8 FM CHANNEL BANDWIDTH
Based on the above discussion the channel bandwidth BW = 2nfm ...(4.11) where fm is the frequency of the modulating wave and n is the number of the significant sidefrequency components. The value of n is determined from the modulation index. Though the higher frequencies in speech or music have much less amplitude as compared to lower audio frequencies, we shall estimate the channel bandwidth for the worst case where even the highest frequency to be transmitted causes maximum permitted frequency deviation. The maximum frequency deviation of commercial FM is limited to 75 kHz, and the modulating frequencies typically cover 25 Hz to 15 kHz. If a 15 kHz tone has unit amplitude, i.e., equal to the maximum allowed amplitude, then mf =

i.e., n = 7. Therefore Had the amplitude been less, the maximum frequency deviation would not be developed, and the bandwidth would be smaller. This brings out an interesting observation that in frequency modulation (with fixed ∆f) the bandwidth depends on the tone amplitude, whereas in amplitude modulation, the bandwidth depends on the tone frequency. Similarly in the 625-B television system where the standards specify that the maximum deviation (∆f) should not exceed ± 50 kHz for the highest modulating frequency of 15 kHz, mf =

50 ≈3 15

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67

This gives a value of n = 5 as seen in the given chart. ∴ BW = 2 × 5 × 15 = 150 kHz. The bandwidth can also be estimated from ‘Carson’s Rule’ which states that to a good approximation, the bandwidth required to pass an FM wave is equal to twice the sum of the deviation and the highest modulating frequency. Thus, for the standard FM transmission the required bandwidth = 2(75 + 15) = 180 kHz. This nearly checks with the value of *210 kHz estimated earlier. Similarly for the 625 line system the Carson’s Rule yields a bandwidth requirement of 2(50 + 15) = 130 kHz and this is close to the value calculated earlier. The resultant deviation of ± 75 kHz around the sound carrier is very much within the guard-band edge and reasonably away from any significant video sideband components. It may be noted that in the American television system where the maximum permissible deviation is ± 25 kHz around the sound carrier, a bandwidth of about 100 kHz is enough for sound signal transmission.

4.9 CHANNEL BANDWIDTH FOR COLOUR TRANSMISSION
As explained in the chapter devoted to the analysis and synthesis of TV pictures the colour video signal does not extend beyond about 1.5 MHz. Therefore, the colour information can be transmitted with a restricted bandwidth much less than 5 MHz. This feature allows the narrow band chrominance (colour) signal to be multiplexed with the wideband luminance (brightness) signal in the standard 7 MHz television channel. This is achieved by modulating the colour signal with a carrier frequency which lies within the normal channel bandwidth. This is called colour subcarrier frequency and is located towards the upper edge of the video frequencies to avoid interference with the monochrome signal.
7 MHz P C S Sound signal spectrum

In the PAL colour system which is compatible with the C.C.I.R. 625 line monochrome system the colour subcarrier frequency is located 4.433 MHz way from the picture carrier. The
*In commercial FM broadcast a frequency spectrum of 200 kHz is allotted for each channel.

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MONOCHROME AND COLOUR TELEVISION

bandwidth of colour signals is restricted to about ± 1.2 MHz around the subcarrier. Fig. 4.10 gives necessary details of the location of monochrome (picture), colour and sound signal spectrums all within the same channel bandwidth of 7 MHz. It may be noted that in the American television system where the channel bandwidth is 6 MHz, the colour subcarrier is located 3.58 MHz away from the picture carrier.

4.10 ALLOCATION OF FREQUENCY BANDS FOR TELEVISION SIGNAL TRANSMISSION
For effective amplitude modulation and better selectivity at the RF and IF tuned amplifiers in the receiver, it is essential that the carrier frequency be chosen about ten times that of the highest modulating frequency. Since the highest modulating frequency for picture signal transmission is 5 MHz, the minimum carrier frequency that can be employed, cannot be much less than 40 MHz. As an illustration consider a carrier frequency fc = 10 MHz. With the highest video modulating frequency = 5 MHz, a deviation of 50 per cent from the centre frequency would be necessary in any tuned circuit to accommodate the lower and upper sideband frequencies. However, if the carrier frequency is fixed at, say 50 MHz, the percentage deviation required to pass the upper and lower sideband frequencies for the same modulating frequency would be only 10 per cent. It is obvious from these observations that selectivity is bound to be poor at the receiver tuned amplifiers with a carrier frequency of 10 MHz. The 3 db down points with a carrier frequency of 50 MHz are within 5 per cent deviation from the carrier frequency and thus the selectivity is bound to be much better. Further, each television channel occupies about 7 MHz. In order to accommodate several TV channels, the carrier frequencies have to be in the region of the spectrum above about 40 MHz. This explains why television transmission has to be carried out at very high frequencies in the VHF and UHF bands. In radio broadcast where the highest modulating frequency is only 5 kHz, lower carrier frequencies can be used, and accordingly transmission is carried out in the medium wave band (550 kHz to 1600 kHz) and short wave bands extending up to about 30 MHz. Transmission at very high frequencies has its own problems and limitations for long distance transmission and these are discussed in another chapter.

4.11 TELEVISION STANDARDS
After having learnt about various aspects of television transmission and reception, it would be instructive to review, in detail, the picture and sound signal standards as specified by the International Radio Consulative Committee (C.C.I.R) for the 625-B monochrome system and also to compare its main characteristics with that of other principal television system. This is detailed in Tables 2 and 3.

5 such pulse, each pulsewidth Interval between field sync pulses Interval between equalizing pulses Post-sync equalizing pulses, 5 pulses Build up time of field blanking edges. Build up time for field sync pulses

(i) England earlier used 405 line system in the 5 MHz channel. (ii) France earlier used 819 line system with a channel bandwidth of 14 MHz.

Review Questions
1. 2. Why is it necessary to mdoulate the picture and sound signals before transmission ? Why is TV transmission carried out in the UHF and VHF bands ? Show that in the 625-B system, a total channel bandwidth of 11.25 MHz would be necessary if both the sidebands of the amplitude mdoulated picture signal are fully radiated along with the frequency modulated picture signal. Why is an attenuation slope of 0.5 MHz allowed at both the edges of the AM picture signal sidebands ? Why is a guard band provided at the sound signal edge of the television channel ? Why is it necessary to affect economy in channel bandwidth ? Why SSB is not used for picture signal transmission ? What is vestigial sideband transmission and why it is used for transmission of TV picture signals ? Why is a portion of the lower sideband of the AM picture signal transmitted along with the carrier and full USB ? Does it need any correction somewhere in the television link ? If so where is it carried out ? Sketch and fully label the desired response of a TV receiver that includes necessary correction on account of the discrepancy caused by VSB transmission. Comment on the response curve drawn by you. Show that a total channel bandwidth of 7 MHz is necessary for successful transmission of both picture and sound signals in the 625 line TV system. Sketch frequency distribution of the channel and mark the location of picture and sound signal carrier frequencies. Why is the sound carrier located 5.5 MHz away from the picture carrier ? Justify the allocation of 8 MHz in the British TV system and 6 MHz in the American system for each TV channel. What is the separation between picture and sound carriers in each of these systems ?

3. 4. 5. 6.

7.

8.

9.

10. What is ‘modulation index’ in FM transmission and how does it affect the bandwidth required for each FM channel ? 11. Explain how you would proceed to determine the channel bandwidth for transmission of sound signals (highest modulating frequency = 15 kHz) by frequency modulation. How does the permitted maximum deviation affect the bandwidth requirements ? 12. Show that in the 625-B system where the maximum allowed frequency deviation is ± 50 kHz, a bandwidth of 150 kHz is necessary for almost distortion free transmission by frequency modulation, the highest modulating frequency being 15 kHz, Repeat this for the American system where the maximum allowed deviation is ± 25 kHz. Verify the results by ‘Carson’s Rule’ of determining channel bandwidth.

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5
The Picture Tube

5
The Picture Tube
The picture tube or ‘kinescope’ that serves as the screen for a television receiver is a specialized from of cathode-ray tube. It consists of an evacuated glass bulb or envelope, inside the neck of which is rigidly supported an electron gun that supplies the electron beam. A luminescent phosphor coating provided on the inner surface of its face plate produces light when hit by the electrons of the fast moving beam. A monochrome picture tube has one electron gun and a continuous phosphor coating that produces a picture in black and white. For colour picture tubes the screen is formed of three different phosphors and there are three electron beams, one for each colour phosphor. The three colours—red, green and blue produced by three phosphors combine to produce different colours. More details of colour picture tubes are given in chapters devoted to colour television.

5.1

MONOCHROME PICTURE TUBE

Modern monochrome picture tubes employ electrostatic focussing and electromagnetic deflection. A typical black and white picture tube is shown in Fig. 5.1. The deflection coils are

Neck Bell

Tension band

Screen (face plate) Envelope or bulb

Fig. 5.1. A rectangular picture tube.

mounted externally in a specially designed yoke that is fixed close to the neck of the tube. The coils when fed simultaneously with vertical and horizontal scanning currents deflect the beam at a fast rate to produce the raster. The composite video signal that is injected either at the 74

THE PICTURE TUBE

75

grid or cathode of the tube, modulates the electron beam to produce brightness variations of the tube, modulates the electron beam to produce brightness variations on the screen. This results in reconstruction of the picture on the raster, bit by bit, as a function of time. However, the information thus obtained on the screen is perceived by the eye as a complete and continuous scene because of the rapid rate of scanning. Electron Gun The various electrodes that constitute the electron gun are shown in Fig. 5.2. The cathode is indirectly heated and consists of a cylinder of nickel that is coated at its end with thoriated tungsten or barium and strontium oxides. These emitting materials have low work-function
External conductive coating

and when heated permit release of sufficient electrons to form the necessary stream of electrons within the tube. The control grid (Grid No. 1) is maintained at a negative potential with respect to cathode and controls the flow of electrons from the cathode. However, instead of a wiremesh structure, as in a conventional amplifier tube, it is a cylinder with a small circular opening to confine the electron stream to a small area. The grids that follow the control grid are the accelerating or screen grid (Grid No. 2) and the focusing grid (Grid No. 3). These are maintained at different positive potentials with respect to the cathode that vary between + 200 V to + 600 V. All the elements of the electron gun are connected to the base pins and receive their rated voltages from the tube socket that is wired to the various sections of the receiver. Electrostatic Focussing The electric field due to the positive potential at the accelerating grid (also known as 1st anode) extends through the opening of the control grid right to the cathode surface. The orientation of

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MONOCHROME AND COLOUR TELEVISION

this field is such that besides accelerating the electrons down the tube, it also brings all the electrons in the stream into a tiny spot called the crossover. This is known as the first electrostatic lens action. The resultant convergence of the beam is shown in Fig. 5.2. The second lens system that consists of the screen grid and focus electrode draws electrons from the crossover point and brings them to a focus at the viewing screen. The focus anode is larger in diameter and is operated at a higher potential than the first anode. The resulting field configuration between the two anodes is such that the electrons leaving the crossover point at various angles are subjected to both convergent and divergent forces as they more along the axis of the tube. This in turn alters the path of the electrons in such a way that they meet at another point on the axis. The electrode voltages are so chosen or the electric field is so varied that the second point where all the electrons get focused is the screen of the picture tube. Electrostatic focusing is preferred over magnetic focusing because it is not affected very much by changes in the line voltage and needs no ion-spot correction. Beam Velocity In order to give the electron stream sufficient velocity to reach the screen material with proper energy to cause it to fluoresce, a second anode is included within the tube. This is a conductive coating with colloidal graphite on the inside of the wide bell of the tube. This coating, called aquadag, usually extends from almost half-way into the narrow neck to within 3 cm of the fluorescent screen as shown in Fig. 5.2. It is connected through a specially provided pin at the top or side of the glass bell to a very high potential of over 15 kV. The exact voltage depends on the tube size and is about 18 kV for a 48 cm monochrome tube. The electrons that get accelerated under the influence of the high voltage anode area, attain very high velocities before they hit the screen. Most of these electrons go straight and are not collected by the positive coating because its circular structure provides a symmetrical accelerating field around all sides of the beam. The kinetic energy gained by the electrons while in motion is delivered to the atoms of the phosphor coating when the beam hits the screen. This energy is actually gained by the outer valence electrons of the atoms and they move to higher energy levels. While recturning to their original levels they give out energy in the form of electromagnetic radiation, the frequency of which lies in the spectral region and is thus perceived by the eye as spots of light of varying intensity depending on the strength of the electron beam bombarding the screen. Because of very high velocities of the electrons which hit the screen, secondary emission takes place. If these secondary emitted electrons are not collected, a negative space charge gets formed near the screen which prevents the primary beam from arriving at the screen. The conductive coating being at a very high positive potential collects the secondary emitted electrons and thus serves the dual purpose of increasing the beam velocity and removing unwanted secondary electrons. The path of the electron current flow is thus from cathode to screen, to the conductive coating through the secondary emitted electrons and back to the cathode through the high voltage supply. A typical value of beam current is about 0.6 mA with 20 kV applied at the aquadag coating.

5.2

BEAM DEFLECTION

Both electric and magnetic fields can be employed for deflecting the electron beam. However, in television picture tubes electromagnetic deflection is preferred for the following reasons :

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(a) As already stated the electron beam must attain a very high velocity to deliver enough energy to the atoms of the phosphor coating. Because of this the electrons of the beam remain under the influence of the deflecting field for a very short time. This necessitates application of high deflecting fields to achieve the desired deflection. For example with an anode voltage of about 1 kV, as would be the case in most oscilloscopes, some 10 V would be necessary for 1 cm deflection of the beam on the screen, whereas in a picture tube with 15 kV at the final anode, about 7500 V would be necessary to get full deflection on a 50 cm screen. It is very difficult to generate such high voltages at the deflection frequencies. On the other hand with magnetic deflection it is a large current that would be necessary to achieve the same deflection. Since it is more convenient to generate large currents than high voltages, all picture tubes employ electromagnetic deflection. (b) With electrostatic deflection the beam electrons gain energy. Thus larger deflection angles tend to defocus the beam. Further, the deflection plates need to be placed further apart as the deflection angle is made larger, thus requiring higher voltages to produce the same deflection field. Magnetic deflection is free from both these shortcomings and much larger deflection angles can be achieved without defocusing or nonlinearities with consequent saving in tube length and cabinet size. (c) For electrostatic deflection two delicate pairs of deflecting plates, are needed inside the picture tube, whereas for magnetic deflection two pairs of deflecting coils are mounted outside and close to the neck of the tube. Such a provision is economical and somewhat more rugged. Deflection Yoke The physical placement of the two pairs of coils around the neck of the picture tube is illustrated in Fig. 5.3 and the orientation of the magnetic fields produced by them is shown in Fig. 5.4. In combination, the vertical and horizontal deflection coils are called the ‘Yoke’. This yoke is fixed outside and close to the neck of the tube just before it begins to flare out (see Fig. 5.2).

Vertical raster plane Horizontal deflection windings Top

Vertical deflection windings

Electron beam H

Horz raster plane

V Bottom

Fig. 5.3. Cross-sectional view of a yoke showing location of vertical and horizontal deflection windings about the neck of the picture tube.

Fig. 5.4. Horizontal and Vertical deflecting coils (pairs) around the neck of the picture tube. Note that the location of the beam on the picture tube screen will depend on the strength and direction of currents in the two pairs of coils. For the directions of current shown the beam will be deflected upwards and to the left.

The magnetic field of the coils reacts with the electron beam to cause its deflection. The horizontal deflection coil which sweeps the beam across the face of the tube from left to right is split into two sections and mounted above and below the beam axis. The vertical deflection coil is also split into two sections and placed left and right on the neck in order to pull the beam gradually downward as the horizontal coils sweep the beam across the tube face. Each coil gets its respective sweep input from the associated sweep circuits, and together they form the raster upon which the picture information is traced. It may be noted that a perpendicular displacement results because the magnetic field due to each coil reacts with the magnetic field of the electron beam to produce a force that deflects the electrons at right angles to both the beam axis and the deflection field. Deflection Angle This is the maximum angle through which the beam can be deflected without striking the side of the bulb. Typical values of deflection angles are 70°, 90°, 110° and 114°. As shown in Fig. 5.5, it is the total angle that is specified. For instance a deflection angle of 110° means the electron beam can be deflected 55° from the centre. The advantage of a large deflection angle is that for equal picture size the picture tube is shorter and can be installed in a smaller cabinet. However, a large deflection angle requires more power from the deflection circuits. For this reason the tubes are made with a narrow neck to put the deflection yoke closer to the electron beam. A 110° yoke has a smaller hole diameter (about 3 cm) compared with neck diameters for tubes with lesser deflection angles. Different screen sizes can be filled with the same deflection angle, because bigger tubes have larger axial lengths.

Cosine Winding With increased deflection angles it becomes necessary to use a special type of winding to generate uniform magnetic fields for linear deflection. In this arrangement the thickness of the deflection winding varies as the cosine of the angle from a central reference line. Such a winding is known as ‘Cosine winding’ and its appearance in a deflection yoke is shown in Fig. 5.3. Nearly all present day yokes are wound in this manner to ensure linear deflection.

5.3

SCREEN PHOSPHOR

The phosphor chemicals are generally light metals such as zinc and cadmium in the form of sulphide, sulphate, and phosphate compounds. This material is processed to produce very fine particles which are then applied on the inside of the glass plate. As already explained the high velocity ellectrons of the beam on hitting the phosphor excite its atoms with the result that the corresponding spot fluoresces and emits light. The phosphorescent characteristics of the chemicals used are such that an afterglow remains on the screen for a short time after the beam moves away from any screen spot. This afterglow is known as persistence. Medium persistence is desirable to increase the average brightness and to reduce flicker. However, the persistence must be less than 1/25 second for picture tube screens so that one frame does not persist into the next and cause blurring of objects in motion. The decay time of picture tube phosphors is approximately 5 ms, and its persistence is referred to as P4 by the industry.

5.4

FACE PLATE

A rectangular image on a circular screen is wasteful of screen area. Therefore, all present day picture tubes have rectangular face plates, with a breadth to height ratio of 4 : 3. A rectangular tube with 54 cm screen means that the distance between the two diagonal points is 54 centimeters. Approximately 1.5 cm thickness provides the strength required for the large face plate to withstand the air pressure on the evacuated glass envelope. In older receivers special glass or plastic shields were placed in the cabinet in front of the picture tube to prevent any glass from hitting the viewer in case of an implosion. Modern picture tubes incorporate integral

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implosion protection. There are a number of different systems in use. In one arrangement, known as kimcode, a metal rim band (see Fig. 5.1) is held around the tube by a tension strap or with a layer of epoxy cement. In another system called Panoply, a special faceplate is held in front of the tube by epoxy cement. In all cases it is essential to check for implosion proofing while replacing any picture tube. Yoke and Centering Magnets The yoke on all black and white tubes is positioned right up against the flare of the tube in order to achieve complete coverge of the full screen area. If the yoke is not moved as far forward as possible, the electron beam will strike the neck of the picture tube and cause a shadow near the corners of the face plate. The mounting system permits positioning of the yoke against the tube funnel and allows rotation of the yoke to ensure that horizontal lines run parallel to the natural horizontal axis. Electrical centering of the beam can be accomplished by supplying direct current through the horizontal and vertical deflection coils. However, this method is not used now because of the added current drain on the low voltage power supply. Modern tubes have a pair of permanent magnets (see Fig. 5.2) for centering, in the form of rings usually mounted on the yoke cover. Poles of both the magnets can be suitably shifted with a pair of projecting tabs provided on the magnetic rings. When the two tabs (one from each ring) coincide with each other, the strongest field is achieved; that is, the beam will be pushed furthest off centre. When the two tabs are 180° apart (on opposite sides) the field is minimum and so is the decentering. The two rings are rotated together to change the direction in which decentering occurs. This is illustrated in Fig. 5.6.
Yoke frame Windings terminal board Movable centering magnets (two) Horz deflection windings

The edge of the yoke linear (see Fig. 5.2) is used to hold small permanent magnets. As shown in Fig. 5.6 these are positioned to correct any ‘pincushion error’. Screen Brightness It is estimated that about 50 per cent of the light emitted at the screen, when the electron beam strikes it, travels back into the tube. Another 20 percent or so is lost in the glass of the

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tube because of internal reflections and only about 20 percent of it reaches the viewer. Image contrast is also impaired because of interference caused by the light which is returned to the screen after reflection from some other points. Also any ions is the beam, which do exist despite best precautions while degassing, damage the phosphor material on hitting it and thus cause a dark brownish patch on the screen. This area usually centers around the middle of the screen because the greater mass of the ions prevents any appreciable deflection during their transit, with the result, that they arrive almost at the centre of the screen. To overcome these serious drawbacks practically all modern picture tubes employ a very thin coating of aluminium on the back surface of the screen phosphor. The aluminized coating is very thin and with a final anode voltages of 10 kV or more, the electrons of the beam have enough velocity to penetrate this coating and excite the phosphor. Thus most of the light that would normally travel back and get lost in the tube is now reflected back to the screen by the metal backing and this results in a much improved brilliancy. The aluminized coating is connected to the high voltage anode coating and thus helps in draining off the secondary emitted electrons at the screen. This further improves the brightness. Ion-trap In older picture tubes a magnetic beam, bender commonly known as ‘ion-trap’ was employed to deflect the heavy ions away from the screen. In present day picture tubes having a thin metal coating on the screen, it is no longer necessary to provide an ion-trap. This is because the ions on account of their greater mass fail to penetrate the metal backing and do not reach the phosphor screen. Thus an aluminized coating when provided on the phosphor screen, not only improves screen brightness and contrast but also makes the use of ‘ion-traps’ unnecessary. High Voltage Filter Capacitor A grounded coating is provided on the outer surface of the picture tube. This provides shielding from stray fields and also acts as one plate of the capacitor, the other plate being the inner anode coating with the glass bulb serving as the insulator between the two. The capacitor thus formed (see Fig. 5.2) serves as a filter capacitor for the high voltage supply. This capacitor can hold charge for a long time after the anode voltage is switched off and so before handling the picture tube the capacitor must be discharged by shorting the anode button to the grounded wall coating. Spark-gap Protection On account of close spacing between the various-electrodes and the use of very high voltages, arcing of flashover can occur in the electron gum especially at the control grid. This arcing causes voltage surges, which result in damage to the associated circuit components. Therefore for protection of the receiver circuit, due to any arcing, metallic spark-gaps are provided as shunt paths for the surge currents. In some designs neon bulbs are used as spark gaps. The gas in the neon tube ionizes when the potential exceeds a certain limit and thus provides a shunt path for the high voltage arc current.

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5.5

PICTURE TUBE CHARACTERISTICS

As shown in Fig. 5.7 the transfer characteristics of picture tubes are similar to the grid-plate characteristics of vacuum tubes. The grid of the picture tube has a fixed bias that is set with the brightness control for optimum average brightness of the picture on the screen. The video signal that finally controls the brightness variations on the screen may be applied either at the grid or cathode of the picture tube. Each method has its own merits and demerits and are discussed in another chapter. This method of varying the beam current to control the instantaneous screen brightness is called intensity or ‘Z’ axis modulation. The peak-to-peak amplitude of the ac video signal determines the contrast in the picture, between peak white with maximum beam current and black at cut-off. The contrast control is in the video amplifier, which controls the peak-to-peak amplitude of the video signal applied to the picture tube.
Anode current mA Peak white 1.6 1.2 0.8 Cut-off bias (black) 0.4 VG1K 0

At cut-off the grid voltage is negative enough to reduce the beam current to a value low enough to extinguish the beam, and this corresponds to the black level in the picture. The parts of the screen without any luminescence look black in comparison with the adjacent white areas.

5.6

PICTURE TUBE CIRCUIT CONTROLS

Manufacturers usually recommend a sufficiently high voltage to the second anode of the picture tube to produce adequate screen brilliancy for normal viewing. This voltage is always obtained from the output of the horizontal deflection circuit. The dc voltages to the screen grid and focus

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grid are also taken from the horizontal stage and adjusted to suitable values by resistive potential divider networks. This is shown in Fig. 5.8.
+ Vcc

Contrast control + + 300 V F F VG1K = – 50 K G1

V VH H

G2 G 3

Picture tube

From video detector

+ 600V R1 R2

Video amplifier

+ 300 V Brightness control R4

R3 EHT 2KV

(Boosted B + Supply)

Fig. 5.8. Picture tube circuit and associated controls.

A variable bias control either in the cathode circuit or control grid lead is provided to control the electron density, which in turn controls the brightness on the screen. This control, known as the ‘brightness control’, is brought out at the front panel of the receiver to enable the viewer to adjust brightness. As discussed earlier most modern picture tubes do not require critical focus adjustment. Therefore no focus control is normally provided and instead dc voltage at the focus electrode is carefully set as explained above. The contrast control through not strictly a part of the picture tube circuit forms part of cathode or control grid circuit. This control is also provided at the front panel of the receiver and its variation enables adjustment of contrast in the reproduced picture. Picture Tube Handling The very high vacuum in a modern picture tube means that there is a danger of implosion if the tube is struck with a hard object or if it is made to rest on its neck. Because of the large volume of the tube, there is a very high pressure on the glass shell. In case it breaks the resulting implosion will often cause tube fragments to fly in all directions at high speed. This may cause severe injury to the persons hit by the tube fragments. Manufacturers recommend the use of protective goggles and gloves whenever picture tubes are handled and such precautions should be observed. The tube neck is particularly fragile and must be handled with care.

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Review Questions
1. 2. 3. 4. 5. 6. 7. 8. 9. Sketch the sectional view of a picture tube that employs electrostatic focusing and electromagnetic deflection and label all the electrodes. Explain briefly, how the electron beam is focused on the tube screen. What is meant by crossover point in the electron gun ? What type of phosphor is employed for picture tube screens ? Why is a medium persistence phosphor preferred ? What is the function of aquadag coating on the inner side of the tube bell ? Why is a grounded coating provided on the outer surface of the picture tube ? Why is an aluminized coating provided on the phosphor screen ? How are any stary ions prevented from hitting the screen ? What do you understand by a 54 cm picture tube ? Why is it necessary to employ implosion protection in picture tubes ? What precautions must be observed while handling a picture tube ? Why is it necessary to provide spark-gap protection between the various electrodes ? Discuss the merits of electromagnetic deflection over electrostatic deflection in television picture tubes. Why is ‘cosine winding’ used for deflection coils ? What is meant by the deflection angle of a picture tube ? What is the advantage of providing a large deflection angle yoke ?

10. Explain how the yoke is mounted on the tube neck. Describe how the centering of the electron beam is accomplished with the help of centering magnets. Why are small permanent magnets provided at the edges of the yoke liner ? 11. Show with a circuit diagram how dc potentials are supplied to the various electrodes of the picture tube. 12. What are the functions of ‘brightness’ and ‘contrast’ controls ? Explain their action with suitable circuit diagrams.

6
Television Camera Tubes

6
Television Camera Tubes
A TV camera tube may be called the eye of a TV system. For such an analogy to be correct the tube must possess characteristic that are similar to its human counterpart. Some of the more important functions must be (i) sensitivity to visible light, (ii) wide dynamic range with respect to light intensity, and (iii) ability to resolve details while viewing a multielement scene. During the development of television, the limiting factor on the ultimate performance had always been the optical-electrical conversion device, i.e., the pick-up tube. Most types developed have suffered to a greater or lesser extent from (i) poor sensitivity, (ii) poor resolution, (iii) high noise level, (iv) undesirable spectral response, (v) instability, (vi) poor contrast range and (vii) difficulties of processing. However, development work during the past fifty years or so, has enabled scientists and engineers to develop image pick-up tubes, which not only meet the desired requirements but infact excel the human eye in certain respects. Such sensitive tubes have now been developed which deliver output even where our eyes see complete darkness. Spectral response has been so perfected, that pick-up outside the visible range (in infra-red and ultraviolet regions) has become possible. Infact, now there is a tube available for any special application.

6.1

BASIC PRINCIPLE

When minute details of a picture are taken into account, any picture appears to be composed of small elementary areas of light or shade, which are known as picture elements. The elements thus contain the visual image of the scene. The purpose of a TV pick-up tube is to sense each element independently and develop a signal in electrical form proportional to the brightness of each element. As already explained in Chapter 1, light from the scene is focused on a photosensitive surface known as the image plate, and the optical image thus formed with a lens system represents light intensity variations of the scene. The photoelectric properties of the image plate then convert different light intensities into corresponding electrical variations. In addition to this photoelectric conversion whereby the optical information is transduced to electrical charge distribution on the photosensitive image plate, it is necessary to pick-up this information as fast as possible. Since simultaneous pick-up is not possible, scanning by an electron beam is resorted to. The electron beam moves across the image plate line by line, and field by field to provide signal variations in a successive order. This scanning process divides the image into its basic picture elements. Through the entire image plate is photoelectric, its construction isolates the picture elements so that each discrete small area can produce its own signal variations. 86

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Photoelectric Effects The two photoelectric effects used for converting variations of light intensity into electrical variations are (i) photoemission and (ii) photoconductivity. Certain metals emit electrons when light falls on their surface. These emitted electrons are called photoelectrons and the emitting surface a photocathode. Light consists of small bundles of energy called photons. When light is made incident on a photocathode, the photons give away their energy to the outer valence electrons to allow them to overcome the potential-energy barrier at the surface. The number of electrons which can overcome the potential barrier and get emitted, depends on the light intensity. Alkali metals are used as photocathode because they have very low work-function. Cesium-silver or bismuth-silver-cesium oxides are preferred as photoemissive surfaces because they are sensitive to incandescent light and have spectral response very close to the human eye. The second method of producing an electrical image is by photoconduction, where the conductivity or resistivity of the photosensitive surface varies in proportion to the intensity of light focused on it. In general the semiconductor metals including sel nium, tellurium and lead with their oxides have this property known as photoconductivity. The variations in resistance at each point across the surface of the material is utilized to develop a varying signal by scanning it uniformly with an electron beam. Image Storage Principle Television cameras developed during the initial stages of development were of the non-storage type, where the signal output from the camera for the light on each picture element is produced only at the instant it is scanned. Most of the illumination is wasted. Since the effect of light on the image plate cannot be stored, any instantaneous pick-up has low sensitivity. Image disector and flying-spot camera are examples of non-storage type of tubes. These are no longer in use and will not be discussed. High camera sensitivity is necessary to televise scenes at low light levels and to achieve this, storage type tubes have been developed. In storage type camera tubes the effect of illumination on every picture element is allowed to accumulate between the times it is scanned in successive frames. With light storage tubes the amount of photoelectric signal an be increased 10,000 times approximately compared with the earlier non-storage type. The Electron Scanning Beam As in the case of picture tubes an electron gun produces a narrow beam of electrons for scanning. In camera tubes magnetic focusing is normally employed. The electrons must be focused to a very narrow and thin beam because this is what determines the resolving capability of the camera. The diameter of the beam determines the size of the smallest picture element and hence the finest detail of the scene to which it can be resolved. Any movement of electric charge is a flow of current and thus the electron beam constitutes a very small current which leaves the cathode in the electron gun and scans the target plate. The scanning is done by deflecting the beam with the help of magnetic fields produced by horizontal and vertical coils in the deflection yoke put around the tubes. The beam scans 312.5 lines per field and 50 such fields are scanned per second.

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In tubes employing photoemissive target plates the electron beam deposits some charge on the target plate, which is proportional to the light intensity variations in the scene being televised. The beam motion is so controlled by electric and magnetic fields, that it is decelerated before it reaches the target and lands on it with almost zero velocity to avoid any secondary emission. Because of the negative acceleration the beam is made to move back from the target and on its return journey, which is very accurately controlled by the focusing and deflection coils, it strikes an electrode which is located very close to the cathode from where it started. The number of electrons in the returning beam will thus vary in accordance with the charge deposited on the target plate. This in turn implies that the current which enters the collecting electrode varies in amplitude and represents brightness variations of the picture. This current is finally made to flow through a resistance and the varying voltage developed across this resistance constitutes the video signal. Figure 6.1 (a) illustrates the essentials of this technique of developing video signal.
Faceplate Light image Electron image v0 RL Camera lens + Photoemissive coating Light image + Faceplate Target Scanning beam Photoconductive coating Electron gun

v0 RL

Fig. 6.1(a). Production of video signal by photoemission.

Fig. 6.1(b). Production of video signal by photoconduction.

In camera tubes employing photoconductive cathodes the scanning electron beam causes a flow of current through the photoconductive material. The amplitude of this current varies in accordance with the resistance offered by the surface at different points. Since the conductivity of the material varies in accordance with the light falling on it, the magnitude of the current represents the brightness variations of the scene. This varying current completes its path under the influence of an applied dc voltage through a load resistance connected in series with path of the current. The instantaneous voltage developed across the load resistance is the video signal which, after due amplification and processing is amplitude modulated and transmitted. Figure 6.1 (b) shows a simplified illustration of this method of developing video signal. Electron Multiplier When the surface of a metal is bombarded by incident electrons having high velocities, secondary emission takes place. Aluminium, as an example, can release several secondary electrons for each incident primary electron. Camera tubes often include an electron multiplier structure, making use of the secondary emission effect to amplify the small amount of photoelectric current

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that is later employed to develop video signal. The electron multiplier is a series of cold anodecathode electrodes called dynodes mounted internally, with each at a progressively higher positive potential as illustrated in Fig. 6.2. The few electrons emitted by the photocathode are accelerated to a more positive dynode. The primary electrons can then force the ejection of secondary emission electrons when the velocity of the incident electrons is large enough. The secondary emission ratio is normally three or four, depending on the surface and the potential applied. The number of electrons available is multiplied each time the secondary electrons strike the emitting surface of the next more positive dynode. The current amplification thus obtained is noise free because the electron multiplier does not have any active device or resistors. Since the signal amplitude is very low any conventional amplifier, if used instead of the electron multiplier, woul cause serious S/N ratio problems.
Glass envelope Dynode 4 + (400 V) Dynode 3 (+ 300 V) Dynode 2 (+ 200 V) Secondary electrons Dynode 1 (+ 100 V) Photoelectrons

Anode (+ 600 V) Dynode 5 (+ 500 V) Photo cathode (0 V)

Incident light

Fig. 6.2. Illustration of an electron-multiplier structure.

Types of Camera Tubes The first developed storage type of camera tube was ‘Iconoscope’ which has now been replaced by image-orthicon because of its high light sensitivity, stability and high quality picture capabilities. The light sensitivity is the ratio of the signal output to the incident illumination. Next to be developed was the vidicon and is much simpler in operation. Similar to the vidicon is another tube known as plumbicon. The latest device in use for image scanning is the solid state image scanner.

6.2

IMAGE ORTHICON

This tube makes use of the high photoemissive sensitivity obtainable from photocathodes, image multiplication at the target caused by secondary emission and an electron multiplier. A sectional view of an image orthicon is shown in Fig. 6.3. It has three main sections: image section, scanning section and electron gun-cum-multiplier section.

(i) Image Section The inside of the glass face plate at the front is coated with a silverantimony coating sensitized with cesium, to serve as photocathode. Light from the scene to be televised is focused on the photocathode surface by a lens system and the optical image thus formed results in the release of electrons from each point on the photocathode in proportion to the incident light intensity. Photocathode surface is semitransparent and the light rays penetrate it to reach its inner surface from where electron emission takes place. Since the number of electrons emitted at any point in the photocathode has a distribution corresponding to the brightness of the optical image, an electron image of the scene or picture gets formed on the target side of the photocoating and extends towards it. Through the convertion efficiency of the photocathode is quite high, it cannot store charge being a conductor. For this reason, the electron image produced at the photocathode is made to move towards the target plate located at a short distance from it. The target plate is made of a very thin sheet of glass and can store the charge received by it. This is maintained at about 400 volts more positive with respect to the photocathode, and the resultant electric field gives the desired acceleration and motion to the emitted electrons towards it. The electrons, while in motion, have a tendency to repel each other and thin can result in distortion of the information now available as charge image. To prevent this divergence effect an axial magnetic field, generated in this region by the ‘long focus coil’ is employed. This magnetic field imparts helical motion of increasing pitch and focuses the emitted electrons on the target into a well defined electron image of the original optical image. The image side of the target has a very small deposit of cesium and thus has a high secondary emission ratio. Because of the high velocity attained by the electrons while in motion from photocathode to the target plate, secondary emission results, as the electrons bombard the target surface. These

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secondary electrons are collected by a wire-mesh screen, which is located in front of the target on the image side and is maintained at a slightly higher potential with respect to the target. The wire-mesh screen has about 300 meshes per cm2 with an open area of 50 to 75 per cent, so that the screen wires do not interfere with the electron image. The secondary electrons leave behind on the target plate surface, a positive charge distribution, corresponding to the light intensity distribution on the original photocathode. For storage action this charge on the target plate should not spread laterally over its surface, during the storage time, since this would destory the resolution of the device. To achieve this the target is made out of extremely thin sheet of glass. The positive charge distribution builds up during the frame storage time (40 ms) and thus enhances the sensitivity of the tube. It should be clearly understood, that the light from the scene being televised continuously falls on the photocathode, and the resultant emitted electrons on reaching the target plate cause continuous secondary emission. This continuous release of electrons results in the building up of positive charge on the target plate. Because of the high secondary emission ratio, the intensity of the positive charge distribution is four to five times more as compared to the charge liberated by the photocathode. This increase in charge density relative to the charge liberated at the photocathode is known as ‘image multiplication’ and contributes to the increased sensitivity of image orthicon. As shown in Fig. 6.3, the two-sided target has the charge image on one side while an electron beam scans the opposite side. Thus, while the target plate must have high resistivity laterally for storage action, it must have low resistivity along its thickness, to enable the positive charge to conduct to the other side which is scanned. It is for this reason that the target plate is very thin, with thickness close to 0.004 mm. Thus, whatever charge distribution builds up on one side of the target plate due to the focused image, appears on the other side, which is scanned, and it is from here that the video signal is obtained. (ii) Scanning Section The electron gun structure produces a beam of electrons that is accelerated towards the target. As indicated in the figure, positive accelerating potentials of 80 to 330 volts are applied to grid 2, grid 3, and grid 4 which is connected internally to the metalized conductive coating on the inside wall of the tube. The electron beam is focused at the target by magnetic field of the external focus coil and by voltage supplied to grid 4. The alignment coil provides magnetic field that can be varied to adjust the scanning beam’s position, if necessary, for correct location. Deflection of electron beam’s to scan the entire target plate is accomplished by magnetic fields of vertical and horizontal deflecting coils mounted on yoke external to the tube. These coils are fed from two oscillators, one working at 15625 Hz, for horizontal deflection, and the other operating at 50 Hz, for vertical deflection. The target plate is close to zero potential and therefore electrons in the scanning beam can be made to stop their forward motion at its surface and then return towards the gun structure. The grid 4 voltage is adjusted to produce uniform deceleration of electrons for the entire target area. As a result, electrons in the scanning beam are slowed down near the target. This eliminates any possibility of secondary emission from this side of the target plate. If a certain element area on the target plate reaches a potential of, say, 2 volts during the storage time, then as a result of its thinness the scanning beam ‘sees’ the charge deposited on it, part

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of which gets diffused to the scanned side and deposits an equal number of negative charges on the opposite side. Thus out of the total electrons in the beam, some get deposited on the target plate, while the remaining stop at its surface and turn back to go towards the first electrode of the electron multiplier. Because of low resistivity across the two sides of the target, the deposited negative charge neutralizes the existing positive charge in less than a frame time. The target can again become charged as a result of the incident picture information, to be scanned during the successive frames. As the target is scanned element by element, if there are no positive charges at certain points, all the electrons in the beam return towards the electron gun and none gets deposited on the target plate. The number of electrons, leaving cathode of the gun, is practically constant, and out of this, some get deposited and remaining electrons, which travel backwards provide signal current that varies in amplitude in accordance with the picture information. Obviously then, the signal current is maximum for black areas on the picture, because absence of light from black areas on the picture does not result in any emission on the photocathode, and there is no secondary emission at the corresponding points on the target, and no electrons are needed from the beam to neutralize them. On the contrary for high light areas, on the picture, there is maximum loss of electrons from the target plate, due to secondary emission, and this results in large deposits of electrons from the beam and this reduces the amplitude of the returning beam current. The resultant beam current that turns away from the target, is thus, maximum for black areas and minimum for bright areas on the picture. High intensity light causes large charge imbalance on the glass target plate. The scanning beam is not able to completely neutralize it in one scan. Therefore the earlier impression persists for several scans. Image Resolution. It may be mentioned at this stage that since the beam is of low velocity type, being reduced to near zero velocity in the region of the target it is subjected to stray electric fields in its vicinity, which can cause defocusing and thus loss of resolution. Also on contact with the target, the electrons would normally glide along its surface tangentially for a short distance and the point of contact becomes ill defined. The beam must strike the target at right angle at all points of the target, for better resolution. These difficulties are overcome in the image-orthicon by the combined action of electrostatic field because of potential on grid 4, and magnetic field of the long focusing coil. The interaction of two fields gives rise to cycloidal motion to the beam in the vicinity of target, which then hits it at right angle no matter which point is being scanned. This very much improves the resolving capability of the picture tube. (iii) Electron Multiplier The returning stream of electrons arrive at the gun close to the aperture from which electron beam emerged. The aperture is a part of a metal disc covering the gun electrode. When the returning electrons strike the disc which is at a positive potential of about 300 volts, with respect to the target, they produce secondary emission. The disc serves as first stage of the electron multiplier. Successive stages of the electron multiplier are arranged symmetrically around and back of the first stage. Therefore secondary electrons are attracted to the dynodes at progressively higher positive potentials. Five stages of multiplication are used, details of which are shown in Fig. 6.4. Each multiplier stage provides a gain of approximately 4 and thus a total gain of (4)5 ≈ 1000 is obtained at the electron multiplier. This is known as signal multiplication. The multiplication so obtained maintains a high signal to noise ratio. The secondary electrons are finally collected by the anode, which is connected to the highest supply

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voltage of + 1500 volts in series with a load resistance RL. The anode current through RL has the same variations that are present in the return beam from the target and amplified by the electron multiplier. Therefore voltage across RL is the desired video signal; the amplitude of which varies in accordance with light intensity variations of scene being televised. The output across RL is capacitively coupled to the camera signal amplifier. With RL = 20 K-ohms and typical dark and high light currents of magnitudes 30 µA and 5 µA respectively, the camera output signal will have an amplitude of 500 mV peak-to-peak.
Dynode no. 3, 880 V Dynode no. 5, 1450 V

Field Mesh Image Orthicon. The tube described above is a non-field mesh image orthicon. In some designs an additional pancake-shaped magnetic coil is provided in front of the face plate. This is connected in series with the main focusing coil. The location of the coil results in a graded magnetic field such that the optically focused photocathode image is magnified by about 1.5 times. Thus the charge image produced on the target plate is bigger in size and this results in improved resolution and better overall performance. Such a camera tube is known as a field mesh Image Orthicon. Light Transfer Characteristics and Applications—During the evolution of image orthicon tubes, two separate types were developed, one with a very close target-mesh spacing (less than 0.001 cm) and the other with somewhat wider spacing. The tube, with very close target mesh spacing, has very high signal to noise ratio but this is obtained at the expense of sensitivity and contrast ratio. This is a worthwhile exchange where lighting conditions can be controlled and picture quality is of primary importance. This is generally used for live shows in the studios. The other type with wider target-mesh spacing has high sensitivity and contrast ratio with more desirable spectral response. This tube has wider application for outdoor or other remote pickups where a wide range of lighting conditions have to be accommodated. More recent tubes with improved photocathodes have sensitivities several times those of previous tubes and much improved spectral response. Overall transfer characteristics of such tubes are drawn in Fig. 6.5. Tube ‘A’ is intended primarily for outdoor pick-ups where as tube ‘B’ is much suited for studio use and requires strong illumination. The knee of the transfer characteristics is reached when the illumination causes the target to be fully charged with respect to the mesh between successive scans by the electron beam. The tube is sometimes operated slightly above the knee, to obtain the black border effect (also known as Halo effect) around the high light areas of the target.

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Output current (mA)

10.0 A

1.0 B

0.1 0.0001

0.001

0.01

0.1

1.0

Illumination on photocathode (ft – candles)

Fig. 6.5. Light transfer characteristics of two different Image Orthicons.

6.3

VIDICON

The Vidicon came into general use in the early 50’s and gained immediate popularity because of its small size and ease of operation. It functions on the principle of photoconductivity, where the resistance of the target material shows a marked decrease when exposed to light. Fig. 6.6
Target connection 30 to 60 V Alignment coil Grid no. 2 (accelerator) 300 V Grid no. 1, 0 to 100 V

illustrates the structural configuration of a typical vidicon, and Fig. 6.7 shows the circuit arrangement for developing camera signal output. As shown there, the target consists of a thin photo conductive layer of either selenium or anti-mony compounds. This is deposited on a transparent conducting film, coated on the inner surface of the face plate. This conductive coating is known as signal electrode or plate. Image side of the photolayer, which is in contact with the signal electrode, is connected to DC supply through the load resistance RL. The beam that emerges from the electron gun is focused on surface of the photo conductive layer by combined action of uniform magnetic field of an external coil and electrostatic field of grid No 3. Grid No. 4 provides a uniform decelerating field between itself, and the photo conductive

Charge Image The photolayer has a thickness of about 0.0001 cm, and behaves like an insulator with a resistance of approximately 20 MΩ when in dark. With light focused on it, the photon energy enables more electrons to go to the conduction band and this reduces its resistivity. When bright light falls on any area of the photoconductive coating, resistance across the thickness of that portion gets reduces to about 2 MΩ. Thus, with an image on the target, each point on the gun side of the photolayer assumes a certain potential with respect to the DC supply, depending on its resistance to the signal plate. For example, with a B + source of 40 V (see Fig. 6.7), an area with high illumination may attain a potential of about + 39 V on the beam side. Similarly dark areas, on account of high resistance of the photolayer may rise to only about + 35 volts. Thus, a pattern of positive potentials appears, on the gun side of the photolayer, producing a charge image, that corresponds to the incident optical image. Storage Action Though light from the scene falls continuously on the target, each element of the photocoating is scanned at intervals equal to the frame time. This results in storage action and the net change in resistance, at any point or element on the photoconductive layer, depends on the time, which elapses between two successive scannings and the intensity of incident light. Since storage time for all points on the target plate is same, the net change in resistance of all elementary areas is proportional to light intensity variations in the scene being televised. Signal Current As the beam scans the target plate, it encounters different positive potentials on the side of the photolayer that faces the gun. Sufficient number of electrons from the beam are then deposited on the photolayer surface to reduce the potential of each element towards the zero cathode potential. The remaining electrons, not deposited on the target, return back and are not utilized in the vidicon. However, the sudden change in potential on each element while the beam

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scans, causes a current flow in the signal electrode circuit producing a varying voltage across the load resistance RL. Obviously, the amplitude of current and the consequent output voltage across RL are directly proportional to the light intensity variations on the scene. Note that, since, a large current would cause a higher voltage drop across RL, the output voltage is most negative for white areas. The video output voltage, that thus develops across the load resistance (50 K-ohms) is adequate and does not need any image or signal multiplication as in an image orthicon. The output signal is further amplified by conventional amplifiers before it leaves the camera unit. This makes the vidicon a much simpler picture tube. Leaky Capacitor Concept Another way of explaining the development of ‘charge image’ on the photolayer is to consider it as an array of individual target elements, each consisting of a capacitor paralleled with a light dependent resistor. A number of such representations are shown in Fig. 6.8. As seen there, one end of these target elements is connected to the signal electrode and the other end is unterminated facing the beam.
C Glass faceplate R Light C Target element Scanning beam Electron gun

R Cc v0 RL + 40V Light dependent resistor

Fig. 6.8. Schematic representation of a Vidicon target area.

In the absence of any light image, the capacitors attain a charge almost equal to the B + (40 V) voltage in due course of time. However, when an image is focused on the target the resistors in parallel with the capacitors change in value depending on the intensity of light on each unit element. For a high light element, the resistance across the capacitor drops to a fairly low value, and this permits lot of charge from the capacitor to leak away. At the time of scanning, more electrons are deposited, on the unterminated end of this capacitor to recharge it to the full supply voltage of + 40 V. The consequent flow of current that completes its path through RL develops a signal voltage across it. Similarly for black areas of the picture, the resistance across the capacitors remains fairly high, and not much charge is allowed to leak from the corresponding capacitors. This in turn needs fewer number of electrons from the beam to recharge the capacitors. The resultant small current that flows, develops a lower voltage across the load resistance.

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97

The electron beam thus ‘sees’ the charge on each capacitor, while scanning the target, and delivers more or less number of electrons to recharge them to the supply voltage. This process is repeated every 40 ms to provide the necessary video signal corresponding to the picture details at the upper end of the load resistor. The video signal is fed through a blocking capacitor to an amplifier for necessary amplification. Light Transfer Characteristics Vidicon output characteristics are shown in Fig. 6.9. Each curve is for a specific value of ‘dark’ current, which is the output with no light. The ‘dark’ current is set by adjusting the target voltage. Sensitivity and dark current both increase as the target voltage is increased. Typical output for the vidicon is 0.4 µA for bright light with a dark current of 0.02 µA. The photoconductive layer has a time lag, which can cause smear with a trail following fast moving objects. The photoconductive lag increases at high target voltages, where the vidicon has its highest sensitivity.
Dark current = 0.2 mA 1.0 0.02 mA A B C

Output current mA

0.1

0.01

0.004 mA

0.001 0.01

0.1

1.0

10

100

1000

Illumination on tube face, (ft-candles)

Fig. 6.9. Light transfer characteristics of Vidicon.

Applications Earlier types of vidicons were used only where there was no fast movement, because of inherent lag. These applications included slides, pictures, closed circuit TV etc. The present day improved vidicon finds wide applications in education, medicine, industry, aerospace and oceanography. It is, perhaps, the most popular tube in the television industry. Vidicon is a short tube with a length of 12 to 20 cm and diameter between 1.5 and 4 cm. Its life is estimated to be between 5000 and 20,000 hours.

6.4

THE PLUMBICON

This picture tube has overcome many of the less favourable features of standard vidicon. It has fast response and produces high quality pictures at low light levels. Its smaller size and light weight, together with low-power operating characteristics, makes it an ideal tube for transistorized television cameras. Except for the target, plumbicon is very similar to the standard vidicon. Focus and deflection are both obtained magnetically. Its target operates effectively as a P–I–N semiconductor diode. The inner surface of the faceplate is coated with a thin transparent conductive

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layer of tin oxide (SnO2). This forms a strong N type (N+) layer and serves as the signal plate of the target. On the scanning side of this layer is deposited a photoconductive layer of pure lead monoxide (PbO) which is intrinsic or ‘I’ type. Finally the pure PbO is doped to form a P type semiconductor on which the scanning beam lands. The details of the target are shown in Fig. 6.10 (a). The overall thickness of the target is 15 × 10– 6 m. Figure 6.10 (b) shows necessary circuit details for developing the video signal. The photoconductive target of the plumbicon functions similar to the photoconductive target in the vidicon, except for the method of discharging each storage element. In the standard vidicon, each element acts as a leaky capacitor, with the leakage resistance decreasing with increasing light intensity. In the plumbicon, however, each element serves as a capacitor in series with a reverse biased light controlled diode. In the signal circuit, the conductive film of tin oxide (SnO2), is connected to the target supply of 40 volts through an external load resistance RL to develop the camera output signal voltage. Light from the scene being televised is focussed through the transparent layer of tin-oxide on the photoconductive lead monoxide. Without light the target prevents any conduction because of absence of any charge carriers and so there is little or no output current. A typical value of dark current is around 4 nA (4 × 10– 9 Amp). The incidence of light on the target results in photoexcitation of semiconductor junction between the pure PbO and doped layer. The resultant decrease in resistance causes signal current flow which is proportional to the incident light on each photo element. The overall thickness of the target is 10 to 20 µm.
n-type layer (SnO2) Signal plate SnO2 Camera signal output Cc 1 I (mA)

Light Transfer Characteristics The current output versus target illumination response of a plumbicon is shown in Fig. 6.1 (c). It is a straight line with a higher slope as compared to the response curve of a vidicon. The higher value of current output, i.e., higher sensitivity, is due to much reduced recombination of photogenerated electrons and holes in the intrinsic layer which contains very few discontinuities. For target voltages higher than about 20 volts, all the generated carriers are swept quickly across the target without much recombinations and thus the tube operates in a photosaturated mode. The spectral response of the plumbicon is closer to that of the human eye except in the red colour region.

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6.5

SILICON DIODE ARRAY VIDICON

This is another variation of vidicon where the target is prepared from a thine n-type silicon wafer instead of deposited layers on the glass faceplate. The final result is an array of silicon photodiodes for the target plate. Figure 6.11 shows constructional details of such a target. As shown there, one side of the substrate (n-type silicon) is oxidized to form a film of silicon dioxide (SiO2) which is an insulator. Then by photomasking and etching processes, an array of fine openings is made in the oxide layer. These openings are used as a diffusion mask for producing corresponding number of individual photodiodes. Boron, as a dopent is vapourized through the array of holes, forming islands of p-type silicon on one side of the n-type silicon substrate. Finally a very thin layer of gold is deposited on each p-type opening to form contacts for signal output. The other side of the substrate is given an antiflection coating. The resulting p-n photodiodes are about 8 µm in diameter. The silicon target plate thus formed is typically 0.003 cm thick, 1.5 cm square having an array of 540 × 540 photodiodes. This target plate is mounted in a vidicon type of camera tube.
Substrate (n-type silicon) Gold coating for signal output

Scanning and Operation The photodiodes are reverse biased by applying +10 V or so to the n + layer on the substrate. This side is illuminated by the light focused on to it from the image. The incidence of light generates electron-hole pairs in the substrate. Under influence of the applied electric field,

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holes are swept over to the ‘p’ side of the depletion region thus reducing reverse bias on the diodes. This process continues to produce storage action till the scanning beam of electron gun scans the photodiode side of the substrate. The scanning beam deposits electrons on the p-side thus returning the diodes to their original reverse bias. The consequent sudden increase in current across each diode caused by the scanning beam represents the video signal. The current flows through a load resistance in the battery circuit and develops a video signal proportional to the intensity of light falling on the array of photodiodes. A typical value of peak signal current is 7 µA for bright white light. The vidicon employing such a multidiode silicon target is less susceptible to damage or burns due to excessive high lights. It also has low lag time and high sensitivity to visible light which can be extended to the infrared region. A particular make of such a vidicon has the trade name of ‘Epicon’. Such camera tubes have wide applications in industrial, educational and CCTV (closed circuit television) services.

6.6

SOLID STATE IMAGE SCANNERS*

The operation of solid state image scanners is based on the functioning of charge coupled devices (CCDs) which is a new concept in metal-oxide-semiconductor (MOS) circuitry. The CCD may be thought of to be a shift register formed by a string of very closely spaced MOS capacitors. It can store and transfer analog charge signals—either electrons or holes—that may be introduced electrically or optically. The constructional details and the manner in which storing and transferring of charge occurs is illustrated in Fig. 6.12. The chip consists of a p-type substrate, the one side of which is oxidized to form a film of silicon dioxide, which is an insulator. Then by photolithographic processes, similar to those used in miniature integrated circuits an array of metal electrodes, known as gates, are deposited on the insulator film. This results in the creation of a very large number of tiny MOS capacitors on the entire surface of the chip.
f3 f2 f1 S1 substrate p-type Surface potential t1 (b) t2 t3 Electron energy

The application of small positive potentials to the gate electrodes results in the development of depletion regions just below them. These are called potential wells. The depth of each well (depletion region) varies with the magnitude of the applied potential. As shown in Fig. 6.12 (a), the gate electrodes operate in groups of three, with every third electrode connected to a common conductor. The spots under them serve as light sensitive elements. When any image is focused onto the silicon chip, electrons are generated within it, but very close to the surface. The number of electrons depends on the intensity of incident light. Once produced they collect in the nearby potential wells. As a result the pattern of collected charges represents the optical image. Charge Transfer The charge of one element is transferred along the surface of the silicon chip by applying a more positive voltage to the adjacent electrode or gate, while reducing the voltage on it. The minority carriers (electrons in this case) while accumulating in the so called wells reduce their depths much like the way a fluid fills up in a container. The acumulation of charge carries under the first potential wells of two consecutive trios is shown in Fig. 6.12 (b) where at instant t1 a potential φ1 exists at the corresponding gate electrodes. In practice the charge transfer is effected by multiphase clock voltage pulses (see Fig. 6.12 (c)) which are applied to the gates in a suitable sequence. The manner in which the transition takes place from potential wells under φ1 to those under φ2 is illustrated in Fig. 6.12 (b). A similar transfer moves charges from φ2 to φ3 and then from φ3 to φ1 under the influence of continuing clock pulses. Thus, after one complete clock cycle, the charge pattern moves one stage (three gates) to the right. The clocking sequence continues and the charge finally reaches the end of the array where it is collected to form the signal current. Scanning of Television Pictures A large number of CCD arrays are packed together to form the image plate. It does not need an electron gun, scanning beam, high voltage or vacuum envelope of a conventional camera tube. The potential required to move the charge is only 5 to 10 volt. The spot under each trio serves as the resolution cell. When light image is focused on the chip, electrons are generated in proportion to the intensity of light falling on each cell.
1 2 3 123 Out

The principle of one-dimensional charge transfer as explained above can be integrated in various ways to render a solid-state area image device. The straightforward approach consists of arranging a set of linear imaging structures so that each one corresponds to a scan line in the display. The lines are then independently addressed and read into a common output diode by application of driving pulses through a set of switches controlled by an address register as shown in Fig. 6.13. To reduce capacitance, the output can be simply a small diffused diode in one corner of the array. The charge packets emerging from any line are carried to this diode by an additional vertical output register. In such a line addressed structure (Fig. 6.13) where the sequence of addressing the lines is determined by the driving circuitry, interlacing can be accomplished in a natural way. Cameras Employing Solid-State Scanners CCDs have a bright future in the field of solid state imaging. Full TV line-scan arrays have already been constructed for TV cameras. However, the quality of such sensors is not yet suitable for normal TV studio use. RCA SID 51232 is one such 24 lead dual-in-line image senser. It is a self-scanned senser intended primarily for use in generating standard interlaced 525 line television pictures. The device contains 512 × 320 elements and is constructed with a 3 phase n-channel, vertical frame transfer organization using a sealed silicon gate structure. Its block diagrams is shown in Fig. 6.14 (a). The image scanner’s overall picture performance is comparable to that 2/3 inch vidicon camera tubes but undesirable characteristics such as lag and microphonics are eliminated.
Bias charge circuit IG1 IG2 fH1 fH2 fH3 fOG OS OD RD fR Optical glass window Output circuit

Review Questions
1. 2. What is the basic principle of a camera pick-up tube ? Describe the two photoelectric effects used for converting variations of light intensity into electrical signals. What do you understand by image storage capability of a modern television pick-up tube ? Explain why storage type tubes have must higher sensitivity as compared to the earlier non-storage type. Draw cross-sectional view of an image orthicon camera tube and explain how it develops video signal when light from any scene is focused on its face plate. What do you understand by image multiplication and signal multiplication in an image orthicon camera tube ? Why is an electron multiplier preferred over conventional amplifiers for amplifying the video signal at the output of the camera tube ? In an image orthicon, what is the function of the wire-mesh screen and why is it located very close to the target plate ? Explain with the help of transfer characteristics the effect of targetmesh spacing on the overall performance of the tube. In an image orthicon : (a) Why is the electron beam given a cycloidal motion before it hits the target plate ? (b) Why is the electron beam velocity brought close to zero on reaching the target plate ? (c) What is the function of the decelerator grid ? 7. 8. Explain with the help of suitable sketches, how video signal is developed in a vidicon camera tube ? How is the vidicon different from an image-orthicon and what are its special applications. What do you understand by ‘dark current’ in a vidicon ? Explain how the inherent smear effect in a vidicon is overcome in a Plumbicon. Explain with a suitable sketch the mechanism by which the video siganl is developed from the P-I-N structure of its target. Give constructional details of the vidicon target prepared from a thin n-type silicon wafer which operates as an array of photodiodes. Explain how the signal voltage is developed from such a target.

3. 4.

5.

6.

9.

10. Explain with suitable sketches the basic principle of a solid state image scanner. Describe briefly the manner in which the CCD array is scanned to provide interlaced scanning.

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7
Basic Television Broadcasting

7
Basic Television Broadcasting
The composite video signal generated by camera and associated circuitry is processed in the control room before routing it to the transmitter. At transmitter, picture carrier frequency assigned to the station is generated, amplified and later amplitude modulated with the incoming video signal. The sound output associated with the scene is simultaneously processed and frequency modulated with channel sound carrier frequency. The two outputs, one from picture signal transmitter and the other from sound signal transmitter are combined in a suitable network and then fed to a common antenna network for transmission. As is obvious, the picture and sound signals, though generated and processed simultaneously pass through two independent transmitters at the broadcasting station. Thus, it is logical, that the two transmitting arrangements be studied separately. The first part of this chapter deals with television studio setup and picture signal transmission, while the later part is devoted to frequency modulation and sound signal transmission.

7.1

TELEVISION STUDIO

A TV studio is an acoustically treated compact anechoic room. It is suitably furnished and equipped with flood lights for proper light effects. The use of dimmerstats with flood lights enables suitable illumination level of any particular area of the studio depending on the scene to be televised. Several cameras are used to telecast the scene from different angles. Similarly a large number of microphones are provided at different locations to pick up sound associated with the programme. The camera and microphone outputs are fed into the control room by coaxial cables. The control room has several monitors to view pictures picked up by different cameras. A monitor is a TV receiver that contains no provision for receiving broadcast signals but operates on a direct input of unmodulated signal. A large number of such monitors are used to keep a check on the content and quality of pictures being telecast. Similarly, headphones are used to monitor and regulate sound output received from different microphones through audio mixers. In addition to a live studio, video tape recording and telecine machine rooms are located close to the control room. In most cases, programmes as enacted in the studio are recorded on a video tape recorder (VTR) through the control room. These are later broadcast with the VTR output passing through the same control room. Figure 7.1 illustrates a typical layout of a television studio setup. As shown, the telecine machines together with a slide scanner are installed next to the control room. Such a facility enables telecasting of cinematograph films and advertisement slides. All the rooms are interconnected by coaxial cables and shielded wires. In large establishments, there are several such studio units with their outputs feeding 106

the transmitter through a switcher in the master control room, which selects one programme at a time. Even in studio set-ups with only one control room, there are several studio rooms, all connected to the same control room. This enables preparation of different programmes in other studios while a programme is being telecast or recorded from one studio.

7.2

TELEVISION CAMERAS

Television cameras may assume different physical and electrical configurations. However, in general they may be divided into two basic groups—self contained cameras and two-unit systems that employ separate camera heads driven by remote camera control equipment located in the central apparatus room (see Fig. 7.1). A self contained camera has all the elements necessary to view a scene and generate a complete television signal. Such units are employed for outdoor locations and normally have a VTR and baby flood lights as an integral part of the televising setup. The remote camera head usually contains only photosensitive pick-up tube, its associated deflection circuitry, video preamplifier and a video monitor. Thus the bulk of the circuitry is contained in the camera control unit, which is connected to the camera head by means of a multiconductor cable. This cable not only carries video, deflection and sync signals but also feeds high voltage supplies necessary for the camera tube. The remote camera control unit contains most of the electrical operating and set-up controls. For this reason, it is usually located near a viewing monitor so that the results of any adjustments may be easily viewed on the monitor screen. All camera controls are available on a panel in the production control room. Camera Lenses Television cameras can produce images to different scales depending on the focal length (viewing angle) of the lens employed. Lenses of longer focal length are narrow angle lenses while those of shorter focal length are wide angle lenses. Narrow angle lenses (below 20°) are suitable for closeups of distant objects because of the magnifying effect due to their longer focal length. Lenses with angles over 60° are most suited for location shots which cover large areas. Medium angle lenses (20 to 60°) are called universal lenses and are used for televising normal scenes. All lenses consist of a combination of simple lens elements to minimize spherical aberration and other optical distortions. Lens Turret A judicious choice of lens can considerably improve the quality of image, depth of field and the impact which is intended to be created on the viewer. Accordingly a number of lenses with different viewing angles are provided. Their focal lengths are slightly adjustable by movement of the front element of the lens located on the barrel of the lens assembly. This lens compliment of the TV camera is mounted on a turret. The lens turret is screwed in the front of the camera and rotation of the turret brings the desired lens in front of the camera tube. An image orthicon turret assembly holds four lenses of focal lengths 35 mm, 50 mm, 150 mm and a zoom lens of 40 to 400 mm. Figure 7.2 shows such a lens turret mounted in front of a television camera.

BASIC TELEVISION BROADCASTING

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View finder TV camera Lens turret

Stand

Fig. 7.2. A television camera with lens turret and view finder.

Zoom Lens A zoom lens has a variable focal length with a range of 10 : 1 or more. In this lens the viewing angle and field view can be varied without loss of focus. This enables dramatic close-up control. The smooth and gradual change of focal length by the cameraman while televising a scene appears to the viewer as it he is approaching or receding from the scene. The variable focal length is obtained by moving individual lens elements of a compound lens assembly. A zoom lens can in principle simulate any fixed lens which has a focal length within the zoom range. It may, however, be noted that the zoom lens is not a fast lens. The speed of a lens is determined by the amount of light it allows to pass through it. Thus under poor lighting conditions, faster fixed focal length lenses mounted on the turret are preferred. In many camera units only a zoom lens is provided instead of the turret lens assembly. This alone enables the camera operator to have close-ups, wide coverage of the scene and distant shots without loss of focus. This is particularly so in colour TV cameras where the scene is often well defined and suitably illuminated for proper reproduction of colour details. Camera Mountings As shown in Fig. 7.2, studio cameras are mounted on light weight tripod stands with rubber wheels to enable the operator to shift the camera as and when required. It is often necessary to be able to move the camera up and down and around its central axis to pick-up different sections of the scene. In such cases, pan-tilt units may be used which typically provide a 360° rotational capability and allow tilting action of plus or minus 90°. In many applications, primarily closed circuit systems, where it is desirable to be able to remotely move the camera both horizontally and vertically, small servo motors are provided as part of the camera mount. Small motors are also used for remote focusing of the lens unit. In exceptional cases when an overview of a scene is necessary, a remotely controlled camera is hung from the ceiling. View Finder To permit the camera operator to frame the scene and maintain proper focus, an electronic view-finder is provided with most TV cameras. This view-finder is essentially a monitor which reproduces the scene on a small picture tube. It receives video signals from the control room

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stabilizing amplifier. The view-finder has its own deflection circuitry as in any other monitor, to produce the raster. The view-finder also has a built-in dc restorer for maintaining average brightness of the scene being televised. Studio Lightings In a television studio it is necessary to illuminate each area of action separately besides providing an average level of brightness over the entire scene. Lighting scheme is so designed that shadows are prevented. As many as 50 to 100 light fittings of different types are often provided in most studios. The light fixtures used include spot lights, broads and flood lights of 0.5 KW to 5 KW ratings. A number of such fittings are suspended from the top so that these can be shifted unseen by the viewer. In big studios catwalks (passages close to the ceiling) are built for ease of changing location of the suspended light fixtures. The brightness level in different locations of the studio is controlled by varying effective current flow through the corresponding lamps. For a smooth current control, dimmerstats (autotransformers) are used for low rating lamps are silicon controlled rectifiers (SCRs) for higher power lights. The power to all the lines is fed through automatic voltage stabilizers in order to maintain a steady voltage supply. The mains distribution boards and switches are located in a separate room close to the studios. The dimmerstats and other light control equipment is mounted on a separate panel in the programme control room. Audio Pick-up The location and placement of microphones depends on the type of programme. For panel discussions, news-reading and musical programmes the microphones may be visible to the viewer and so can be put on a desk or mounted on floor stands. However, for plays and many other similar programmes the microphones must be kept out of view. For such applications these are either hidden suitably or mounted on booms. A microphone boom is an adjustable extended rod from a stand which is mounted on a movable platform. The booms carry microphones close to the area of pick-up but keep them high enough to be out of the camera range. Boom operators manipulate boom arms for distinct sound pick-up yet keeping the microphones out of camera view.

7.3

PROGRAMME CONTROL ROOM

As explained earlier all video and audio outputs are routed through a common control room. This is necessary for a smooth flow and effective control of the programme material. This room is called the Programme Control Room (PCR). It is manned by the programme director, his assistant, a camera control unit engineer, a video mixer expert, a sound engineer and a lighting director. The programme director with the help of this staff effects overall control of the programme while it is telecast live or recorded on a VTR. The camera and sound outputs from the announcer’s booth and VIP studios are also routed through the programme control room. The video and audio outputs from different studios and other ancillary sources are terminated on separate panels in the control room. One panel contains the camera control unit and video mixer. In front of this panel are located a number of monitors for editing and previewing all incoming and outgoing programmes. Similarly another panel (see Fig. 7.1) houses microphone controls and switch-in controls of other allied equipment. This panel is under

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control of the sound engineer who in consulation with the programme director selects and controls all available sound outputs.

Fig. 7.3. A typical programme control room.

The producer and the programme assistant have in front of them a talk-back control panel for giving instructions to the cameramen, boom operator, audio engineer and floor manager. The producer can also talk over the intercom system to the VTR and telecine machine operators. The lighting is controlled by switches and faders from a dimmer console which is also located in the programme control room. Figure 7.3 shows a view of a typical programme control room. Camera Control Unit (C.C.U.) The camera control unit has provision to control zoom lens action and pan-tilt movement besides beam focus and brightness control of camera tubes. The C.C.U. engineer manipulates various controls under directions from the producer. In broadcast stations, the video signal must be maintained within very close tolerances of amplitudes. The C.C.U. engineer has the necessary facilities to adjust parameters such as video gain, camera sensitivity, blanking level video polarity etc. For live broadcast of programmes televised far away from the studios, microwave links are used. The modulated composite video signal received over the microwave link is demodulated and processed in the usual manner by the C.C.U. engineer for transmission on the channel allocated to the station.

7.4

VIDEO SWITCHER

A video switcher is a multicontact crossbar switch matrix with provision for selecting any one or more out of a large number of inputs and switching them on to outgoing circuits. The input sources include cameras, VTRs and telecine machine outputs, besides test signals and special

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effects generators. Thus at this point the programme producer with the assistance of video switcher may select the output of any camera, or mix the output of two or more cameras. Similarly various effects such as fades, wipes, dissolves, supers and so on may be introduced and controlled with a mixer. The results obtained from these switching procedures are quite familiar to any one who has watched commercial television programmes. Through switching, a rather restricted two-dimensional picture presented by one camera can be given additional perspective by changing the display to another camera that views the same scene from a somewhat different angle. Also information being viewed by a number of cameras at various locations can be presented on a single monitor. The ultimate destination of the outputs from the video switcher may be transmitter or a VTR. It could as well as be a string of monitors in a closed circuit television system. Types of Video Switchers. Broadcast switchers incorporate some method of vertical blanking interval controlled switching. Switching in this manner, during the vertical blanking period, eliminates any visible evidence of switching that might be observed as a disruption during the normal vertical scan. There are three types of video switchers : (i) Mechanical Pushbutton Switcher. In this type the signals are terminated on the actual switch contacts. The bank of switches is interlocked to prevent simultaneous operation. This type of switcher is used primarily for portable field units or in CCTV systems because switching is not frequent and momentary disturbances in the picture during switching can be tolerated.
C-1 C-2 C-3 MON 1

MON 2

MON 3

MON 4

MON 5

Fig. 7.4. A 3 × 5 Switching matrix.

(ii) Relay Switcher. The relay switcher or relay cross-bar is an electromechanical switcher. Here magnetically activated read switch contacts are used to effect switching. The

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relays can be operated by remote control lines. Reed relays have fast operate time (around 1 ms) and so can be used to enable switching during the vertical blanking interval. Figure 7.4 shows a 3 × 5 switching matrix employing a reed relay switcher. This is an example of a distribution-type switcher system where all inputs are available to all monitors. The X’s indicate the possibility of combinations that may be achieved. If the crosspoint indicated by the ‘X’ on the intersection of the lines from camera 3 and monitor 2 were selected, the scene viewed by camera 3 would appear an monitor 2. Similarly any or all of the remaining monitors may be selected to view any camera that is desired. Isolation amplifiers, though not shown, are used in-between the cameras and monitors. (iii) Electronic Switcher. These are all electronic switchers and use solid state devices that provide transition times of the order of a few micro-seconds. Their size is generally very small and due to inherent reliability need much less maintenance. Almost all present day switchers employed in broadcasting are electronic switchers.
Camera-1 100% % Brightness Camera-2 (a) v0 Mixing network
Amp Amp

Types of Switching Transitions. The actual switching transition is either carried out by a lap-dissolve operation or a fade out-fade in form of switching. Both methods are illustrated in Fig. 7.5. The lap-dissolve switching (Fig. 7.5(a)) may be accomplished by two potentiometers

Camera-2 output

Camera-1 output

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connected to the two signals that are to be switched. Signal amplitude from say camera number 1 is slowly reduced, while that from camera number 2 is increased at the same rate. This, as shown in Fig. 7.5(b) may be done through variable resistors connected at the inputs of two amplifiers. The values of these potentiometers may be changed as described above to control the gain of amplifiers whose outputs are then combined, Fig. 7.5(c) illustrates the fade outfade in method of video transfer. It can also be carried out by two potentiometers employed in the lap-dissolve method. However, in this case, by separate actuation of the two potentiometers, signal number 1 is slowly reduced in amplitude until zero signal level is obtained, and then signal No. 2 is slowly raised from 0 to 100 per cent by the second potentiometer. The amplifiers used are commonly called mixers or faders. Electronic Switcher Configuration Figure 7.6 is a functional block diagram of very simple broadcast switcher-mixer. It has five inputs out of which any two may be selected to drive the two buffer amplifiers. These, in turn feed into a mixer amplifier.
Camera inputs 1 Bus A Buffer amplifiers Bus B Mixer control Mixer amplifiers Switched output 2 3 Captions camera VTR 4 5

Sync input

Fig. 7.6. A simple switcher for mixing outputs from two buses.

The mixer transfers video signals by fade out-fade in method. The potentiometers at the remote mixer amplifier can be positioned to select 100 per cent output from either A or B bus. Assume A and B inputs were at 100 per cent and 0 per cent levels respectively. If camera No. 2 is selected on the A bus, it would appear at the output. Similarly, if camera No. 3 is selected on the B bus, it will not appear at the mixer output. However, when the levers that control the potentiometers are moved through their full travel, the output from the mixer amplifier would transfer from A bus to B bus at a relatively slow rate providing a transition from camera No. 2 to camera No. 3. Similarly more complex switchers can be designed to provide different switching matrices. Special Effects Generator A special effects generator is normally located along with the camera control units in the camera apparatus room. It is programmed to generate video signals for providing special effects. Its output is available at a panel in the production control room. The special effects signals include curtain moving effects, both horizontal and vertical. These are inserted while changing from one scene to another. Similarly many other patterns are available which can be interposed in-between any two programmes. Infact several options are available and can be selected while ordering the equipment.

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7.5

SYNCHRONIZING SYSTEM

To generate a meaningful picture on the raster of a monitor or receiver, some means are needed to synchronize the scanning systems of both the camera and the monitor. In a multicamera system, as is often the case in broadcasting, it is necessary to have them all synchronized by a single sync pulse generator. Accordingly a common sync drive circuitry is provided which controls scanning sequence, insertion and timing of sync pulses in all the cameras. With such a control, when the scene shifts from one camera to another, the synchronizing waveforms are in phase so that the monitor or home receiver is not interrupted in its scanning process. In the absence of such a provision, while switching from one camera to another, the monitor or receiver would have to read just its scanning procedure for the incoming camera and the picture might roll momentarily. Figure 7.7(a) shows one method of driving multiple cameras from a single sync generator. The sync line is terminated in a 75 ohm resistor because the output of most TV camera equipment is designed to work into a 75 ohm load.
Camera Sync generator Camera 1 Distribution amplifier MON 3 MON 1

MON 2

Camera 2

MON 4

Camera 3 75 W termination resistor 75 W

MON 5

Fig. 7.7(a) A sync generator driving several camera units

Fig. 7.7(b) Connections to several monitors for displaying the output of a single camera.

Camera 1

Camera 2

Camera 3

MON Push buttons

Fig. 7.7(c). Switcher for selecting any camera output to one monitor.

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Monitors usually have well designed video amplifiers with bandwidth as large as 30 MHz. This enables excellent reproduction of pictures. Any defects are clearly seen. This is useful while testing and adjusting studio and other allied equipment. Several monitors may be used to display the scene viewed by one camera. When the number of monitors is large and they are located at a considerable distance from each other, a distribution amplifier (see Fig. 7.7(b)) is used to route the video signal to all of them. Similarly it is often desirable to provide means for viewing the output from different cameras on a monitor. This is simply done by selecting the monitor inputs from one camera or another by push button switches as shown in Fig. 7.7(c). The switches not depressed connect terminating resistors to the appropriate cameras. In operation all switches are interlocked so that only one camera can be connected to the monitor at any time. Depressing one switch releases all other.
From mains (50 Hz) Shaping circuit Mains lock error voltage

A synchronized master oscillator generates an output at 2fH, viz., 2 × 15625 = 31250 Hz which drives other circuits. As shown in the figure, this oscillator can be synchronized from (i) a crystal controlled oscillator operating at exact 2fH, (ii) an external source, or (iii) 50 Hz mains supply. In the mains frequency lock mode, the 2fH multivibrator is controlled by a phase detector circuit which compares the 50 Hz square wave derived from the master oscillator through a divider chain with the 50 Hz square wave derived from mains supply. The error signal which develops at output of the phase detector corrects the frequency of the multivibrator and locks it with the mains frequency. While broadcasting programme received from another TV station, its sync pulses are processed and fed at the ‘external source’ input to slave the sync circuitry of the station to that of the incoming station. The master oscillator frequency is twice the horizontal frequency, and is coincident with the frequency of the equalizing and serration pulses which are driven from its output. The buffer amplifier isolates the master oscillator from the rest of the circuitry. The divider and gating block serves two functions. The most important function is to accurately divide the master oscillator frequency by a factor of 625, thus deriving a 50 Hz output that is phase locked with the original 31.25 kHz source. The output is correctly shaped in a shaper circuit before using it to initiate vertical drive and vertical blanking waveforms. The sync developing circuit provides equalizing and serration pulses. The output from the master oscillator (31.25 kHz) is also fed to a 2 : 1 divider to derive output at 15625 Hz, the horizontal frequency. This is used to derive horizontal blanking, horizontal drive and horizontal sync signals in appropriate circuits. The basic building block necessary for generating and shaping the various sync and drive sources include frequency dividers, pulse shapers or stretchers, delay circuits, adders and logic gates. The circuitry of a modern SPG (sync pulse generator) employs ICs and transistors. This results in a compact, accurate and reliable unit. The various outputs from the SPG are derived through distribution amplifiers which develop the necessary power and act as buffers between the generation and distribution points.

7.6

MASTER CONTROL ROOM (MCR)

In small broadcasting houses the PCR has a master switcher for routing the composite video signal and allied audio output directly to the transmitter. The ancillary equipment is mostly located in the Central apparatus room. However, in bigger establishments which have a large number of studios and production control rooms, all outputs from various sources are routed through the master control room. This room houses centralized video equipment like sync pulse generators, special effects generator, test equipment, video and audio monitors besides a master routing switcher. Picture Signal Transmission At the production control room video signal amplitude as received from the camera is very low and direct coupled amplifiers are used to preserve dc component of the signal. Further on, ac coupling is provided because it is often technically easier and less expensive to use such a coupling. This involves loss of dc component which, however, is reinserted at the transmitter before modulation. This is carried out by a dc restorer circuit often called a blanking level clamp.

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In the master control room the composite video signal is raised to about one volt P-P level before feeding it to the cable that connects the control room to the transmitter. Though the transmitter is located close to the studios, often in the same building, matching networks are provided at both ends of the connecting cable to avoid unnecessary attenuation and frequency distortion.

7.7

GENERATION OF AMPLITUDE MODULATION

In AM transmitters where efficiency is the prime requirement, amplitude modulation is effected by making the output current of a class C amplifier proportional to the modulating voltage. This amounts to applying a series of current pulses at the frequency of the carrier to the output tuned (tank) circuit where the amplitude of each pulse follows the variations of the modulating signal. The resonant frequency of the tuned circuit is set equal to the carrier frequency. In the tank circuit each current pulse causes a complete sine wave at the resonant frequency whose amplitude is proportional to the applied current pulse. The accumulative effect of this flywheel action of the resonant circuit is generation of a continuous sine wave voltage at the output of tank circuit. The frequency of this voltage is equal to carrier frequency having amplitude variations proportional to magnitude of the modulating signal.
RF input 0 Total bias t ip

Fixed bias – Vg RF in CB(RF) + CN (neutralizing capacitor) AF in

0

t v0 v0 RL 0 t

B+

Fig. 7.9. Grid modulated class C amplifier.

In practice AM may be generated by applying the modulating voltage source in series with any of the DC supplies of the class C amplifier. Thus grid (or base), plate (or collector) and cathode (or emitter) modulation are all possible. As an illustration Fig. 7.9 shows the basic circuit and corresponding waveforms of a grid modulated class C amplifier. The modulating voltage is in series with the fixed negative grid bias and so the amplitude of the total bias is proportional to the amplitude of the modulating signal, and varies at the rate of the modulating frequency. Since the carrier RF source is also in series with the bias and modulating voltage, these get superimposed and the total bias appears as shown in Fig. 7.9. The resulting plate

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current flows in pulses and the amplitude of each pulse is proportional to the instantaneous bias and therefore to the instantaneous modulating voltage. As explained earlier the application of these pulses to the tank circuit then yields amplitude modulation. In an AM transmitter, amplitude modulation can be generated at any point after the crystal oscillator. If the output stage in the transmitter is plate modulated the method is called high level modulation. If modulation is applied at any other point, including some other electrode of the output amplifier then so-called low-level modulation is produced. The end product, however, of both the system is the same. It is not practicable to use plate modulation at the output stage in a television transmitter, because of the difficulty of generating high video powers at the large bandwidths required. Accordingly, grid modulation of the output stage is the highest level of modulation employed in TV transmitters. It is called ‘high-level’ modulation in TV broadcasting and anything else is then called ‘low-level’ modulation. In recent television transmitter designs, it has become a standard practice to affect modulation in two stages. A carrier frequency of 40 MHz is employed in a balanced modulator configuration and a band ranging from 35 to 45 MHz is obtained as its output. In some designs vestigial sideband correction is also carried out in the modulator output circuit. Its output is then mixed with an appropriate high frequency to get the desired channel carrier frequency and its sidebands as the difference product. This process is illustrated for channel 4 (Band-1) in Fig. 7.10.
USB = 40 to 45 MHz LSB = 35 to 40 MHz Balanced modulator Mixer Bandpass filter (LSB) 57.25 to 67.25 MHz

A simplified functional block diagram of a television transmitter is shown in Fig. 7.11. Necessary details of video signal modulation with picture carrier of allotted channel are shown in picture transmitter section of the diagram. Note the inclusion of a dc restorer circuit (DC clamp) before the modulator. Also note that because of modulation at a relatively low power level, an amplifier is used after the modulated RF amplifier to raise the power level. Accordingly this amplifier must be a class-B push-pull linear RF amplifier. Both the modulator and power amplifier sections of the transmitter employ specially designed VHF triodes for VHF channels and klystrons in transmitters that operate in UHF channels. Vestigial Sideband Filter The modulated output is fed to a filter designed to filter out part of the lower sideband frequencies. As already explained this results in saving of band space.

Antenna The filter output feeds into a combining network where the output from the FM sound transmitter is added to it. This network is designed in such a way that while combining, either signal does not interfere with the working of the other transmitter. A coaxial cable connects the combined output to the antenna system mounted on a high tower situated close to the transmitter. A turnstile antenna array is used to radiate equal power in all directions. The antenna is mounted horizontally for better signal to noise ratio.

7.9

POSITIVE AND NEGATIVE MODULATION

When the intensity of picture brightness causes increase in amplitude of the modulated envelope, it is called ‘positive’ modulation. When the polarity of modulating video signal is so chosen that sync tips lie at the 100 per cent level of carrier amplitude and increasing brightness produces decrease in the modulation envelope, it is called ‘negative modulation’. The two polarities of modulation are illustrated in Fig. 7.12.
+v Modulating signal +v

Comparison of Positive and Negative Modulation (a) Effect of Noise Interference on Picture Signal. Noise pulses created by automobile ignition systems are most troublesome. The RF energy contained in such pulses is spread more or less uniformly over a wide frequency range and has a random distribution of phase and amplitude. When such RF pulses are added to sidebands of the desired signal, and sum of signal and noise is demodulated, the demodulated video signal contains pulses corresponding to RF noise peaks, which extend principally in the direction of increasing envelope amplitude. This is shown in Fig. 7.13. Thus in negative system of modulation, noise pulse extends in black direction of the signal when they occur during the active scanning intervals. They extend in the direction of sync pulses when they occur during blanking intervals. In the positive system, the noise extends in the direction of the white during active scanning, i.e., in the opposite direction from the sync pulse during blanking.
Noise pulse Noise pulse extends towards black Black

Obviously the effect of noise on the picture itself is less pronounced when negative modulation is used. With positive modulation noise pulses will produce white blobs on the screen whereas in negative modulation the noise pulses would tend to produce black spots which are less noticeable against a grey background. This merit of lesser noise interference on picture information with negative modulation has led to its use in most TV systems. (b) Effect of Noise Interference on Synchronization. Sync pulses with positive modulation being at a lesser level of the modulated carrier envelope are not much affected by noise pulses. However, in the case of negatively modulated signal, it is sync pulses which exist at maximum carrier amplitude, and the effect of interference is both to mutilate some of these, and to

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produce lot of spurious random pulses. This can completely upset the synchronization of the receiver time bases unless something is done about it. Because of almost universal use of negative modulation, special horizontal stabilizing circuits have been developed for use in receivers to overcome the adverse effect of noise on synchronization. (c) Peak Power Available from the Transmitter. With positive modulation, signal corresponding to white has maximum carrier amplitude. The RF modulator cannot be driven harder to extract more power because the non-linear distortion thus introduced would affect the amplitude scale of the picture signal and introduce brightness distortion in very bright areas of the picture. In negative modulation, the transmitter may be over-modulated during the sync pulses without adverse effects, since the non-linear distortion thereby introduced, does not very much affect the shape of sync pulses. Consequently, the negative polarity of modulation permits a large increase in peak power output and for a given setup in the final transmitter stage the output increases by about 40%. (d) Use of AGC (Automatic Gain Control) Circuits in the Receiver. Most AGC circuits in receivers measure the peak level of the incoming carrier signal and adjust the gain of the RF and IF amplifiers accordingly. To perform this measurement simply, a stable reference level must be available in the signal. In negative system of modulation, such a level is the peak of sync pulses which remains fixed at 100 per cent of signal amplitude and is not affected by picture details. This level may be selected simply by passing the composite video signal through a peak detector. In the positive system of modulation the corresponding stable level is zero amplitude at the carrier and obviously zero is no reference, and it has no relation to the signal strength. The maximum carrier amplitude in this case depends not only on the strength of the signal but also on the nature of picture modulation and hence cannot be utilized to develop true AGC voltage. Accordingly AGC circuits for positive modulation must select some other level (blanking level) and this being at low amplitude needs elaborate circuitry in the receiver. Thus negative modulation has a definite advantage over positive modulation in this respect. The merits of negative modulation over positive modulation, so far as picture signal distortion and AGC voltage source are concerned, have led to the use of negative modulation in almost all TV systems now in use.

7.10 SOUND SIGNAL TRANSMISSION
The outputs of all the microphones are terminated in sockets on the sound panel in the production control room. The audio signal is accorded enough amplification before feeding it to switchers and mixers for selecting and mixing outputs from different microphones. The sound engineer in the control room does so in consultation with the programme director. Some prerecorded music and special sound effects are also available on tapes and are mixed with sound output from the studio at the discretion of programme director. All this needs prior planning and a lot of rehearsing otherwise the desired effects cannot be produced. As in the case of picture transmission, audio monitors are provided at several stages along the audio channel to keep a check over the quality and volume of sound.

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Preference of FM over AM for Sound Transmission Contrary to popular belief both FM and AM are capable of giving the same fidelity if the desired bandwidth is allotted. Because of crowding in the medium and short wave bands in radio transmission, the highest modulating audio frequency used in 5 kHz and not the full audio range which extends up to about 15 kHz. This limit of the highest modulating frequency results in channel bandwidth saving and only a bandwidth of 10 kHz is needed per channel. Thus, it becomes possible to accommodate a large number of radio broadcast stations in the limited broadcast band. Since most of the sound signal energy is limited to lower audio frequencies, the sound reproduction is quite satisfactory. Frequency modulation, that is capable of providing almost noise free and high fidelity output needs a wider swing in frequency on either side of the carrier. This can be easily allowed in a TV channel, where, because of very high video frequencies a channel bandwidth of 7 MHz is allotted. In FM, where highest audio frequency allowed is 15 kHz, the sideband frequencies do not extend too far and can be easily accommodated around the sound carrier that lies 5.5 MHz away from the picture carrier. The bandwidth assigned to the FM sound signal is about 200 kHz of which not more than 100 kHz is occupied by sidebands of significant amplitude. The latter figure is only 1.4 per cent of the total channel bandwidth of 7 MHz. Thus, without encroaching much, in a relative sense, on the available band space for television transmission all the advantages of FM can be availed.

7.11 MERITS OF FREQUENCY MODULATION
Frequency modulation has the following advantages over amplitude modulation. (a) Noise Reduction The greatest advantage of FM is its ability to eliminate noise interference and thus increase the signal to noise ratio. This important advantage stems from the fact that in FM, amplitude variations of the modulating signal cause frequency deviations and not a change in the amplitude of the carrier. Noise interference results in amplitude variations of the carrier and thus can be easily removed by the use of amplitude limiters. It is also possible to reduce noise in FM by increasing frequency deviation. This deviation can be made as large as required without increasing the transmitter power. Higher audio frequencies are mostly harmonics of the lower audio range. They have low amplitudes and so cause a small deviation of the carrier frequency. Noise power interference is also generally low in amplitude and so results in frequency deviation similar to that caused by higher audio frequencies. Thus higher audio frequencies are most susceptible to noise effects. If these frequencies were artificially boosted in amplitude at the transmitter and correspondingly reduced at the receiver, improvement in noise immunity could be expected. This in fact is the standard practice in all FM transmission and reception. In AM on the other hand, the signal modulation can be increased relative to noise modulation only by increasing the transmitter output power. A 20 db improvement in signal-to-noise voltage ratio requires ten-times increase in frequency deviation in FM but an increase of 100 times in AM power output. Evidently an AM system in this respect reaches an economical limit long before the FM system, provided additional bandwidth is available for FM transmission. In an FM receiver, if two signals are being received simultanesouly, the weaker signal will be eliminated almost entirely if it possesses less than half the amplitude of the other

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stronger signal. However, in AM the interfering signal or station can be heard or received even when a 100 : 1 relationship exists between their amplitudes. Pre-emphasis and De-emphasis. The boosting of higher audio modulating frequencies, in accordance with a prearranged response curve is termed pre-emphasis, and the compensation at the receiver is called de-emphasis. Examples of circuits used for each function are shown in Fig. 7.14. As is obvious from these configurations, the pre-emphasis and de-emphasis networks are high-pass and low-pass filters respectively. The time constant of the filter for pre-emphasis at transmitter and later de-emphasis at receiver has been standardized at 50 µs in all the CCIR systems. However, in systems employing American FM and TV standards, networks having a time constant of 75 µs are used. A 50 µs (= RC) de-emphasis corresponds to a frequency

1 , which comes to 3180 Hz. 2πRC Figure 7.15 shows pre-emphasis and de-emphasis curves corresponding to a time constant of 50 µs.
response curve that is 3 db down at the frequency given by
+ Vcc
+ Vcc

(b) Transmitter Efficiency The amplitude of the FM wave is independent of the depth of modulation, whereas in AM it is dependent on this parameter. This means that low level modulation can be used in FM and all succeeding amplifiers can be class ‘C’ which are more efficient. Thus, unlike AM, all amplifiers handle constant power and this results in more economical FM transmitters. (c) Adjacent Channel Interference Because of the provision of a guard band in between any two TV channels, there is less interference than in conventional AM broadcasts. (d) Co-channel Interference The amplitude limiter in the FM section of the receiver works on the principle of passing the stronger signal and eliminating the weaker. In this manner, a relatively weak interfering signal or any pick-up from a co-channel station (a station operating at the same carrier frequency) gets eliminated in a FM system. It may be noted that from general broadcast point of view FM needs much wider bandwidth than AM. It is 7 to 15 times as large as that needed for AM. Besides, FM transmitters and receivers tend to be more complex and hence are expensive. However, in TV transmission and reception, where handling of the picture signal is equally complex, FM sound does not add very much to the cost of equipment.

7.12 GENERATION OF FREQUENCY MODULATION
The primary requirement of an FM generator is a variable output frequency, where the variations are proportional to the instantaneous value of the modulating voltage. In one method of FM generation, either the inductance or capacitance of the tank circuit of an LC oscillator is varied to change the frequency. If this variation can be made directly proportional to the amplitude of the modulating voltage, true FM will be obtained. A voltage-variable reactance is generally placed across the oscillator tank circuit. The oscillator is tuned to deliver the assigned carrier frequency with the average reactance of the variable element present in parallel with its own tank circuit. The capacitance (or inductance) of the reactance element changes on application of modulating voltage to cause frequency deviations in the oscillator frequency. Larger the departure of modulating voltage from zero, greater is the reactance variation, and in turn higher is the frequency deviation. Basic Reactance Modulator An FET, tube or transistor when suitably biased can be used as a variable reactance element. Similarly a varactor diode can also be used for this purpose. The basic circuit arrangements of a reactance modulator either with an FET or with a vacuum tube are shown in Fig. 7.16. Provided certain simple conditions are met, the impedance Z as seen at the terminals marked A – A′ in the figures, is almost entirely reactive. The conditions which must be met are, (i) the current ig should be negligible compared to the plate (or drain) current, and (ii) the impedance XC >> R, preferably by more than 5 : 1.

With these assumptions, the following analysis, which is valid for both the circuits can be made : With a voltage v applied across the terminals A – A′, currents i and ig will flow in the plate (or drain) circuit and bias circuit respectively. The current ig will develop a voltage vg = ig ×R=

The following conclusions follow from this result : (i) The equivalent capacitance depends on the device transconductance and can therefore be varied with bias voltage. The approximate relation between the bias voltage and gm is illustrated in Fig. 7.16 (c). (ii) The capacitance can be originally adjusted to any value (within reasonable limits), by variation of the components R and C. (iii) gmRC has the dimensions of capacitance. (iv) From the circuits and analysis made, it is clear that if R and C are interchanged, the impedance across A – A′ will become inductive with Leq =
RC . gm

(v) Similarly it can be shown that by using L and R instead of C and R in the biasing circuit, both capacitive and inductive reactances can be obtained across the terminals A – A′.

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Transistor Reactance Modulator Figure 7.17 shows a circuit of an LC oscillator that is frequency modulated by a capacitive reactance (RC) transistor modulator. As shown, the audio frequency voltage is applied at the base of the transistor Q1. The amplitude variations of this driving voltage vary the forward bias to change the transistor collector current. This changes β of the transistor which results in a proportionate change in the equivalent capacitance across the oscillator tank circuit. Note the use of RF chokes in the circuit. They are used to isolate various points of the circuit for ac current while providing a dc path.
+ Vcc RFC R1 C Q1 R R2 AF in C(RF) RE CE Cf R4 CE C2 RE C1 L C Reactance modulator Q2 Oscillator Tank circuit FM output Cc R3 RFC

Fig. 7.17. Reactance modulator circuit.

Varactor Diode Modulator The circuit of Fig. 7.18 shows such a modulator. It is seen that the varactor diode has been back-biased to provide the junction capacitance effect. The bias is varied by the modulating voltage which is injected in series with the dc bias source through transformer T1. The instantaneous changes in the bias voltage cause corresponding changes in the junction capacitance, which in turn vary the oscillator frequency accordingly. It is often used for automatic frequency control and remote tuning.
To oscillator tank circuit Cc

RFC

T1 AF in

Varactor diode

Cb (RF) – VD

Fig. 7.18. Varactor diode modulator.

7.13 STABILIZED REACTANCE MODULATOR
Although the oscillator on which the reactance modulator operates cannot be crystal controlled, it must nevertheless have the stability of a crystal if it is to be a part of a commercial transmitter.

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This suggests that frequency stabilization of the reactance modulator is required. The block diagram of Fig. 7.19 shows a typical AFC (automatic frequency control) system used with the reactance controlled FM transmitter. Here a fraction of the output is taken from the limiter and fed to a mixer, which also receives the signal from a crystal oscillator. The resulting difference signal at a frequency which is much less than that of the master oscillator, is amplified and fed to a phase* discriminator. The output of the discriminator which is connected to the reactance modulator, provides a dc control voltage to counteract rapidly any drift in the average frequency of the master oscillator. The time constant of the discriminator load is quite large (of the order of 100 ms). Hence the discriminator will react to slow changes in the incoming frequency, but not to normal frequency changes due to frequency modulation, which are too fast. Note that the discriminator must be connected to give a positive output if the input frequency is higher than the discriminator tuned frequency and a negative output if it is lower. Thus any drift in the oscillator frequency towards the higher side will produce a positive control voltage and this fed in series with the input of the reactance modulator, will increase its transconductance. This increases its output capacitance (Ceq = gmRC) and thus lowers the oscillator’s centre frequency. Analogous sequence of operations will take place to raise the oscillator frequency when it drifts on the lower side of its centre frequency. In the transmitter block diagram of Fig. 7.11 a reactance modulator with such a frequency control has been incorporated.
Master oscillator Buffer Frequency multiplier Limiter FM out

Reactance modulator

DC control voltage

RF discriminator

AF in Crystal oscillator

fs f0 Mixer

fs – f0

IF amplifier

Fig. 7.19. A typical transmitter AFC system.

7.14 GENERATION OF FM FROM PM
The direct modulators have the disadvantage of being based on an oscillator which is not stable enough for communication or broadcast purposes. As explained above, it needs stabilization which adds to circuit complexity. Note that the use of a crystal oscillator is not possible because it cannot be successfully frequency modulated. It is possible, however, to generate FM via Phase Modulation (PM) where a crystal oscillator can be used. It has been

where both ∆f and ∆φ are independent of fm and depend only on the modulating signal amplitude and system constants. Hence the only difference between FM and PM is that if the modulating signal is integrated before performing PM then we get FM. An integrator is a low-pass (bassboost) circuit. A simple R-L integrator is shown in Fig. 7.20 (a). Here R/(2πL) is set at about 30 Hz, so that in the audio range the response falls with frequency at 6 db/decade. Armstrong FM System Figure 7.20 (b) shows the functional block diagram of an Armstrong FM system. As explained above the audio voltage enters the modulator, which is essentially a phase modulator, after bass-boosting through an equalizer. The carrier frequency from the crystal oscillator after a phase shift of 90° is fed to a balanced modulator which also receives equalized audio signal. The two sidebands obtained from the balanced modulator are added to the unmodulated carrier in the combining amplifier. The amplitude of carrier voltage obtained from the crystal oscillator through the buffer stage is kept quite large in comparison with that of the sidebands. This as usual, is essential for effective modulation.
L AF in R Equalized AF out E out/E in a R/w L fd (a) RL equalizer (c) FM phasor diagram fc Ð0° Crystal oscillator Carrier frequency (fc) Buffer amplifier Combining amplifier Frequency multipliers Power amplifier FM out (d) AM phasor diagram Em Em Ec fd Ec

Em Em

l = L/R

Side bands 1/f network (audio equalizer)

90° phase shifter

Balanced modulator

AF in

fc Ð90°

(b) Block diagram

Fig. 7.20. The Armstrong frequency-modulation system.

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As shown in the phasor diagram of Fig. 7.20 (c), the two sideband phasors rotate in unison but in opposite direction, and the sum of the carrier and sidebands causes the carrier to change phase by the angle φd. It will be noted that the phase relation of carrier and side frequencies is at 90° with respect to the AM situation as shown in Fig. 7.20 (d) where only the amplitude of the carrier is varied and not the phase. It may also be noted that the resultant phasor (see Fig. 7.20 (c)) in the case of phase modulation varies in amplitude besides phase deviation and thus a little amplitude modulation is also present in the output. However, the change in amplitude is made very small, though at a price, by making the phase deviation very small. A typical value is ∆f = 50 Hz at a carrier frequency of 1 MHz corresponding to a maximum phase deviation of 5 × 10– 5 radians. For such a small phase deviation, incidental amplitude variation is negligible. The most convenient crystal oscillator frequency is around 1 MHz, while TV broadcast carrier frequencies are in the region of 50 to 100 MHz. Frequency multipliers are used to raise the carrier frequency. It may be noted that in frequency multiplication, ∆f and the carrier frequency get multiplied by the same factor. Usually to achieve the final ∆f of ± 75 kHz, a larger multiplying factor is required than that needed to raise the carrier frequency to the required value. This is accomplished by shifting the carrier frequency down by a hetrodyning process at some point within the multiplier chain.

7.15 FM SOUND SIGNAL
As explained in an earlier section, audio signals from different microphones are received at the sound panel in the production control room. After due amplification all the outputs are fed into a switcher, where if necessary they are mixed and the desired output is selected. The final output goes to a distributor in the master control room, where both picture and sound signals from different studios are received. This distributor is switched to select corresponding picture and sound signals from the desired studio. As in case of video signals the audio signals are also routed to the sound transmitter through a cable (see Fig. 7.11) with matching networks on either side. At transmitter the audio signal is frequency modulated and transferred to assigned channel sound carrier frequency by the use of multipliers. It is later amplified through several stages of power amplifiers to raise the power output to desired level. Audio monitors are provided at various points to keep a check on the sound quality. It is finally fed to the common antenna array through a combining network for radiation along with the modulated picture signal.

Review Questions
1. 2. 3. Draw the layout of a typical television studio and explain how the picture and sound signals are processed in the control room. What is the role of a special effects generator ? What is the difference between a self-contained and a two-unit camera system ? What is the function of view finder which is provided at the hood of camera ? Explain how in a multicamera system, synchronization is maintained between the cameras and control monitor. Explain with a functional block diagram how sync and equalizing pulses are generated and kept phase-locked with a common master oscillator.

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4.

Draw a 3 × 5 switching matrix designed to select output from different cameras on any of the five monitors. Explain how an electronic video switcher accomplishes a smooth change-over from one output to another through a mixer amplifier. Draw a block diagram to show how the video signal is modulated and processed at the picture transmitter. Why is high level modulation not used in a TV transmitter ? Discuss the merits and demerits of positive and negative amplitude modulation and justify the choise of negative modulation in most TV systems. Why is FM chosen for transmission of sound signal in TV systems ? Why are pre-emphasis and de-emphasis circuits provided at the FM transmitter and receiver respectively ? How is frequency modulation produced ? Draw the circuit of a basic reactance modulator and prove that its output reactance varies with changes in the amplitude of the drive voltage. Draw the block diagram of an AFC circuit that forms part of a reactance FM modulator to stabilize the centre frequency of the master oscillator. Explain its control action.

5. 6. 7. 8. 9.

10. Explain briefly how a phase modulator can be used to generate frequency modulation. Draw the block schematic diagram of an Armstrong modulator and explain how an FM output is obtained from it. What is the main merit of this modulator ?

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8
Television Receiver

8
Television Receiver
The preceding chapters were devoted to the complexities and essential requirements of generation and transmission of composite video and sound signals associated with the televised scene. As is logical, we should now turn our attention to the receiver. In effect, a television receiver is a combination of an AM receiver for the picture signal and an FM receiver for the associated sound. In addition, the receiver also provides suitable scanning and synchronizing circuitry for reproduction of image on the screen of picture tube. We shall confine our discussion to monochrome (black and white) receivers. The basic principle and essential details of colour receivers are described in Chapters 25 and 26. However, it may be noted that all the circuits for a black-and-white picture are also needed in a colour receiver. The colour television picture is just a monochrome picture with colour added in the main areas of picture information.

8.1

TYPES OF TELEVISION RECEIVERS

The receiver may use tubes for all stages, have all solid-state devices-transistors and integrated circuits, or combine tubes and transistors as a hybrid receiver. A typical chassis of a monochrome receiver is shown in Fig. 8.1.
Video and sound (intercorrier) IF section Video and audio output circuits

Fig. 8.1. Rear view of a black and white receiver with the back removed.

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(a) All Tube Receivers. This type mainly applies to earlier monochrome receivers. All the functions are provided by about 12 tubes including several multipurpose tubes with two or three stages in one glass envelope. The dc supply for tubes is between 140 to 280 volts. (b) Solid-State Receivers. In this type all states except the picture tube use semiconductor diodes, transistors and integrated circuits. The dc supply is between 12 to 100 volts for various stages. The heater power to the picture tube is supplied through a separate filament transformer. (c) Hybrid Receivers. In this type the deflection circuits generally use power tubes, while the signal circuits use transistor and integrated circuits. These receivers usually have a line connected power supply, with series heaters for the tubes. Two dc sources, one for semiconductor devices and the other for tubes are provided.

8.2

RECEIVER SECTIONS

It is desirable to have a general idea of the organization of the receiver before going into circuit details. Figure 8.2 shows block schematic diagram of a typical monochrome TV receiver. As shown there, the receiver has been divided into several main sections depending on their functions and are discussed below.
UHF antenna UHF Tuner VHF antenna UHF mixer Local osc Fine tuning RF amplifier Mixer
Video IF amplifier 2/3 stages

Antenna System Strongest signal is induced in the antenna if it has same polarization as the transmitting antenna. All TV antennas are mounted in horizontal position for better reception and favourable signal to noise ratio. The need for good signal strength has led to the use of tuned antennas. For channels located in the VHF band, a half wave-length antenna is most widely used. Such antennas behave like low ‘Q’ tuned circuits and a single antenna tuned to the middle frequency of various channels of interest can serve the purpose. Various antennas in use are of dipole type with reflectors and directors. A folded dipole with a reflector is used because its response is more uniform over a band of frequencies. A Yagi antenna, i.e., a dipole with one reflector and two or more directors, is a compact high gain directional array, and is often used in fringe areas. In areas where signal strength is very low, booster amplifiers with suitable matching network are used. On the other hand, in areas situated close to a transmitting antenna, where signal strength is quite high, various types of indoor antennas are frequently employed. Since it is not possible for one dipole antenna to cover both upper and lower VHF band channels effectively; high and low band dipoles are mounted together and connected to a common transmission line. For channels in the UHF bound, where the attenuation is very high and the signals reaching the antenna are weaker, special antennas like fan dipole, rhombic and parabolic reflector type are often used. A transmission line connects antenna to the receiver input terminals for the RF tuner. A twin-lead is generally used. This type is an unshielded balanced line with characteristic impedance equal to 300 ohms. When there is a problem of interference, a shielded coaxial cable is used. This cable has high attenuation, especially at UHF channel frequencies. It has a characteristic impedance of 75 ohms. The current practice is to design input circuit of the TV receiver for a 300 ohm transmission line. It has been found that a 300 ohm transmission line used with a half-wave dipole produces a broad frequency response without too large a loss due to mismatching. A folded dipole has an impedance close to 300 ohms at its resonant frequency, and a much uniform response is obtained with this antenna. Receivers designed to receive UHF channels have two inputs; one to match a 300 ohm transmission line and the other for a 75 ohm coaxial cable. A signal strength of the order of 500 µV to 1 mV and a signal to noise ratio of 30 : 1 are considered adequate for satisfactory reception of both picture details and sound output. RF Section This section consists of RF amplifier, mixer and local oscillator and is normally mounted on a separate sub-chassis, called the ‘Front End’ or ‘RF Tuner’. Either tubes or transistors can be used. With tubes, local oscillator and mixer functions are usually combined in one stage called the ‘frequency converter’. The purpose of the tuner unit is to amplify both sound and picture signals picked up by the antenna and to convert the carrier frequencies and their associated bands into the intermediate frequencies and their sidebands . The receiver uses superhetrodyne principle as used in radio receivers. The signal voltage or inoformation from various stations modulated over different carrier frequencies is hetrodyned in the mixer with the output from a local oscillator to transfer original information on a common fixed carrier frequency called the intermediate frequency (IF). The setting of the local oscillator frequency enables selection of desired station. The standard intermediate frequencies for the 625-B system are-Picture IF = 38.9 MHz, Sound IF = 33.4 MHz.

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In principle an RF amplifier is not necessary and signal could be fed directly to the tuned input circuit of the mixer. However, the problems of a relatively weak input signal with low signal to noise ratio, local oscillator radiation and image rejection are such, that a stage of amplification ahead of the mixer is desirable. The tuning for different channels is carried out with a channel selector switch which changes resonant frequencies of the associated tuned circuits by varying either inductance or capacitance of these circuits. The RF section is shown separately in Fig. 8.3 (a), where the channel selector switch has been set for channel 4 (Band I),

Fig. 8.3 (a). Block diagram of a VHF tuner. Selector switch set for channel 4 band I (61-68 MHz).

Fig. 8.3 (b). Ideal response curve of the RF amplifier when set for channel 4.

i.e., 61 to 68 MHz. The picture carrier frequency in this channel is 62.25 MHz and the sound carrier 67.75 MHz. The RF amplifier must have sufficient bandwidth to accept both the picture and sound signals. This is illustrated in Fig. 8.3 (b). The local oscillator frequency is set at 101.15 MHz. In the mixer, both sum and difference (sideband) frequencies are generated. The output circuit of the mixer is however, tuned to deliver difference frequencies i.e., the intermediate frequencies and their sidebands. The required IFs are then produced as here: (Local oscillator frequency of 101.15 MHz)–(Picture carrier frequency of 62.25 MHz) = Picture IF of 38.9 MHz, (Local oscillator frequency of 101.15 MHz)–(Sound carrier frequency of 67.75 MHz) = 33.4 MHz. The desired output response from the mixer is shown in Fig. 8.4. Notice that frequency changing process reverses the relative positions of the sound and picture signals. This is obvious, since the oscillator works above the signal frequencies, and ‘difference’ frequencies produced, when the picture and sound frequencies are substracted must give a

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higher IF from the lower frequency picture signal. It may be noted that picture and sound signals would remain in the same relative position, i.e., with sound carreir frequency higher than picture carrier frequency if local oscillator frequency is set below, instead of above the carriers. The local oscillator frequency is kept higher because of ease of oscillator design and several other merits. The ratio of highest to lowest radio frequency that the local oscillator must generate, when the oscillator frequency is chosen to be higher than the incoming carrier frequency, is much less than when the local oscillator frequency is kept below the incoming channel frequency. It is much easier to design an oscillator that maintains almost a constant output amplitude and a sinusoidal waveshape when its overall frequency range is less. This justifies the choice of higher local oscillator frequency.

Fig. 8.4. Location of sound and picture IF frequencies at the output of mixer.

The tuning of RF and oscillator tuned circuits is pre-set for switching in different channels. Despite the fact that modern tuner units are remarkably stable, most receiver manufacturers provide a fine tuning control for small adjustments of local oscillator frequencies. The control is varied to obtain best picture results on the screen. IF Amplifier Section A short length of coaxial cable feeds tuner output to the first IF amplifier. This section is also called video IF amplifier since composite video signal is the envelope of the modulated picture IF signal. Practically all the gain and selectivity of the receiver is provided by the IF section. With tubes, 2 or 3 IF stages are used. With transistors, 3 to 4 If stages are needed. In integrated circuits, one IC chip contains all the IF amplifier stages. Essential Functions of the IF Section The main function of this sections is to amplify modulated IF signal over its entire bandwidth with an input of about 0.5 mV signal from the mixer to deliver about 4 V into the video detector. This needs an overall gain of about 8000. This gain should be adjustable, by automatic gain control, over a wide range to accommodate input signal variations at the antenna from 50 µV to 0.5 V, to deliver about 4 V peak-to-peak signal at the input of the video detector. To achieve desired gain, atleast three stages of tuned amplifiers are cascaded and to obtain desired bandwidth the resonant frequencies of these stages are staggered. Such an arrangement provides desired gain and selectivity.

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8.3

VESTIGIAL SIDEBAND CORRECTION

Another important function assigned to IF section is to equalize the amplitudes of side-band components, because of vestigial sideband transmission. The need for this was fully explained in Chapter 4, and a reference to Fig. 4.8 will show, that for vestigial side-band correction the picture carrier frequency gain must be 6 db down on the IF frequency response curve. It is also necessary to shape the IF response curve around the picture IF frequency in such a way that all lower video frequencies, which got a boost on account of partial lower side-band transmission (besides the full upper side band), are duly attenuated and get restored to their actual level. This is achieved by suitable tuning and shaping the response of the IF stages. This is fully illustrated in Fig. 8.5, which shows ideal overall response of the IF section.
Lower adjacent channel–2 fcP = 48.25 MHz fcS = 53.75 MHz db
0

In the IF amplifier circuitry, provision must be made for rejection of signals from adjacent channels. For this purpose special tuned circuits, called trap circuits, are connected in the signal path in such a way that the offending frequencies are removed. These trap circuits are disposed at convenient places in the IF amplifiers. Their position will vary from receiver to receiver, but generally they are placed in the input circuit of the first IF amplifier. The way in which these unwanted adjacent channel IF signals appear is illustrated in Fig. 8.5. As an example, suppose that the receiver is switched to channel 3 on Band I. The local oscillator

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frequency for channel 3 is (55.25 + 38.9) 91.15 MHz. This would beat with the channel picture and sound carrier frequencies to give the desired picture and sound IF frequencies. Besides these, the sound carrier of channel 2, which is close to the beginning of channel 3, will beat with the local oscillator to give unwanted difference frequency of 40.4 MHz (94.15 – 53.75), which would lie close to the upper skirt of desired IF response. Similarly, the picture carrier of upper adjacent channel 4, will also beat with the local oscillator to produce another unwanted difference frequency signal of 31.9 MHz (94.15 – 62.25). This is close to the lower skirt of IF response. The trap circuits are designed to attenuate these two adjacent channel interfering frequencies by about 40 db as shown in the figure. It is understood that such interference would occur only if transmitters operating at channels 2 and 4 are located close to the transmitter operating at channel 3.

8.4

CHOICE OF INTERMEDIATE FREQUENCIES

Since the picture and sound carriers in any channel are spaced by 5.5 MHz, it is natural that the corresponding IF frequencies are also located at the same difference. Accordingly, if the picture IF is fixed at a certain frequency the sound IF automatically gets fixed at a frequency 5.5 MHz less than the picture IF frequency. Therefore we shall refer mostly to picture IF frequency. The factors which influence the choice of intermediate frequencies in TV receivers are: (i) Image Rejection Ratio For a desired input signal at 100 MHz the local oscillator frequency is set at 110 MHz if the IF frequency is fixed at 10 MHz. However, for the same input signal frequency, if the IF frequency is chosen to be 40 MHz, the local oscillator must be set to give an output at 140 MHz. This is shown in Fig. 8.6 (a) and (b). In the first case, if another station is operating at 120 MHz, it will
Desired signal frequency = 100 MHz Image signal frequency = 120 MHz Input Mixer Desired signal frequency = 100 MHz Image signal frequency = 180 MHz Mixer IF out 40 MHz

also be received with equal strength because the incoming signal will beat with the local oscillator frequency of 110 MHz to develop output at 10 MHz. Similarly in the second case, a station operating at 180 MHz will be received equally well because the output circuit of mixer is tuned to deliver output at 40 MHz. Note that in each case the undesired signal which gets received is spaced at a gap of twice the IF frequency, and is known as ‘Image Signal’. The image rejection ratio is defined as the output due to desired station divided by output due to image signal. Without the use of an RF amplifier prior to the mixer, there is nothing that can stop the

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reception of image signal if that is present. With RF amplifier the output due to image signal can be very much reduced or completely eliminated. With lower IF frequency, say 10 MHz, the image frequency at 120 MHz is not very far away from desired frequency of 100 MHz and might pass through the pass-band of RF amplifier through somewhat attenuated. But if the IF frequency is kept high, as shown in Fig. 8.6 (b), image signal frequency is 80 MHz away from the desired signal and has no chance of passing through the RF amplifier. Thus the use of RF amplifier helps in reducing interference due to image signals and a higher IF frequency results in a very high image rejection ratio. By choosing an IF greater than half the entire band to be covered it is possible to eliminate image interference. For the lower VHF band (41 to 68 MHz) the IF frequency comes to 13 MHz. In the upper VHF band (174 to 230 MHz) desired IF frequency is 28 MHz. In the UHF band (470 to 528 MHz), where the image problem is most serious, half of the difference of entire band results in the choice of an IF frequency of 56 MHz. But this is higher than the lowest frequency used in the lower VHF band and because of direct pick-up problems in that band, it cannot be used. Therefore, the IF frequency must be less than 41 MHz. (ii) Pick-up Due to Local Oscillator Radiation from TV Receivers If the output from the local oscillator of a TV receiver gets coupled to the antenna, it will get radiated and may cause interference in another receiver. This is shown in Fig. 8.6 (c). Here again advantage lies with higher IF frequency, because with higher IF there is a greater separation between the resonant circuits of local oscillator and RF amplifier circuits. Thus lesser signal is coupled from the local oscillator through the RF amplifier to the antenna circuit and interference due to local oscillator radiation is reduced. (iii) Ease of Separation of Modulating Signal from IF Carrier at the Demodulator For ease of filtering out the IF carrier freuency, it is desirable to have a much higher IF frequency as compared to the highest modulating frequency. In radio receivers the IF frequency is 455 KHz and the highest audio frequency is only 5 KHz. In TV receivers, with the highest modulating frequency of 5 MHz, an IF frequency of atleast 40 MHz is desirable. (iv) Image Frequencies Should Not Lie in the FM Band The FM band is from 88 MHz to 110 MHz. With IF frequency chosen close to 40 MHz, the image frequencies of the lower VHF band fall between 121 to 148 MHz and thus cannot cause any interference in the FM band. Higher TV channels are much above the FM band. (v) Interference or Direct Pick-Up from Bands Assigned for other Services Amateur and industrial applications frequency band lies between 21 to 27 MHz. If the IF frequency is chosen above 40 MHz, even the second harmonics of this band will not cause any serious direct pick-up problems. (vi) Gain It is easier to build amplifiers with large gain at relatively low frequencies. The TV sets manufactured some 30 years back used IF frequency as low as 12 MHz. This was mostly due to limitations of active devices available, and the poor quality of components marketed at that time. With the rapid strides made by electronics industry during the past three decades, active devices which can perform very well at high frequencies are now easily available. The quality

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of components and other techniques have also considerably improved. Thus the gain criteria is no longer a constraint in choosing higher IF frequencies. The merits of having high IF frequency are thus obvious and this has lead to the choice of IF frequencies close of 40 MHz. In the 625B system adopted by India and several other countries the recommended IF frequencies are : Picture IF = 38.9 MHz, Sound IF = 33.4 MHz. It will be of interest to note that sets manufactured in USA have picture IF = 45.75 MHz and sound IF = 41.25 MHz. In the British 625 line system, because of channel bandwidth difference, the picture IF = 39.5 MHz and sound IF = 33.5 MHz. Video Detector. Modulated IF signals after due amplification in the IF section are fed to the video detector. The detector is designed to recover composite video signal and to transform the sound signal to another lower carrier frequency. This is done by rectifying the input signal and filtering out unwanted frequency components. A diode is used, which is suitably polarized to rectify either positive or negative peaks of the input signal. Figure 8.7 shows a simplified circuit arrangement of a video detector. Note the use of an L-C filter instead of the usual RC configuration employed in ratio receiver detectors. This is to avoid undue attenuation of the video signal while filtering out carrier components. The video signal shown in Fig. 8.7 is of correct polarity for feeding to the cathode of picture tube after one stage of video amplification.

Video Amplifier. The picture tube needs video signal with peak-to-peak amplitude of 80 to 100 volts for producing picture with good contrast. With an input of about 2 volts from the detector, the video amplifier is designed to have a gain that varies between 40 to 60. A contrast control which essentially is gain control of the video amplifier is provided on the front panel of the receiver to adjust contrast between black and white parts of the picture. A large constrast makes the picture hard, whereas too low a value leaves it weak or soft. The video amplifier is dc coupled from the video detector to the picture tube, in order to preserve the dc component for correct brightness. However, in some video amplifier designs, on account of complexities of a direct coupled amplifier, ac coupling is instead used. The dc component of the video signal is restored by a diode clamper before feeding it to cathode or grid of the picture tube. In video amplifiers that employ tubes, one stage is enough to provide the desired gain. In transistor amplifier designs, a suitable configuration of two transistors and a driver often becomes necessary to obtain the same gain.

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Besides gain, response of the amplifier should ideally be flat from dc (zero) to 5 MHz to include all essential video components. This needs rigorous design considerations because the band of frequencies to be covered extends from dc through audio range to radio frequencies. A loss in gain of high frequency components in the video signal would reduce sharpness of the picture whereas a poor low frequency response will result in loss of boundary details of letters etc. It is also essential that phase distortion in the amplifer is kept to a minimum. Excessive phase distortion at low frequencies results in smear effect over picture details. Thus the video amplifier of a television receiver needs careful design to achieve desired characteristics. Various wide-banding techniques are employed to extend bandwidth of the amplifier.

8.5

PICTURE TUBE CIRCUITRY AND CONTROLS

The output from the video amplifier may be fed either at the cathode or control grid of the picture tube. In either case a particular polarity of the video signal is essential for correct reproduction of picture details. In most cases cathode drive is preferred. The grid is thus left free to receive retrace blanking pulses to ensure that no retrace lines are seen on the screen for any setting of the brightness control. Figure 8.8 shows the passage of video signal from video detector to the picture tube.
Composite video signal Amplified video signal 80 V P – P vin Modulated Video IF signal detector Video amplifier EHT

B+ Contrast control Brightness control

Fig. 8.8. Passage of video signal from detector to picture tube.

8.6

SOUND SIGNAL SEPARATION

The picture and sound signals on their respective carriers are amplified together in the IF section. On application of the two signals to the video detector, the picture IF (38.9 MHz) acts as the carrier and beats with the sound carrier (33.4 MHz) and its associated FM side-band frequencies, to produce difference i.e., 5.5 MHz ± 50 KHz components.This is called intercarrier beat signal and is in effect the second conversion of sound carrier frequency. The resultant product, however, retains its original FM modulation. If amplitude variations in the FM modulated difference signal of 5.5 MHz is to be avoided to suppress audio signal distortion, the amplitude of sound IF carrier (33.4 MHz) together with its side bands must be attenuated by about 20 db below the picture IF carrier level. This is achieved by shaping the lower skirt of the IF section response in such a way that the sound IF lies at — 26 db (5 per cent of the maximum) on the voltage gain axis of the IF response. This

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is clearly shown in Fig. 8.5, where a small pedestal has been created at 33.4 MHz to achieve the desired objective. Note that any drift in the local oscillator frequency has no effect on the inter-carrier sound beat frequencies. This is so, because any shift in the local oscillator frequency, shifts both the picture and sound IFs by the same amount. The video detector circuit is modified (see Fig. 8.7) by providing a resonant trap circuit to isolate the sound signal. In some receivers the inter-carrier sound signal is separated after one stage of amplification in the video amplifier.

8.7

SOUND SECTION

As shown in the receiver block diagram (Fig. 8.2), the relatively weak FM sound signal, now on a carrier frequency of 5.5 MHz is given at least one stage of amplification before feeding it to the FM detector. This stage is a tuned amplifier, with enough bandwidth to pass the FM sound signal. This tuned amplifier is known as sound IF. The FM detector is normally a ratio detector or a discriminator preceded by a limiter. The characteristics of a typical FM detector are shown in Fig. 8.9. As shown, the audio output is proportional to deviations from the carrier frequency. The frequency of audio signal depends on the rate of frequency deviation. At the output of FM detector, a de-emphasis circuit is provided that has the same time constant (50 µs) as that of the pre-emphasis circuit employed at the sound transmitter. This restores the amplitude of higher audio frequencies to their correct level. The audio signal receives atleast one stage of amplification before it is passed on to the audio output (power) amplifier. The volume and tone controls form part of the audio amplifiers. These are brought out at the front panel of the receiver. The power amplifier is either a single ended or push-pull configuration employing tubes or transistors. Special ICs have been developed which contain FM demodulator and most parts of the audio amplifier. These are fast replacing discrete circuits hitherto used in the sound section of the receiver. The audio amplifier feeds into one or two loudspeakers provided at a convenient location along front panel of the receiver.

Fig. 8.9. Response curve of an FM detector.

Automatic Gain Control (AGC) AGC circuit controls gain of RF and IF stages to deliver almost constant signal voltage to the video detector, despite changes in the signal picked up by the antenna. The change in gain is

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achieved by shifting the operating point of the amplifying devices (tubes or transistors) used in the amplifiers. The operating point is changed by a bias voltage that is developed in the AGC circuit. Any shift in the operating point changes gm (mutual conductance) of the tube or power gain of the transistor circuit which in turn results in change of stage gain. Sync level in the composite video signal is fixed irrespective of the picture signal details. Hence, sync pulse tops represent truly the signal strength. A peak rectifier is used to develop a control voltage which is proportional to the sync level. The composite video signal to the peak rectifier in the AGC circuit is either obtained from the output of video detector or after one stage of video amplification. The output is filtered and the dc voltage thus obtained is fed to the input (bias) circuits of the RF and IF amplifiers to control their gain. Decoupling circuits are used to avoid interaction between different amplifier stages. AGC is normally not applied to the last IF stage because at that level the signal strength is quite large and any shift in the initially chosen operating point can cause distortion because of partial operation on the nonlinear portion of the device characteristics. Since AGC voltage is proportional to the signal strength, even weak RF signals will also develop some control voltage. This when applied to the RF amplifier will tend to reduce its gain, though the stage should provide maximum possible gain for weak signals to maintain high signal to noise ratio. Therefore, the RF amplifier is not fed any AGC voltage till the signal strength attains a certain predetermined level. This is achieved by providing a voltage delay circuit in the AGC line. Such a provision is known as delayed AGC. The AGC control, as explained above is illustrated in Fig. 8.10 by a block schematic circuit arrangement.
Composite video signal (input) To IF amplifiers Decoupling network R2 AGC delay circuit C1 AGC bias line C2 R1 RL Peak AGC rectifier Rectified output v0

To RF amplifier

R3 C3

AGC filter

Fig. 8.10. Block diagram of AGC system.

Sync Separation The horizontal and vertical sync pulses that form part of the composite video signal are separated in the sync separator. The composite video signal is either taken from the video detector output or after one stage of video amplification. A sync separator is a clipper that is suitably biased to produce output, only during sync pulse amplitude of the video signal. In some receivers, a noise gate preceds the sync separator. This suppresses strong noise pulses if present in the video signal. A sync pulse train as obtained from a sync separator is shown in Fig. 3.5.

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8.8

SYNC PROCESSING AND AFC CIRCUIT

The pulse train as obtained from the sync separator is fed simultaneously to a differentiating and an integrating circuit. The differentiated output (see Fig. 3.5) provides sharp pulses for triggering the horizontal oscillator, while output from the integrator controls the frequency of the vertical oscillator. As explained in Chapter 3, pre and post equalizing pulses ensure identical vertical sync pulses both after the first and second fields. An integrating circuit is a low-pass filter and hence sharp noise pulses do not appear at its output. However, the differentitator, being a high-pass filter, develops output in response to noise pulses in addition to the spiked horizontal sync pulses. This results in occasional wrong triggering of the horizontal oscillator which results in diagonal tearing of the reproduced picture. To overcome this difficulty, a special circuit known as automatic frequency control (AFC) circuit (Fig. 8.11) is employed. The AFC circuit employs a discriminator arrangement which compares the incoming horizontal sync pulses and the voltage that develops across the output of the horizontal deflection amplifier. The AFC output is a dc control voltage that is free of noise pulses. This control voltage is used to synchronize the horizontal oscillator with the received horizontal sync pulses.
Sync voltage

Sync discriminator Filter Horz fly back pulses from H.O.T.

R C

D.C. control voltage

Fig. 8.11. Block diagram of AFC circuit.

8.9

VERTICAL DEFLECTION CIRCUIT

Blocking oscillators or cathode coupled multivibrators are normally employed as vertical deflection oscillators. The controlling time constants are suitably chosen to develop output corresponding to trace and retrace periods. The necessary sawtooth voltage is developed by charging and discharging a capacitor with different time constants. This capacitor forms part of the waveshaping circuit which is connected across the oscillator. The frequency of the oscillator is controlled by varying the resistance of the RC coupling network and is locked in synchronizm by the vertical sync pulses. A part of the coupling network resistance is a potentiometer that is located on the fornt panel of the reciver. This is known as ‘Vertical Hold Control’ (see Fig. 8.2) and enables resetting of the vertical oscillator frequency when it drifts far away from 50 Hz. The output from the oscillator-cum-waveshaping circuit if fed to a power amplifier, the output of which is coupled to the vertical deflection coils to produce vertical deflection of the beam on picture tube screen.

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8.10 HORIZONTAL DEFLECTION CIRCUIT
The horizontal oscillator (see Fig. 8.2) is similar to the vertical oscillator and is set to develop sweep drive voltage at 15625 Hz. However, the frequency of this oscillator is controlled by dc control voltage developed by the AFC circuit. Since the noise pulses in the control voltage are completely suppressed, most receivers do not provide any horizontal frequency (hold) control, as is normally done for the vertical oscillator. The oscillator output is waveshaped to produce linear rise of current in the horizontal deflection coils. Since the deflection coils need about one amp of current to sweep the entire raster, the output of the oscillator is given one stage of power amplification (as for vertical deflection) and then fed to the horizontal deflection coils. Low Voltage Power Supply The usual B + or low voltage power supply is obtained by rectifying and filtering the ac mains supply. If necssary the mains voltage is stepped up or down before rectification. Silicon diodes are normally used for rectification. In some power supply designs, which do not employ a mains transformer, voltage doubler circuits are used to raise the dc voltage. For circuits that employ transistors and integrated circuits, regulated low voltage power supplies are normally provided. While branching the dc supply to various sections of the receiver, decoupling networks are used to avoid any undue coupling between different sections of the receiver. The filament power is supplied by either connecting all the heaters in series across the ac mains or by a low voltage winding on the mains transformer. High Voltage (EHT) Supply As already stated in the chapter on picture tubes, an anode voltage of the order of 15 kV is needed for sufficient brightness in black and white picture tubes. This is known as HV or EHT (extra high tension) supply.

High voltage rectifier HV winding EHT (15–18 kV)

From horz osc

Horz deflection amplifier

To horz deflection coils

Boosted B+ supply

Fig. 8.12. Basic circuit of E.H.T. supply.

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To obtain such a high voltage by stepping up the mains voltage with a transformer is almost impossible and prohibitive in cost. A novel method used for obtaining EHT source is illustrated in Fig. 8.12. During retrace intervals of horizontal scanning, high voltage pulses of amplitude between 6 to 9 kV are developed across the primary winding of the horizontal ouptut transformer. As shown in the figure these are stepped up by an autotransformer winding to about 10 to 15 kV and then fed to a high voltage rectifier. The output of the rectifier is filtered to provide required dc voltage. Such an arrangement does not load very much the horizontal output stage because the current demand from this high voltage source is less than 1 mA. The horizontal output circuit is so designed, that in addition to providing EHT source, the energy stored in the horizontal deflection coils during retrace is tapped through a diode called damper diode to charge a capacitor. The voltage thus developed across the capacitor, actually adds 200 to 300 volts to normal B + voltage to give a boosted B + supply of 400 to 700 volts. This voltage is also suitable for first and second anodes of the picture tube. This arrangement makes the horizontal deflection circuit very efficient.

Review Questions
1. Draw block diagram of an RF Tuner (Front End) and explain how incoming signals from different stations are translated to common picture IF and sound IF frequencies. Illustrate your answer by choosing carrier frequencies of any channel in the VHF band. What do you understand by image rejection ratio ? Explain how by providing an RF amplifier, image signal reception is greatly minimized. What are the other merits of using an RF amplifier before the frequency converter ? Describe briefly the factors that influenced the choice of picture IF = 38.9 MHz and sound IF = 33.4 MHz in the 625-B monochrome television system. What are the essential functions which are assigned to IF section of the receiver ? Show by sketching output voltage verses frequency response of the IF section, how vestigial sideband correction is carried out. Why is the sound signal amplitude attenuated to about 5 per cent of the maximum output voltage ? Explain how composite video signal is detected ? How is the polarity of the video output signal decided ? Why is it dependent on the number of video amplifier stages ? What do you understand by intercarrier sound system ? Explain why any shift in the local oscillator freuency does not effect the frequency of the intercarrier beat signal. Where and how is the intercarrier sound signal separated from the video signal ? Draw a block diagram of the sound channel in a TV receiver. Explain briefly how the intercarrier sound signal as obtained at the video detector, is processed to produce sound output. Why is a de-emphasis circuit provided after the FM detector ? Explain briefly how sync pulses are separated from the composite video signal and processed to synchronize the vertical and horizontal oscillators. Describe briefly how EHT and boosted B + voltages are developed from the horizontal output circuit of the sweep amplifier.

2.

3. 4.

5. 6.

7.

8. 9.

10. Draw block diagram of a monochrome TV receiver and label its various sections Indicate by waveshapes the nature of signal at the input and output of each block of the receiver.

9
Television Signal Propagation and Antennas

9
Television Signal Propagation and Antennas
9.1 RADIO WAVE PROPAGATION
Radio waves are electromagnetic waves, which when radiated from transmitting antennas, travel through space to distant places, where they are picked up by receiving antennas. Although space is the medium through which electromagnetic waves are propagated, but depending on their wavelengths, there are three distinctive methods by which propagation takes place. These are: (a) ground wave or surface wave propagation, (b) sky wave propagation, and (c) space wave propagation. (a) Ground Wave Propagation Vertically polarized electromagnetic waves radiated at zero or small angles with ground, are guided by the conducting surface of the ground, along which they are propagated. Such waves are called ground or surface waves. The attenuation of ground waves, as they travel along the surface of the earth is proportional to frequency, and is reasonably low below 1500 kHz. Therefore, all medium wave broadcasts and longwave telegraph and telephone communication is carried out by ground wave propagation. (b) Sky Wave Propagation Ground wave propagation, above about 1600 kHz does not serve any useful purpose as the signal gets very much attenuated within a short distance of its transmission. Therefore, most radio communication in short wave bands up to 30 MHz (11 metres) is carried out by sky waves. When such waves are transmitted high up in the sky, they travel in a straight line until the ionosphere is reached. This region which begins about ‘120 km above the surface of the earth, contains large concentrations of charged gaseous ions, free electrons and neutral molecules. The ions and free electrons tend to bend all passing electromagnetic waves. The angle by which the wave deviates from its straight path depends on (i) frequency of the radio wave (ii) angle of incidence at which the wave enters the ionosphere (iii) density of the charged particles in the ionosphere at the particular moment and (iv) thickness of the ionosphere at the point. Figure 9.1 illustrates the path of several waves entering the ionosphere at different incident angles. With increase in frequency, the allowable incident angle at the ionosphere becomes smaller until finally a frequency is reached, when it becomes impossible to deflect the beam back to earth. For ordinary ionospheric conditions this frequency occurs at about 35 to 40 MHz. Above this frequency, the sky waves cannot be used for radio communication between distant points on the earth. This is why no frequencies beyond about 30 MHz (11 metres) are allotted for radio communication. 150

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151

Fig. 9.1. Ray paths for different angles of incidence (φ) at the ionosphere.

(c) Space Wave Propagation As explained above, propagation of radio waves above about 40 MHz (which is the beginning of television transmission band) is not possible through either surface or sky wave propagation. Thus, the only alternative for transmission in the VHF and UHF bands, despite large attenuation, is by radio waves which travel in a straight line from transmitter to receiver. This is known as space wave propagation. Its maximum range, because of the nature of propagation, is limited to the line of sight distance between the transmitter and receiver. For not too large distances, the surface of the earth can be assumed to be flat and different rays of wave propagation can reach the receiver from transmitter as shown in Fig. 9.2(a). As seen there, ht and hr, are the heights of transmitting and receiving antennas respectively, and d is the distance that separates them from each other. Both the direct wave AB and reflected wave ACB contribute to the field strength at the receiving antenna. Assuming the earth’s surface to be perfectly reflecting, the total field strength E, due to both direct and reflected waves, for reasonably large value of d can be expressed as: E* =

Fig. 9.2(a). Space wave propagation. For the sake of clarity, the antenna heights have been greatly exaggerated in comparison with the distance between them. *E =

Fig. 9.2(b). Computation of line-of-sight distance. The height of antennas has been greatly exaggerated in comparison with R, the radius of earth.

2 E0 2πfht hr sin but when d is large, as is often the case, the sine of the angle can be d d replaced by the angle and thus the above expression becomes 4πfht hr E= E0 d2

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where E0 is the field strength at unit distance from the transmitter, and f is the frequency of the transmitted signal. The field strength varies inversely as tbe square of the distance between the two antennas but is directly proportional to their heights. Various Aspects of Space Wave Propagation (i) Effect of Earth’s Curvature. Earth’s curvature limits the maximum distance between the transmitting and receiving antennas. This is depicted in Fig. 9.2(b). The maximum line of sight distance d between the two antennas can be easily found out. Neglecting (hr)2 and (ht)2, being very small as compared to R, the radius of the earth, the line-of-sight distance d ≈ 4.22( ht +
hr ) km.

where ht and hr are expressed in metres. In reality etectromagnetic waves are bent slightly as they glide along the surface of the earth and this increases the line-of sight distance by a small amount. It is evident that the ground coverage will increase with increase in height of both transmitting and receiving antennas. It is due to this reason that television transmitting and receiving antennas are placed as high as possible, for example atop tall buildings and on high plateaus. Another advantage is that the local noise disturbance pick-up is reduced by placing the antennas at high altitudes. (ii) Effect of Atmospherics. The presence of gas molecules and water vapour affects the dielectric constant and hence the refractive index of the atmosphere. As a result, the space waves are differently refracted or reflected under varying conditions of atmosphere. This under certain conditions enables the propagation to reach points very much beyond the line of sight. Similarly under adverse weather conditions the signal attenuation increases, thereby reducing effective distance of transmission. Occasionally the concentration of charged particles in the ionosphere increases sharply and it becomes possible for waves up to 60 MHz to return to earth. Though this enhances the range of sky wave propagation, but the exact time and place of occurrence of such phenomena cannot be predicted Thus this phenomenon has little value for commercial operation, but does explain to some extent the distant reception of high frequency and TV signals, which occurs at times under unusual conditions. (iii) Effect of Obstacles. Tall and massive objects like hills and buildings, will obstruct surface waves, which travel close to ground. Consequently, shadow zones and diffraction will result. For this reason in some areas antennas higher than those indicated by theoretical expressions are needed. On the other hand, some areas receive such signals by reflection only. Again, in some areas strong reflected signals are received besides direct signals. This can cause a form of interference known as ‘ghosting’ on the screen of a television receiver because of a phase difference between the two signals.

9.2

TELEVISION SIGNAL TRANSMISSION*

The RF carrier power output of commonly used VHF television transmitters varies from 10 to 50 kW. A satisfactory level of signal strength is said to exist when the image produced on the
*Television broadcast channels are given in Appendix C.

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screen of the receiver overrides noise by an acceptable margin. Signal strength is a function of power radiated, transmitting and receiving antenna heights, and the terrain above which the propagation occurs. The acceptable signal to noise ratio at the picture tube screen is measured in terms of peak-to-peak video signal voltage (half tone), injected at the grid or cathode of the picture tube versus the r.m.s. random noise voltage at that point. A peak signal to r.m.s. noise ratio of 45 db is generally considered adequate to produce a good quality picture. Field strength is indicated by the amount of signal received by a receiving antenna at a height of 10 metres from ground level, and is measured in microvolts per metre of antenna dipole length. The field strength for very good reception in thickly populated and built-up areas is 2500 µV/ metre for channels 2 to 4 (47 to 68 MHz), and 3550 µV/metre for channels 5 to 11 (174 to 223 MHz). For channels in the UHF band, a field strength of about 5000 µV/metre becomes necessary. This is so because of the lower sensitivity of the receiver for higher channels. Range of Transmission A sample calculation shows that for a transmitting antenna height of 225 metres above ground level the radio horizon is 60 km. If the receiving antenna height is 16 metres above ground level the total distance is increased to 76 km. Greater distance between antennas may be obtained by locating them on top of very tall buildings or hillocks. However, links longer than 120 km are hardly ever used for TV transmission because of limitations of radiated power, high channel frequencies and antenna heights. Thus, depending on the transmitter power and other factors the service area may extend up to 120 km for the channels in the VHF band but drops to about 60 km for UHF channels. Booster Stations Some areas are either shadowed by mountains or are too far away from the transmitter for satisfactory television reception. In such cases booster stations can be used. A booster station must be located at such a place, where it can receive and rebroadcast the programme to receivers in adjoining areas. Mussoorie (U.P.) is one such booster station. Its receiving and transmitting antennas are located on top of a hill. The station receives Delhi TV station (channel 4) programmes and relays them in channel 10 to the surrounding areas and regions on the other side of the hills.

9.3

INTERFERENCE SUFFERED BY CARRIER SIGNALS

In addition to thermal and man-made noise, the carrier signal must compete with various other forms of interfering signals originating from other television stations, radio transmitters, industrial radiating devices and TV receivers. When the interfering signal has a frequency that lies within the channel to which a TV receiver is tuned, the extent of interference depends only on the relative field strengths of the desired signal and the interfering signal. If the interfering signal frequency spectrum lies outside the desired channel, selectivity of the receiver aids in rejecting the interference. (a) Co-channel Interference Two stations operating at the same carrier frequency, if located close by, will interfere with each other. This phenomenon which is common in fringe areas is called co-channel interference. As the two signal strengths in any area almost equidistant from the two co-channel stations

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become equal, a phenomenon known as ‘venetian-blind’ interference occurs. This takes the form of horizontal black and white bars, superimposed on the picture produced by the tuned channel. These bars tend to move up or down on the screen. As the strength of the interfering signal increases, the bars become more prominent, until at a signal-to-interference ratio of 45 db or so, the interference becomes intolerable. The horizontal bars are a visible indication of the beat frequency between the two interfering carriers. Figure 9.3 shows the bar pattern that appears on the screen. The frequency of the beat note, which is equal to frequency separation between the two carriers, is usually of the order of a few hertz. This is so because most transmitters operate almost at the correct assigned frequencies. Motion of the bars, upwards or downwards occurs whenever the beat frequency is not an exact multiple of the field frequency. Co-channel interference was a serious problem in early days of TV transmission, when the channel allocation was confined to VHF band only. This necessitated the repetition of channels at distances not too far from each other. Now, when a large number of channels in the UHF band are available such a problem does not exist. The sharing of channel numbers is carefully planned so that within the ‘service area’ of any station, signals from the distant stations under normal conditions of reception are so weak as to be imperceptible. However, during a period of abnormal reception conditions (often during spring) when the signals from distant VHF stations are received much more strongly, co-channel interference can occur in fringe areas. The use of highly directional antennas is very helpful in etiminating co-channel interference.

(b) Adjacent Channel Interference Stations located close by and occupying adjacent channels, present a different interference problem. Adjacent channel interference (see Fig. 8.5) may occur as a result of beats between any two of these frequencies or between a carrier and any sidebands. A coarse dot structure is produced on the screen if picture carrier of the desired channel beats with sound carrier of the lower adjacent channel. The beat pattern is more pronounced if the lower adjacent sound carrier is relatively strong and is not sufftciently attenuated in the receiver. The next most prominent source of interference is the one produced by picture sideband components of the upper adjacent channel. The beat frequency between adjacent picture carrier is 7 MHz. Since this is far beyond the video frequency range, the resultant beat pattern is not discernible. However, the picture sidebands of the upper adjacent channel may beat with the desired channel carrier and produce an interfering image. To prevent adjacent channel interference, several sharply tuned band eliminator filters (trap circuits) are provided in the

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IF section of the receiver. This was explained in Chapter 8 while discussing desired IF response characteristics of the receiver. In addition to this, the guard band between two adjacent channels also minimizes the intensity of any adjacent channel interference. A space of about 150 km between adjacent channel stations is enough to eliminate such an interference and is normally allowed. (c) Ghost Interference Ghost interference arises as a result of discrete reflections of the signal from the surface of buildings, bridges, hills, towers etc. Figure 9.4 (a) shows paths of direct and reflected electromagnetic waves from the transmitter to the receiver. Since reflected path is longer than the direct path, the reflected signal takes a longer time to arrive at tke receiver. The direct signal is usually stronger and assumes control of the synchronizing circuitry and so the picture, due to the reflected signal that arrives late, appears displaced to the right. Such displaced pictures are known as ‘trailing ghost’ pictures. On rare occasions, direct signal may be the weaker of the two and the receiver synchronization is now controlled by the reflected signal. Then the ghost picture, now caused by direct signal, appears displaced to the left and is known ‘as leading ghost’ picture. Figure 9.4 (b) shows formation of trailing and leading ghost pictures on the receiver screen.

Reflecting surface or object Reflected path T Direct path

Ghost image

Direct image R Trailing ghost image Leading ghost image

Fig. 9.4 (a). Geometry of multiple path transmission.

Fig. 9.4 (b). Ghost interference.

The general term for the propagation condition which causes ghost pictures is ‘multipath transmission’. Ghost pictures are particularly annoying when the relative strengths of the two signals, vary such, that first one and then the other assume control of the receiver synchronism. In such cases the ghost image switches over from a leading condition to a trailing one or viceversa at a very fast rate. The effect of such reflected signals (ghost images) can be minimized by using directional antennas and by locating them at suitable places on top of the buildings.

9.4

PREFERENCE OF AM FOR PICTURE SIGNAL TRANSMISSION

At the VHF and UHF carrier frequencies there is a displacement in time between the direct and reflected signals. The distortion which arises due to interference between multiple signals is more objectionable in FM than AM because fhe frequency of the FM signal continuously changes. If FM were used for picture transmission, the changing best frequency between the multiple paths, delayed with respect to each other, would produce a bar interference pattern in the image with a shimmering effect, since the bars continuously change as the beat frequency

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changes. Hence, hardly any steady picture is produced. Alternatively if AM were used, the multiple signal paths can atmost produce a ghost image which is steady. In addition to this, circuit complexity and bandwidth requirements are much less in AM than FM. Hence AM is preferred to FM for broadcasting the picture signal.

9.5

ANTENNAS

In the preceding sections of this chapter, while explaining various methods of propagation, it was taken for granted that transmitters can somehow transmit and receivers have some means of receiving what is transmitted. Actually a structure must be provided, both for effective radiation of energy at the transmitter and efficient pick up at the receiver. An antenna is such a structure. It is generally a metallic object, often a rod or wire, that is used to convert highfrequency current into electromagnetic waves, and vice versa. Though their functions are different, transmitting and receiving antennas behave identically. Radiation Mechanism An antenna may be thought of as a short length of a transmission line. When high frequency alternating source is applied at its one end, the resulting forward and reverse travelling waves combine to form a standing wave pattern on the line. However, all the forward energy does not get reflected at the open end, and a small portion escapes from the system and is thus radiated. The electromagnetic radiation from the transmitting antenna has two components—a magnetic field associated with current in the antenna and an electric fleld associated with the potential. The two fields are perpendicular to each other in space and both are perpendicular to he direction of propagation of the wave. This is illustrated in Fig. 9.5. An electromagnetic wave is horizontally polarized if its electric field is in the horizontal direction. Thus an antenna fixed horizontally produces horizontally polarized waves. Similarly a vertical antenna produces vertically polarized waves.
x z y E E Dipole antenna H E-Electric field H-Magnetic field y H x z Direction of propagation

R.F. energy

Fig. 9.5. Transverse electromagnetic wave in free space.

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The amount of energy that is radiated in space by a transmission line antenna is however, extremely small, unless the wires of the line are suitably oriented and their lengths made comparable to the wavelength, If the two wires of the transmission line are opened up, there is less likelihood of cancellation of radiation from the two-wire tips and this improves the efficiency of radiation. This type of radiator is called a dipole. When total length of the two wires is halfwavelength, the antenna is called a half-wave dipole. Figure 9.6 shows the evolution of such an antenna from the basic transmission line. As shown there, the antenna is effectively a piece of quarter-wavelength transmission line bent out and open circuited at the far ends. Such a length has low impedance at the end connected to the main feeder transmission line. This in turn means that a large current will flow at the input of the half-wave dipole and efficient radiation will take place.

T.line l/2

l/2

(a) Transmission line

(b) Opened-out transmission line

(c) Conductors in line

(d) Half-wave dipole (centrefed)

Fig. 9.6. Evolution of the dipole.

With the help of Maxwell’s equations it is possible to deduce expressions for the energy radiated by an antenna, the direction or directions in which it propagates and the field strength at any distance from it.* The results show that the field strength depends on the power transmitted and is inversely proportional to the distance from the radiating source. The coefficient of current (I2rms) in the expression for the radiated power has the dimensions of resistance and is called radiation resistance. In effect, radiation resistance is the equivalent resistance which dissipates the same amount of power as that radiated from the antenna when same current flows through them. For a quarter-wave antenna the radiation resistance is 36.5 ohms, and for a half-wave antenna it is 73 ohms. Radiation Patterns of Resonant Antennas A resonant antenna is a transmission line whose length is an exact multiple of wavelengths and is open at both ends. The current distribution and radiation patterns of such resonant wires of different wavelengths which are remote from the ground are shown in Figs. 9.7 and 9.8. As seen there, for a λ/2 dipole the radiation is maximum at right angles to it, and eventually falls to zero in line with the antenna. This may be explained by considering that at right angles to the short length of the antenna the distance from a remote point to any part of the antenna wire is practically the same. Thus reinforcement of radiation will take place in this direction.
*Such analysis is beyond the scope of this book.

When the distant point lies in a direction other than normal, there will be some cancellation and finally full cancellation will take place at points that are in line with the antenna. Thus the radiation pattern cross section, as presented in Fig. 8.8 (a) is a figure of eight with its axis at right angles to the dipole. Moreover, exactly the same radiation pattern will exist in any other plane, and so the three-dimensional pattern is the surface of revolution obtained by rotating the cross section about an axis coinciding with the dipole. For an antenna of length equal to a whole wavelength the polarity of current, (as shown in Fig. 9.7 (b)) on one half of the antenna is opposite to that on the other half As a result, the radiation at right angles from this antenna will be zero because the field due to one half fully cancels the field due to the other half of the antenna. The direction of maximum radiation exists at 54° to the antenna. The pattern acquires lobes and there are four such for this antenna. As the length of the dipole is increased to three half-wavelengths, the current distribution is changed to that of Fig. 9.7(c) and the radiation pattern takes the shape shown in Fig. 9.8 (c). As the length of the aerial wire is further increased, the number of lobes in the radiation pattern increases, but the angle of largest lobe with the direction of antenna decreases. Nonresonant Antennas A nonresonant antenna (Fig. 9.9(a)) is one which is correctly terminated and as such only a forward travelling wave exists and there are no standing waves. As shown in Fig. 9.9(b) the
Voltage and current distribution Antenna L R (Terminating resistance) (a) (b) Feed Antenna wire

radiation pattern, though similar to the corresponding resonant antenna, is unidirectional. In fact there are half the number of lobes compared to the resonant antenna. This is due to absence of the reflected wave, which otherwise combines vectorially with the forward wave to create the radiation pattern. Ungrounded Antennas When the antenna is very close to the ground, its radiation pattern gets modified on account of reflections from the ground. If the ground is assumed to be a perfect conductor, a true mirror image of the actual antenna is considered to exist below the ground. The overall radiation pattern is then the sum of patterns caused by an array of two nearby antennas. Some typical radiation patterns for various heights above ground are shown in Fig. 9.10.

Grounded Antennas When one end of the antenna is actually grounded, the image of the antenna behaves as if it has been joined to the physical antenna and the combination acts as an antenna of double the size. The current distribution and radiation patterns of different earthed vertical antennas are shown in Fig. 9.11. As shown there, the voltage and current distribution on such a grounded λ/4 antenna, commonly known as the basic Marconi antenna, is the same as those of the ungrounded half-wave Hertz antenna. As is obvious Marconi antenna needs half the length as compared to Hertz antenna to produce the same radiation pattern.

Antenna Gain As explained above, all practical antennas concentrate their radiation in some direction, to a greater or lesser extent. Thus the field (power) density in the direction is greater than what it

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would have been if the antenna were omnidirectional. This may be interpreted that the antenna has a gain in a particular direction. The directive gain is thus defined as the ratio of the power density in the direction of maximum radiation to the power density that would be radiated by an isotropic antenna. The gain being a ratio of powers is expressed in decibels. Antenna Arrays It is clear from previous discussion that radiation from different types of antennas is not uniform in all directions. Though an antenna can be suitably oriented to get maximum response in any desired direction, additional directive gain in preferred directions can be obtained by using more than one radiator arranged in a specific manner in space. Such arrangements of radiators are known as antenna arrays. The simplest type of array consists of two antennas A1 and A2 separated by a distance d. A special case of directivity is obtained when d = λ/4 and the currents in the two antennas have a phase difference of 90° between them. This results in a cardioid shaped directional pattern as shown in Fig. 9.12.
d = l/4 A1 d A2

Fig. 9.12. Cardioid shaped directional pattern formed by parallel half-wave antennas. A1 and A2 are the locations of two antennas with currents 90° out of phase.

A broadside array consists of a number of identical radiators equally spaced along a line and carrying same amount of current in phase from the same source. As indicated in Fig. 9.13(a), this arrangement is strongly directional at right angles to the plane of the array.
Dipoles Radiation pattern Dipoles Radiation pattern Support Feed line l/4 l/4

Fig. 9.13 (a). Broadside array and pattern.

Fig. 9.13 (b). End-fire array and pattern.

Another arrangement known as end-fire array consists of a number of conductors equally spaced in a line (Fig. 9.13(b)), carrying same magnitude of current but with a progressive phase difference between them. The directional pattern of such an antenna has directivity along the array axis in the direction, in which antenna currents become more lagging. It is possible to combine several different arrays to obtain highly directional radiation patterns. Such combinations are often used in HF transmission/reception for point to point communication. Gains, well in excess of 50, are not uncommon with such arrangements.

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Folded Dipole As shown in Fig. 9.14 (a), the folded dipole is made of two half-wave antennas joined at the ends with one open at the centre where the transmission line is connected. The spacing between the two conductors is small compared with a half wave length. This antenna has the same directional characteristics and signal pick up as that of a straight dipole but its impedance is approximately 300 ohms. This is nearly four times that of a dipole, because for the same power applied, the antenna now draws half the current than it would have, in the case of a dipole. Hence the impedance (Z0) = 4 × 72 = 288 ohms for a half-wave dipole with equal diameter arms. This is generally considered as 300 ohms.
0.5l nominal 0.48l actual I/2 I/2 Transmission line (300W) l/2

I

Fig. 9.14 (a). Folded dipole antenna.

Fig. 9.14 (b). High impedance folded dipole antenna.

If elements of unequal diameter are used, or an additional closed conductor of the same diameter is added in between the two (Fig. 9.14(b)), an impedance as large as 650 ohms can be obtained. Parasitic Elements It is not necessary for all the elements of an array to be connected to the output of the transmitter. An element connected direct to the transmitter is called a driver, whereas a radiator not directly connected is called a parasitic element. Such a parasitic radiator receives energy through the induction field of a driven element, rather than by a direct connection to the transmission line. In general, a parasitic element longer than the driver and close to it reduces signal strength in its own direction, and increases it in the opposite direction. This in effect amounts to reflection of energy towards the driver and thus, this element is called a reflector. Again, a parasitic element shorter than the driver from which it receives energy, tends to increase radiation in its own direction, and is therefore called a director. The number of directors and their lengths can be varied to obtain increased directivity and broad band response.

9.6

TELEVISION TRANSMISSION ANTENNAS

As already explained, television signals are transmitted by space wave propagation and so the height of antenna must be as high as possible in order to increase the line-of-sight distance. Horizontal polarization is standard for television broadcasting, as signal to noise ratio is favourable for horizontally polarized waves when antennas are placed quite high above the surface of the earth. Turnstile Array To obtain an omnidirectional radiation pattern in the horizontal plane, for equal television signal radiation in all directions, an arrangement known as ‘turnstile array’ is often used. In

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this type of antenna two crossed dipoles are used in a turnstile arrangement as shown in Fig. 9.15(a). These are fed in quadrature from the same source by means of an extra λ/4 line. Each dipole has a figure of eight pattern in the horizontal plane, but crossed with each other. The resultant field in all directions is equal to the square root of the sum of the squares of fields radiated by each conductor in that direction. Thus the resultant pattern as shown in Fig. 9.15(b) is very nearly circular in the plane of the turnstile antenna. Fig. 9.15(c) shows several turnstiles stacked one above the other for vertical directivity.
A Individual antenna B Combined pattern

Patterns of individual antennas

90° lag

Fig. 9.15 (a). Turnstile array.

Fig. 9.15 (b). Directional pattern in the plane of turnstile.

l/2 l/2

l/2

l/2

Horizontal plane

Fig. 9.15 (c). Stacked turnstile array.

Dipole Panel Antenna System Another antenna system that is often used for band I and band III transmitters consists of dipole panels mounted on the four sides at the top of the antenna tower as shown in Fig. 9.16. Each panel consists of an array of full-wave dipoles mounted in front of reftectors. For obtaining unidirectional pattern the four panels mounted on the four sides of the tower are so fed that the current in each lags behind the previous by 90°. This is achieved by varying the field cable length by λ/4 to the two alternate panels and by reversal of polarity of the current.

Combining Network The AM picture signal and FM sound signal from the corresponding transmitters are fed to the same antenna through a balancing unit called diplexer. As illustrated in Fig. 9.17, the antenna combining system is a bridge configuration in which first two arms are formed by the two radiators of the turnstile antenna and the other two arms consist of two capacitive reactances. Under balanced conditions, video and sound signals though radiated by the same antenna, do not interfere with the functioning of the transmitter other than their own.

Antenna load east-west turnstile elements

Antenna load north-south turnstile elements

Balun Picture transmitter Reactance Reactance

Sound transmitter

Fig. 9.17. Equivalent bridge circuit of a diplexer for feeding picture and sound transmitters to a common turnstile array.

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9.7

TELEVISION RECEIVER ANTENNAS

For both VHF and UHF television channels, one-half-wave length is a practical size and therefore an ungrounded resonant dipole is the basic antenna often employed for reception of television signals. The dipole intercepts radiated electromagnetic waves to provide induced signal current in the antenna conductors. A matched transmission line connects the antenna to the input terminals of the receiver. It may be noted that the signal picked up by the antenna contains both picture and sound signal components. This is possible, despite the 5.5 MHz separation between the two carriers, because of the large bandwidth of the antenna. In fact a single antenna can be designed to receive signals from several channels that be close to each other. Antennas for VHF Channels Although most receivers can produce a picture with sufficient contrast even with a weak signal, but for a picture with no snow and ghosts, the required antenna signal strength lies between 100 and 2000 µV. Thus, while a half-wave dipole will deliver satisfactory signal for receivers located close to the transmitter, elaborate arrays become necessary for locations far away from the transmitter. Yagi-Uda Antenna The antenna widely used with television receivers for locations within 40 to 60 km from the transmitter is the folded dipole with one reflector and one director. This is commonly known as Yagi-Uda or simply Yagi antenna. The elements of its array as shown in Fig. 9.18(a) are arranged collinearly and close together. This antenna provides a gain close to 7 db and is relatively unidirectional as seen from its radiation pattern drawn in Fig. 9.18(b). These characteristics are most suited for reception from television transmitters of moderate capacity. To avoid pickup from any other side, the back lobe of the radiation pattern can be reduced by bringing the radiators closer to each other. The resultant improvement in the front to back ratio of the signal pick-up makes the antenna highly directional and thus can be oriented for efficient pick-up from a particular station. However, bringing the radiators closer has the adverse effect of lowering the input impedance of the array. The separation shown in Fig. 9.18(a) is an optimum value.
Reflector Director Radiation pattern l/10 0.55l l/10 0.45l

Driven element (a) (b)

Fig. 9.18. Yagi-Uda antenna (a) antenna (b) radiation pattern.

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Antenna Length As mentioned earlier, it is not necessary to erect a separate antenna for each channel because the resonant circuit formed by the antenna is of low ‘Q’ (quality factor) and as such has a broad band response. For the lower VHF channels (Band I—channels 2 to 4) the length of the antenna may be computed for a middle value. While this antenna will not give optimum results at other frequencies, the reception will still be quite satisfactory in most cases if the stations are not located far away. Though the antenna length used should be as computed by the usual expression: Wavelength (λ) =
3 × 10 8 metres, but in practice it is kept about 6 per cent less than the f (Hz)

calculated value. This is necessary because the self-capacitance of the antenna alters the current distribution at its ends. The small distance between the two quarter wave rods of the driver, where the lead-in line is connected can be taken as too small and hence neglected. Note that this gap does not affect the current distribution significantly. Antenna Mounting The receiving antenna is mounted horizontally for maximum pick-up from the transmitting antenna. As stated earlier, horizontal polarization results in more signal strength, less reflection and reduced ghost images. The antenna elements are normally made out of 1/4″ (0.625 cm) to 1/2″ (1.25 cm) dia aluminium pipes of suitable strength. The thickness of the pipe should be so chosen that the antenna structure does not get bent or twisted during strong winds or occasional sitting and flying off of birds. A hollow conductor is preferred because on account of skin effect, most of the current flows over the outer surface of the conductor. The antenna is mounted on a suitable structure at a height around 10 metres above the ground level. This not only insulates it from the ground but results in induction of large signal strength which is free from any interference. The centre of the closed section of the half-wave folded dipole is a point of minimum voltage, allowing direct mounting at this point to the grounded metal mast without shorting the signal voltage. A necessary precaution while mounting the antenna is that it should be at least two metres away from other antennas and large metal objects. In crowded city areas close to the transmitter, the resultant signal strength from the antenna can sometimes be very low on account of out of phase reflections from surrounding buildings. In such situations, changing the antenna placement only about a metre horizontally or vertically can make a big difference in the strength of the received signal, because of standing waves set up in such areas that have large conductors nearby. Similarly rotating the antenna can help minimize reception of reflected signals, thereby eliminating the appearance of ghost images. In areas where several stations are located nearby, antenna rotators are used to turn its direction. These are operated by a motor drive to set the broad side of the antenna for optimum reception from the desired station. However, in installations where a rotating mechanism is not provided, it is a good practice to connect the antenna to the receiver before it is fixed in place permanently and proceed as detailed below: (i) Try changing the height of the antenna to obtain maximum signal strength. (ii) Rotate the antenna to check against ghost images and reception of signals from faroff stations.

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(iii) When more than one station is to be received, the final placement must be a compromise for optimum reception from all the stations in the area. In extreme cases, it may be desirable to erect more than one antenna. Indoor Antennas In strong signal areas it is sometimes feasible to use indoor antennas provided the receiver is sufficiently sensitive. These antennas come in a variety of shapes. Most types have selector switches which are used for modifying the response pattern by changing the resonant frequency of the antenna so as to minimize interference and ghost signals. Generally the switch is rotated with the receiver on, until the most satisfactory picture is obtained on the screen. Almost all types of indoor antennas have telescopic dipole rods both for adjusting the length and also for folding down when not in use. Fringe Area Antenna In fringe areas where the signal level is very low, high-gain antenna arrays are needed. The gain of the antenna increases with the number of elements employed. A Yagi antenna with a large number of directors is commonly used with success in fringe areas for stations in the VHF band. As already mentioned, a parasitic element resonant at a lower frequency than the driven element will act as a mild reflector, and a shorter parasitic element will act as a mild ‘concentrator’ of radiation. As a parasitic element is brought closer to the driven element, then regardless of its precise length, it will load the driven element more and therefore reduce its input impedance. This is perhaps the main reason for invariable use of a folded dipole as the driven element of such an array. A gain of more than 10 db with a forward to back ratio of about 15 is easily obtained with such an antenna. Such high gain combinations are sharply directional and so must be carefully aimed while mounting, otherwise the captured signal will be much lower than it should be. A typical Yagi antenna for use in fringe areas is shown in Fig. 9.19 (a). In such antennas the reflectors are usually 5 per cent longer than the dipole and may be spaced from it at 0.15 to 0.25 wavelength depending on design requirements. Similarly the directors may be 4 per cent shorter than the antenna element, but where broadband characteristics are needed successive directors are usually made shorter (see Fig. 9.19 (a)) to be resonant for the higher frequency signals of the spectrum.
Reflector Antenna
2.12 m 2.41 m 1.2 m 2.46 m

In some fringe area installations, transistorised booster amplifiers are also used along with the Yagi antenna to improve reception. These are either connected just close to the antenna or after the transmission line, before the signal is delivered to the receiver. Yagi Antenna Design The following expressions can be used as a starting point while designing any Yagi antenna array. Length of dipole (in metres) ≈

143 ( f is the centre frequency of the channel) f (MHz)

Length of reflector (in metres) ≈ 152/f (MHz) Length of first director (in metres) ≈ 137/f (MHz) Length of subsequent directors reduces progressively by 2.5 per cent. Spacing between reflector and dipole = 0.25λ ≈ 75/f (MHz) Spacing between director and dipole = 0.13λ ≈ 40/f (MHz) Spacing between director and director = 0.13λ ~ 39/f (MHz) − The above lengths and spacings are based on elements of 1 to 1.2 cm in diameter. It may be noted that length of the folded dipole is measured from centre of the fold at one end to the centre of the fold at the other end. It must be remembered that the performance of Yagi arrays can only be assessed if all the characteristics like impedance, gain, directivity and bandwidth are taken into account together. Since there are so many related variables, the dimensions of commercial antennas may differ from those computed with the expressions given above. However, for single channel antennas the variation is not likely to be much. Figure 9.19 (b) shows a dimensioned sketch of channel four (61 to 68 MHz) antenna designed for locations not too far from the transmitter. It has an impedance = 40 + j20 Ω, a front to back pick up ratio = 30 db, and an overall length = 0.37 wavelength. Multiband Antennas It is not possible to receive all the channels of lower and higher VHF band with one antenna. The main problem in using one dipole for both the VHF bands is the difficulty of maintaining a broadside response. This is because the directional pattern of a low-band dipole splits into side lobes at the third and fourth harmonic frequencies in the 174 to 223 MHz band. On the other hand a high-band dipole cut for a half wavelength in the 174 to 233 MHz band is not suitable for the 47 to 68 MHz band because of insufficient signal pick-up at the lower frequencies. As a result, antennas for both the VHF bands generally use either separate dipoles for each band or a dipole for the lower VHF band modified to provide broadside unidirectional response in the upper VHF band also. Diplexing of VNF Antennas When it is required to combine the outputs from the lower and upper VHF band antennas to a common lead-in wire (feeder) it is desirable to employ a filter network that not only matches the signal sources to the common feeder but also isolates the signals in the antennas from each other. Such a filter arrangement is called a ‘diplexer’. Its circuit with approximate component

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values for bands I (47 to 68 MHz) and III (174 to 263 MHz) is given in Fig. 9.20 (a). The manner in which it is connected to the two antennas is shown in Fig. 9.20 (b). Similarly a triplexer filter can be employed when three different antennas are to feed their outputs to a common socket in the receiver. The combining arrangement can be further modified to connect the output from a UHF antenna to the same feeder line.
UHF band - III antenna VHF band - I antenna

Conical Dipole Antenna The VHF dual-band antenna pictured in Fig. 9.21 (a) is generally called a conical or fan dipole. As shown in Fig. 9.21 (b), this antenna consists of two half-wave dipoles inclined at about 30° from the horizontal plane, similar to a section of a cone. In some designs a horizontal dipole is provided in between the two half-wave dipoles. The dipoles are tilted by about 30° inward towards the wavefront of the arriving signal. This as shown in the figure results in a total included angle of 120° between the two conical sections in the broadside direction. A straight reflector is provided behind the conical dipoles.
Reflector Reflector 120° Dipoles Lead wire Mast Direction of wave travel (a) (b) l/2 l/4 30°

With the dipole lengths chosen for channel 2, this antenna is extensively used as a receiving antenna to cover both the VHF bands. The antenna resistance is about 150 ohms. The response pattern of the antenna contains only one major lobe on all the channels. This is so because for the 174-223 MHz band, the tilting of the dipole rods shifts the direction

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of the split lobes produced at the third and fourth harmonic frequencies so that they combine to produce a main forward lobe in the broadside direction. This is an improvement over the conventional dipole where an element cut for the low frequencies will have a multilobed pattern on the higher channels, and an element cut for the high frequencies will have a poor response on the lower channels. Though, one conical antenna array may be adequate for all VHF channels, sometimes three or four such arrays are stacked high for better and more uniform reception. In-line Antenna Another combination antenna which is known as in-line antenna is shown in Fig. 9.22. It consists of a half-wave folded dipole with reflector for the lower VHF band, that is in line with the shorter half-wave folded dipole meant for the upper VHF band. The distance between the two folded dipoles is approximately one-quarter wavelength at the high-band dipole frequency. This is the length of the line connecting the short dipole to the long dipole, where the transmission line to the receiver is connected. The directivity pattern of the in-line antenna is relatively uniform on all VHF channels, with a unidirectional broadside response. Its input resistance is about 150 ohms. UHF Antennas The basic principle of antennas for picking up signals from stations that operate in the UHF band is more or less the same as that in the VHF band. However, on account of higher attenuation suffered by the UHF signals, it becomes necessary to have very high gain and directive antennas. Besides this, higher gain is also necessary because receivers are less sensitive and tend to be more noisy at these frequencies than at lower frequencies. Therefore at microwave frequencies, some special type of antennas are used, in which the optical properties of reflection, refraction and diffraction are utilized to concentrate the radiated waves for higher directivity and more gain. Though a large number of microwave antennas for specific applications have been developed, the two types that find wide application for television reception are briefly described below.
Low-band folded dipole

Reflector

High-band folded dipole Directors

Fig. 9.22. In-line YAGI antenna array for lower and upper VHF bands.

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This di-fan half-wave dipole is the simplest type of UHF antenna as the basic Yagi is for the VHF band. As shown in Fig. 9.23, the dipoles are triangular in shape made out of metal sheet, instead of rods. This unit has a broad band response with radiation pattern resembling the figure of eight. When a screen reflector is placed at its back the response becomes unidirectional. For greater gain two or four sets of dipoles can be put together to form an array. Note the use of a mesh screen reflector instead of a rod as used in VHF antennas. Screen reflectors are more efficient than rods but their big size and bulk makes it impossible to use them in VHF antennas. Parabolic Reflector Antenna In this type of antenna (see Fig. 9.24) the dipole is placed at the focal point of a parabolic reflector. The principle is the same as that of parabolic reflectors of the headlights of a vehicle though in an inverse way. The incoming electromagnetic waves are concentrated by the reflector towards the dipole. This provides both high gain and directivity. Note that instead of using an entire parabolic structure only a section is used. The use of such a reflector provides a gain of 8 db over that of a resonant half-wave dipole.
Mesh-screen reflector

Reflector

Dipole

Specially designed dipoles

Lead-in wire

Fig. 9.23. Fan dipole UHF antenna.

Fig. 9.24. Parabolic reflector antenna.

In areas where both VHF and UHF stations are in operation, combination antennas serve to simplify reception problems from all the channels. Various combinations of different VHF and UHF antennas are in use. One such combination consists of a low-band conical antenna for VHF signals and a broad-band fan dipole for the microwave frequency region. A single lead-in line delivers signals to the receiver through the use of a special coupling device which is mounted directly on the antenna itself.

9.8

COLOUR TELEVISION ANTENNAS

The requirements to be met by colour television antennas are somewhat different than those for monochrome receivers. In monochrome receiver antennas, the emphasis is on higher gain

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while it may vary from channel to channel because of the wide frequency range. This, in itself, is no problem. In fact manufacturers generally design antennas to deliver more gain on higher frequency channels than on lower channels in order to compensate for higher transmission losses and lower receiver sensitivity at very high frequencies. However, in such antennas the gain not only varies from channel to channel but also from one end of the channel to the other. As an illustration Fig. 9.25 shows the response curve of a typical wide range monochrome antenna for channel 4, i.e., from 61 to 68 MHz. As shown there, the gain changes by about 4 db from beginning to end of the channel. If this antenna is used for colour TV reception from the same channel, the colour signal spectrum, which lies around 66.68 MHz will receive almost 2 db less gain than the video carrier and most of its sidebands. While it is true that the channel sound signal spectrum receives even lesser gain than the colour components but this does not affect the receiver reproduction. This is so because the picture contrast and sound volume controls provided in the receiver can be varied independently to get desired picture and sound outputs. However, the reduction in gain of colour signal frequencies cannot be separately compensated for and this results in poor colour picture quality. In colour television, the relative phase angle of the combined colour difference signal phasor determines the colour in the picture. Any change in gain is accompanied by a phase shift of the incoming signal. Thus a change in gain in the region of colour signal spectrum tends to change the hues in the picture. For example a slight shift of the colour signal phasor can turn red colour to orange and yellow to green. In fact, if a large phase shift occurs, no colours may get reproduced. Therefore the most important requirement of a colour TV antenna is a flat response over the entire channel. As labelled along the response curve in Fig. 9.25, the antenna output should not vary by more than one db over the frequency range of any one channel for satisfactory reproduction of colour details.
10 Permissible variation for colour reception

Log Periodic Antennas The stringent requirement of almost flat response besides high gain over any single channel has led to the development of a relatively new class of broadband antennas. The most popular of this type is the log periodic antenna. The name log periodic stems from the fact that the impedance of the antenna has a periodic variation when plotted against logarithm of frequency.

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Figure 9.26 (a) illustrates this periodic nature of the antenna impedance. Such a behaviour results from the geometric relationship between the relative lengths of the antenna elements and the distances which separate them from each other. This naturally results in the antenna getting larger and larger as the distance from the smallest element increases. The basic construction of a log periodic antenna consisting of a six element array is illustrated in Fig. 9.26 (b). As shown there, the largest dipole is at the back and each adjacent element is shorter by a fixed ratio typically 0.9. Also the distance between the dipoles becomes shorter and shorter by a constant factor which is typically 35 per cent of quarter wave spacing. As a result, the resonant frequencies for the dipoles overlap to cover the desired frequency range. All the dipoles are active elements without parasitic reflectors or directors. The active dipoles, as shown in the figure, are interconnected by a crossed wire net which transposes the signal by 180°.
Staggered feeder system

Impedance

log f

Transmission line

Fig. 9.26 (a). Periodic nature of the impedance of a log periodic antenna when plotted on a logarithmic scale.

Fig. 9.26 (b). Constructional details of a log periodic antenna.

When this antenna is pointed in the direction of the desired station, only one or two of the dipole elements in the antenna react to that frequency and develop the necessary signal. All the other elements remain inactive, i.e., do not develop any signal at that particular frequency. However, for any other incoming channel some other elements will resonate to develop the signal. Thus only one or two elements combine to deliver the signal from any one channel. Such an arrangement results in a uniform gain response over each channel. When the largest dipole is cut for channel 2, the array will cover all the low-band VHF channels as antenna resonance moves towards the shorter elements at the front. However, for the high-band VHF channels from 174 to 223 MHz the elements operate as 3 λ/2 dipoles. They are angled in as a ‘V’ to line up with the split lobes in the directional response for third harmonic resonance. Figure 9.27 shows such a log periodic antenna. When the largest dipole is cut for the lowest channel in the UHF band of 470 to 890 MHz, the array can cover all the UHF channels. The ‘V’ angle is not necessary in the UHF array because this frequency range is less than 2 : l. The UHF antenna array can be mounted along with the VHF array where a U/V splitter, i.e., a diplexer network connects the two antennas to a common transmission line for the downlead. It may be noted that the antenna described above is only one type of log periodic antenna out of a wide variety, quite different in appearance.

When colour transmission is to be received from only one channel, there is no need for a specially designed antenna. For example, the antenna designed for monochrome reception on channel four only can also be used with good results for receiving colour transmission from the same channel. However, the elements of the antenna must be cut and spaced accurately to ensure almost uniform gain over the entire channel. The antenna shown in Fig. 9.19 (b) can be used with colour receivers for receiving colour transmission from channel four.

9.9

TRANSMISSION LINES

A transmission line is used for delivering the antenna signal to the receiver. The desirable requirements of a transmission line are: (i) the losses along the line should be minimum. (ii) there should be no reflection of the signal on the line. (iii) the line itself should not pick up any stray signals. To prevent this the line should be balanced or shielded or both. The two main types ef transmission lines are the two wire parallel conductor type and the concentric (co-axial) type. Flat twin-lead, tubular twinlead, open wire line and co-axial type transmission lines are shown in Fig. 9.28. Flat-twin lead and tubular-twin lead transmission lines are constructed in the form of a plastic ribbon and are generally called twin-lead either flat or tubular. These together with the open wire line though balanced are not shielded lines. A line is balanced when each of the two conductors has the same capacitance (or voltage) to ground. The balance corresponds to the dipole antenna itself which has balanced signals of opposite phase in the arms. The connections of a balanced line between the antenna and receiver are shown in Fig. 9.29 (a). The balanced line is connected to the two ends of a centre-tapped antenna input transformer. Then any in-phase stray field cutting across both the wires of a balanced line will induce an equal voltage in each line. The consequent currents tend to induce voltages of opposite polarity in the centre tapped input transformer which cancel each other. However, the antenna signal from the dipole has opposite phases in the two sides of the line

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and the voltages that are induced in the input transformer reinforce each other and consequently a large signal voltage is delivered to the receiver from the secondary side of the input transformer. A shielded line, i.e., the coaxial cable is completely enclosed with a metal sheath or braid that is grounded to serve as a shield for the inner conductor. The shield prevents stray signals from inducing current in the centre conductor. Usually the shield is grounded to the receiver chassis. With only one conductor the line is unbalanced. The connections of a co-axial transmission line between the antenna and receiver are shown in Fig. 9.29 (b). Though the line is unbalanced, a balanced transformer (balun) can be used at the input of the receiver for converting the input signal from unbalanced to balanced form, if necessary. It may be noted that if the two inner conductors (insulated) are used within the shield then the line is both balanced and shielded. Shielded lines generally have more capacitance and higher losses. The attenuation is caused by I2R losses in the a.c. resistance of the line. This reduces the amplitude of the antenna signal delivered by the line to the receiver. The longer the line and higher the frequency, the greater is the attenuation. The characteristic impedance of the line which results from uniform spacing between the two conductors is the same regardless of length of the line.
Conductors Plastic insulation Plastic insulation (a) Flat twin lead (b) Tubular twin lead Braided conductor shield Insulating spacer Bare wire Inner insulation Inner conductor Conductors

Flat Twin-Lead The flat parallel wire is one of the most popular transmission lines in use for the VHF range. The wires are encased in a plastic ribbon of polyethylene which is strong, flexible and unaffected

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by sunlight, water or cold. The characteristic impedance ranges from 75 ohms to 300 ohms. As stated earlier this line is balanced. It is, however, unshielded and therefore not recommended for noisy locations. It should not be run close to power lines to avoid pick-up of 50 Hz hum. It should also be kept away from large metal structures which can alter the balance of the line. Since most receivers have a balanced input impedance of 300 ohms, the 300 ohms twin-lead (spacing about 1 cm, wire gauge 20 to 22) is convenient for impedance matching. The losses in a flat twin lead are much greater when the line is wet. Tubular Twin-Lead In this type the two parallel conductors are embedded in a polyethylene plastic tubing with air as dielectric for most of the inside area. Though expensive it has low losses and is especially suited for the UHF band of frequencies. The twin line is enclosed in a strong ptastic jacket for protection against adverse weather conditions. Open-wire Line As shown in Fig. 9.28 (c), this line is constructed with low loss insulating spacers between the parallel bare-wire conductors. The open wire line causes least attenuation because air is the dielectric between conductors. However, the characteristic impedance is relatively high. With a spacing of about 1.5 cm the impedance of this line is about 450 ohms. Co-axial Cable This line consists of a central conductor in a dielectric that is completely enclosed by a metallic shield which may be a tubing or a flexible braid of copper or aluminium. A plastic jacket moulded over the line provides protection. Because of the grounded shield the coaxial cable is immune to any stray pick-ups. With an outside diameter of about 1 cm the characteristic impedance is 75 ohms. Because of higher attenuation and higher costs, a shielded line is used only when the surrounding noise is very severe or where multiple lines must be run close to each other. In cable distribution systems, coaxial cable is a necessity despite its high losses. The losses in this system are compensated by the use of distribution amplifiers. Special connectors are available for terminating coaxial lines. Foam coaxial cable is also available. The use of foam as dielectric reduces the attenuation by about 20 per cent at 100 MHz. Characteristic Impedance When a transmission line has a length comparable with a wavelength of the signal, the characteristic impedance of the line depends on the small inductance of the conductors and the distributed capacitance between the conductors. It can be shown that the characteristic impedance Z0 =

L / C ohms, where L is the inductance per unit length and C the capacitance per unit length. The closer the conductor spacing, the greater is the capacitance and smaller the Z0 of the line.
Resonant and Non-resonant Lines When a line is terminated by a resistive load equal to Z0 of the line, there is no reflection and maximum power transfer takes place from source to load. Such a line is non-resonant because there are no reflections. A line terminated by Z0 then becomes effectively an infinitely long line because there is no discontinuity at the load. The length of the line is not critical. When such a line having Z0 = 300 ohms is connected to the 300 ohms antenna input terminals of the receiver, there is no reflection and maximum energy is delivered to the receiver from the

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antenna. Because of correct termination the line can be cut to any length without any loss in match of impedances. However, more length will produce more I2R losses. When the line is not terminated by Z0, there will be reflections and standing waves will be set up in the line. This effect makes the line resonant. The signal strength at any point on the transmission line will now be a function of the length of the transmission line unlike the case with non-resonant lines. The ratio of the voltage maximum to that of voltage minimum along the line is defined as the voltage standing wave ratio, abbreviated VSWR. Note that for a line terminated in Z0, the VSWR is one. In a resonant line, the greater the mismatch, the higher the VSWR which is greater than one. The extreme cases of high VSWR correspond to a short or an open terminated line. If one touches a resonant line, the added hand-capacitance can mean much more or much less signal delivery depending on where one touches the line. The use of transmission lines as resonant circuits is illustrated in Fig. 9.30. The maximum impedance is at the point of highest voltage on the line, say at the open end of an equivalent quarter-wave section of the line and minimum at the point of highest current, say at the short
I Shorted end I Shorted end

I Open end l/4 (a) Quarter-wave sections

I Open end l/2 (b) Half-wave sections

Fig. 9.30. Transmission-line sections as resonant circuits.

circuited end of an equivalent quarter-wave line. Note that the impedance at any point equals the ratio of voltage to current. As shown in the figure, a quarter-wave section shorted at the end is equivalent to a parallel-tuned circuit at the generator side because there is a high impedance across the terminals at the resonant frequency. For a line-length shorter than a quarter-wave, the line is equivalent to an inductance. The open quarterwave section provides a very low impedance at the generator side of the line. A line less than a quarter-wave makes the line appear as a capacitance. The half-wave sections however repeat the impedances at the end of the line to furnish the same impedance at the generator side. The main features of quarter-wave (λ/4) and half-wave (λ/2) sections are given in the table below.
Length Quarter-wave Quarter-wave Half-wave Hatt-wave Termination shorted open shorted open Input impedance Open circuit Short circuit Short circuit Open circuit Phase shift 90° 90° 180° 180°

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Such transmission lines are often called STUBS. A stub can be used (i) for impedance matching, (ii) as an equivalent series resonant circuit to short an interfering r.f. signal, and (iii) for phasing signals correctly in antennas. A quarter-wave line produces a phase change of 90° whereas a halt-wave section shifts the phase by 180°. To reduce interference, an open λ/4 stub at the interfering signal frequency can be used. One side is connected across the antenna input terminals of the receiver, while the open end produces a short at the receiver input one quarter-wave back. A 300 ohms twin-lead is designed to have almost a constant impedance for all the channels in a band. This is used to connect the antenna output to the input of the receiver. Matching the impedance of a multiband antenna to the characteristic impedance of the line is not critical because an impedance mismatch of 2.5 to 1 results in a one db loss of the signal. Quarter-wave Matching Section A quarter-wave section can be used for matching two unequal impedances Z1 and Z2. The characteristic impedance Z0 of the quarter-wave section should then be, Z0 =
Z1 Z2 . This is

iliustrated in Fig. 9.31, where an antenna with an impedance equal to 75 ohms is matched to a 300 ohms transmission line by a quarter-wave matching section. This section should then have an impedance equal to Z1 Z2 = 75 × 300 = 150 ohms. The required length of the quarterwave section can be estimated using the expression λ/4 (metres) = v

Fig. 9.31. Use of a quarter-wave section for matching antenna to transmission line.

Balun (Balancing Unit) Two quarter-wave sections of the type discussed above can be combined to make a balancing and impedance transforming unit. This is illustrated in Fig. 9.32 (a). This is called a Balun and is used for matching balanced and unbalanced impedances. Two quarter-wave lines each having an impedance equal to 150 ohms are connected in parallel at one end, resulting in 75 ohms impedance across points A and B. Either A or B can be grounded to provide an unbalanced impedance at the ungrounded point with respect to ground. At the other end the two 150 ahms quarter-wave sections are connected in series to provide 300 ohms impedance between points C and D. The quarter wavelength of the line isolates the ground point from C or D, allowing a balanced impedance with respect to ground. Either side of the Balun can be used for input

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with the other side as output. It is usefully employed for matching a 72 ohms coaxial line to the 300 ohms receiver input or in the reverse direction, i.e., a 300 ohms twin-lead to a 72 ohms unbalanced input. As illustrated in Fig. 9.32 (b) bifilar windings are uscd to simulate the 150 ohms transmission line sections. The windings, as illustrated in Fig. 9.32 (c), are on a ferrite core to increase the inductance, thus making the line electrically longer.
A 75W unbalanced 3 B 4 (a) A B Ferrite core (c) C D Twin wire 1 2 6 300W balanced 7 8 D B 75W unbalanced 3 4 (b) 5 C A 1 2 6 Bifilar windings 7 8 D 300W balanced 5 C

Review Questions
1. 2. Describe briefly the different methods by which radio waves of different frequencies are propagated. Why space wave propagation is the only effective mode of radiation above about 40 MHz ? What do you understand by line-of-sight distance in space wave propagation ? What are the effects of atmospherics and obstacles on space waves ? Why is it necessary to keep both the transmitting and receiving antennas as high as possible for television ? What do you understand by wave polarization ? Why is horizontal polarization preferred for television and FM broadcasts ? Why is TV transmission limited to about 100 km ? What are booster stations and under what conditions are they employed ? Describe briefly co-channel and adjacent channel interference effects. Discuss the techniques employed to eliminate such interference in fringe areas. What is a ghost image and what causes it to appear on the receiver screen along with the reproduced picture ? Differentiate between leading and trailing ghost pictures. Why is AM preferred over FM for picture signal transmission ? Describe briefly radiation mechanism from an antenna. Explain the evolution of a dipole for effective radiation. Sketch approximate radiation patterns for ungrounded resonant antennas of lengths λ/4, λ/2 and 3λ/2 and justify them. Define directional gain and front to back ratio as applied to receiving antennas. What is an antenna array ? Sketch radiation patterns for (i) a broadside array and (ii) end-fire array. How are these patterns modified when the antennas are very close to the ground ? What is the function of a reflector and a director in a Yagi antenna. Explain why such antenna configurations are highly directional. What is the effect of increasing the number of director elements ? What are the special requirements of a fringe area television antenna and how are these achieved ? Give constructional details of a typical fringe area antenna and explain the precautions that must be taken while mounting it. Give constructional details of a turnstile antenna and explain by drawing radiation pattern its suitability for television transmission. Draw the circuit of a diplexer arrangement employed for feeding the output from both picture and sound signal transmitters to the same antenna. Why is it not possible to use the same antenna for reception for both lower and upper VHF channels ? Describe any one type of multiband array commonly employed to cover all the channels in the VHF band. Describe briefly the basic principle of bow-tie (di-fan) and parabolic reflector type of antennas commonly employed for reception from UHF television channels. Explain why an antenna used for colour TV reception must deliver almost constant output voltage over any one channel. Sketch a typical log periodic antenna and explain its special characteristics. Why are its elements bent in a ‘V’ shape ? Describe with suitable sketches various types of lead-in wires used for connecting the antenna to the TV receiver. What is the essential difference betwecn balanced and unbalanced lines and how are they connected to the receiver ? Why is a coaxial cable preferred for connecting a UHF antenna ? What is a stub ? Explain how quarter-wave line sections can be used for providing an impedance match between a low impedance antenna and 300 ohms lead-in line. What is a ‘Balun’ ? Give its constructional details and explain how it can be used as an impedance matching network between two different impedances at high frequencies. Under what conditions does it become necessary to use an attenuator pad between the transmission line and receiver ? Draw ‘H’ and ‘T’ pad configurations and explain how besides providing a match between the line and receiver the desired attenuation is also achieved.

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10
Television Applications

10
Television Applications
Television, by its use in broadcasting has opened broad new avenues in the fields of entertainment and dissemination of information. The not-so-well-known applications are in the area of science, industry and education, where the television camera has contributed immeasurably to man’s knowledge of his environment and of himself. The television camera is probably best described as an extension of the human eye because of its ability to relay information instantaneously. Its capability to view events occurring in extremely hazardous locations has led to its use in areas of atomic radiation, underwater environments and outer space. Some of its important applications which are of direct interest to our society are described briefly in this chapter.

10.1 TELEVISION BROADCASTING
Broadcasting means transmission in all directions by electromagnetic waves from the transmitting station. Broadcasting, that deals mostly with entertainment and advertising, is probably the most familiar use of television. Millions of television sets in use around the world attest to its extreme popularity. Most programmes produced live in the studio are recorded on video tape at a convenient time to be shown later. Initially television transmission was confined to the VHF band only but later a large number of channel allocations were made in the UHF band also. The distance of transmission, as explained earlier, is confined to line of the sight between the transmitting and receiving antennas. The useful service range is up to 120 km for VHF stations and about 60 km for UHF stations. Television broadcasting initially started with monochrome picture but around 1952 colour transmission was introduced. Despite its complexity and higher cost, colour television has become such a commercial success that it is fast superseding the monochrome system.

10.2 CABLE TELEVISION
In recent years master antenna (MATV) and community antenna (CATV) television systems have gained widespread popularity. The purpose of a MATV system is to deliver a strong signal (over 1 mV) from one or more antennas to every television receiver connected to the system. Typical applications of a MATV system are hotels, motels, schools, apartment buildings and so on. The CATV system is a cable system which distributes good quality television signal to a very large number of receivers throughout an entire community. In general, this system feeds 182

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increased TV programmes to subscribers who pay a fee for this service. A CATV system may have many more active (VHF and UHF) channels than a receiver tuner can directly select. This requires use of a special active converter in the head-end. (a) MATV The block diagram of a basic MATV system is shown in Fig. 10.1 (a). One or more antennas are usually located on roof top, the number depending on a available telecasts and their direction. Each antenna is properly oriented so that all stations are received simultaneously. In order to allow a convenient match between the coaxial transmission line and components that make up the system, MATV systems are designed to have a 75 Ω impedance. Since most antennas have a 300 Ω impedance, a balun is used to convert the impedance to 75 ohms. As shown in the figure, antenna outputs feed into a 4-way hybrid. A hybrid is basically a signal combining linear mixer which provides suitable impedance matches to prevent development of standing waves. The standing waves, if present, result in ghosts appearing in an otherwise good TV picture. The output from the hybrid feeds into a distribution amplifier via a preamplifier. The function of these amplifiers is to raise the signal amplitude to a level which is sufficient to overcome the losses of the distribution system while providing an acceptable signal to every receiver in the system. The output from the distribution amplifier is fed to splitters through coaxial trunk lines. A splitter is a resistive-inductive device which provides trunk line isolation and impedance match. Coaxial distribution lines carry television signals from the output of splitters to points of delivery called subscriber tap-offs. The subscriber taps, as shown in Fig. 10.1 (b), can be either transformer coupled, capacitive coupled or in the form of resistive pads. They provide isolation between receivers on the same line thus preventing mutual interference. The taps look like ac outlets and are normally mounted in the wall. Wall taps may be obtained with 300 Ω output 75 Ω output and a dual output. The preferred method is to use a 75 Ω type with a matching transformer. The matching transformer is usually mounted at the antenna terminals of the receiver and will have a VHF output and a UHF output. Since improperly terminated lines will develop standing waves, the end of each 75 Ω distribution cable is terminated with a 75 Ω resistor called a terminator. (b) CATV Formerly CATV system were employed only in far-fringe areas or in valleys surrounded by mountains where reception was difficult or impossible because of low level signal conditions. However, CATV systems are now being used in big cities where signal-level is high but all buildings render signals weak and cause ghosts due to multipath reflections. In either case, such a system often serves an entire town or city. A single antenna site, which may be on top of a hill, mountain or sky-scraper is chosen for fixing antennas. Several high gain and properly oriented antennas are employed to pick up signals from different stations. In areas where several signals are coming from one direction, a single broad based antenna (log-periodic) may be used to cover those channels. Most cable television installations provide additional services like household, business and educational besides commercial TV and FM broadcast programmes. These include news, local sports and community programmes, burgler and fire alarms, weather reports, commercial data retrieval, meter reading, document reproduction etc. Educational

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services include computer aided instructions, centralized library services and so on. Many of the above options require extra subscription fee from the subscriber.

Since several of the above mentioned service need two-way communication between the subscriber and a central processor, the coaxial distribution network has a large number of cable pairs, usually 12 or 24. This enables the viewer to choose any channel or programme out of the many that are available at a given time. CATV Plan. Figure 10.2 shows the plan of a typical CATV system. The signals from various TV channels are processed in the same manner as in a MATV system. In fact, a CATV system can be combined with a MATV set-up. When UHF reception is provided in addition to VHF, as often is the case, the signal from each UHF channel is processed by a translator. A translator is a frequency converter which hterodynes the UHF channel frequencies down to a VHF channel. Translation is advantageous since a CATV system necessarily operates with lengthy coaxial cables and the transmission loss through the cable is much greater at UHF than at VHF frequencies. As in the case of MATV, various inputs including those from translators are combined in a suitable mixer. The set-up from the antennas to this combiner is called a head-end.
FM channels From VHF and UHF antenna systems Local programme

Head end

FM band amplifier

TV

TV

TV

TV

TV

TV modulator

Amplifiers and translators

Combining network Coaxial cable Amplifier

Equalizer

Level adjustment Trunk lines to distant locations Amplifiers

Trunk lines to distant locations

Distribution amplifier

Splitter

Splitter

Tap-off points

R

R

R

R

R

R

Termination resistor 75W

Fig. 10.2. A simplified block diagram of a CATV system.

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Further, as shown in the figure the CATV outputs from the combiner network are fed to a number of trunk cables through a broadband distribution amplifier. The trunk cables carry signals from the antenna site to the utilization site (s) which may be several kilometres away. Feeder amplifiers are provided at several points along the line to overcome progressive signal attenuation which occurs due to cable losses. Since cable losses are greater at higher frequencies it is evident that high-band attenuation will be greater than low-band attenuation. Therefore, to equalize this the amplifiers and signal splitters are often supplemented by equalizers. An equalizer or tilt control consists of a bandpass filter arrangement with an adjustable frequency response. It operates by introducing a relative low-frequency loss so that outputs from the amplifiers or splitters have uniform relative amplitude response across the entire VHF band. The signal distribution from splitters to tap-off points is done through multicore coaxial cables in the same way as in a MATV system. In any case the signal level provided to a television receiver is of the order of 1.5 mV. This level provides good quality reception without causing acompanying radiation problems from the CATV system, which could cause interference to other installations and services.

10.3 CLOSED CIRCUIT TELEVISION (CCTV)
Closed circuit television is a special application in which camera signals are made available only to a limited number of monitors or receivers. The particular type of link used depends on distance between the two locations, the number and dispersion of receivers and mobility of either camera or receiver. Figure 10.3 illustrates various link arrangements which are often used. The simplest link is a cable where video signal from the camera is connected directly through a cable to the receiver. A television monitor, which is a receiver, without RF and IF circuits, is only required for reception in such a link arrangement. About one volt peak-to-peak signal is required by the monitor. Since the video signal is normally delivered via cables and even when transmitted, it is over a limited region and for restricted use, CCTV neede not follow television broadcast standards.
Video amplifier Camera Cable

(d) Output of a remotely controlled camera feeding several TV receivers located at a distance

Fig. 10.3. Commonly used closed-circuit television (CCTV) systems.

CCTV Applications There are numerous applications of CCTV and a few are briefly described here. (i) Education. One instructor may lecture to a large number of students sitting at different locations. Similarly close-ups of demonstration experiments and other aids can be shwon on monitors during these lectures. (ii) Medicine. Several monitors and camera units can be installed to observe seriously ill patients in intensive care units. In medical institutions, operations when performed can be shown to medical students without their actually gathering around the oepration table. (iii) Business. Television cameras can be installed at different locations in big departmental stores to keep an eye over customers and sales personnel. (iv) Surveillance. In banks, railway yards ports, traffic points and several other similar locations, closed circuit TV can be effectively used for surveillance. (v) Industry. In industry CCTV has applications in remote inspection of materials. Observance of nuclear reactions and other such phenomena would have been impossible without television. Similarly television has played a great role in the scanning of earth’s surface and probing of other planets. (vi) Home. In homes a CCTV monitor finds its application in seeing the caller before opening the door. (vii) Aerospace and Oceanography. Here a wireless link is used between the transmitter and receiver. In some applications camera is remotely controlled over a microwave radio link. As shown in Fig. 10.3 (c), for aerospace and oceanography a carrier is used for transmitting the signal and a complete receiver is then necessary for reception.

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10.4 THEATRE TELEVISION
Television programmes can be shown to a large audience in theatres. Similarly cinematographic films can be telecast for viewing on television receivers. Some examples of such applications are as follows. (i) TV Programmes in Theatres Special programmes can be shown on a large screen by optical projection in a theatre where spherical mirrors and reflectors are used to enlarge the image. With about 80 KV on the final anode of the picture tube, there is enough light to show pictures on a standard theatre screen. The same idea can be used for projecting TV programmes at home on a small screen. (ii) Film Recorders Film recorders produce a cinematographic film by photographing a television picture displayed on the screen of a picture tube. For doing so the film has to be pulled down frame by frame during successive blanking intervals. A video tap recording can only be rebroadcast in countries where the same TV standards are in use, whereas film recordings offer a ready means of exporting programmes to other countries using different standards. (iii) Telecine Machines and Slide Projectors Many television programmes originate from 35 mm and 16 mm photographic cinema films. Slides are also often used in TV programmes. Therefore, telecine machines and slide projectors form part of the television studio equipment for transmitting motion pictures and advertisement slides. Telecine machines are cinema projectors equipped with mirror or prism reflector arrangement for focusing pictures, as produced by them, on the face of a TV camera. Slide scanners also have a similar optical arrangement for transmitting still from different slides. For high utilization of the projectorcamera chain, an optical multiplexer is often used. This switches or directs one of the several optical image sources to the lens of a single camera, thus enabling the use of one TV camera for receiving programmes from three or four film and slide projectors. For the accompanying sound signal pick-up, the usual optical or magnetic track playback facility is incorporated in the multiplexer setup. An additional problem of using telecine projectors is the difference in frame rate of motion pictures and television scanning. Motion pictures are taken at the rate of 24 frames/sec but while screening, each frame is projected twice to reduce flicker. This amounts to an effective frame rate of 48/sec. However, in TV transmission, while the frame rate is 25, the field rate is 50 on account of interlaced scanning. Thus, there is a difference of one picture frame/sec between the two and if not corrected, would cause a rolling bar on the raster besides loss of some signal output. In order to overcome this discrepancy, the film in the telecine projector is pulled down by the shutter mechanism at the rate of 25 frames/sec. It is achieved by a suitable speed correction in the drive mechanism. This naturally results in a little faster movement of scenes and objects on the television screen but the distortion caused is so small that it is hardly noticeable. The corresponding small increase in the pitch of reproduced sound also goes undetected. Similarly in the 525 line system where the frame rate is 30/sec, it becomes necessary to suitably modify the film rate of 24 to achieve compatibility between the two in order to prevent rolling and loss of any picture information. The necessary correction is carried out by projecting

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one frame three times and the following frame two times. This sequence is repeated alternately by means of an intermittent shutter mechanism of 3 : 2 pull-down cycle for the film. The pulldown is carried out by a shutter which operates at the rate of 60 pulls per second. Thus, out of a total of 24 frames, 12 are projected three times while the rest only two times. This then makes the film rate of 24 frames equal to the scanning rate of (12 × 3 + 12 × 2 = 60) 60 fields per second. The speed alteration and sequencing explained above makes direct scanning of motion pictures possible through any TV network. While slides are used for stills, small advertisement films are recorded beforehand on video tapes for telecasting when required. (iv) Pay Television Special programmes like first-run films, sports events and cultural programmes that are normally not broadcast on the usual TV channels because of their higher cost are made available to television subscribers either through the cable television network or on special channels. At the receiver a special decoder is used to receive the picture. The decoder also has the provision to register an extra charge for such special screenings. This service is optional but a fixed charge is made for initial installation.

10.5 PICTURE PHONE AND FACSIMILE
This is another fascinating application of television where two people can see each other while talking over the telephone line. A picture phone installation includes a unit that contains a small picture tube and a miniature TV camera. The highest modulating frequency in picturephone services is normally limited to about 1.5 MHz. Facsimile is another application of electronic transmission of visual information, usually a still picture, over telephone lines. Since there is no motion, a slow scanning rate is employed. Facsimile is employed for sending copies of documents over telephone lines.

10.6 VIDEO TAPE RECORDING (VTR)
Video tape recording was introduced in 1956 and it proved to be a vast improvement over the earlier method of recording motion pictures taken from the screen of the television receiver. Video tape retains the ‘live’ quality of broadcasting and has the capability of being edited and duplicated without any delay. The other advantages are (i) immediate playback capability, (ii) convenience of repeating the recorded material as many times as the viewer wishes, and (iii) ease of duplication for distribution to a large number of users. The video signal can be recorded on a magnetic tape for picture reproduction in a similar way as the audio tape is used for reproduction of sound. In an audio recorder, the plastic tape (mylar) that has a very fine coating of ferric oxide is made to move in physical contact with the tape head. Any electrical signal applied to the tape head magnetizes the magnetic particles on the tape, as it passes across the head. For each cycle of the signal, two tiny bar magnets are produced on the tape and the length of bar magnets is inversely proportional to the frequency of applied signal. Thus, on a recorded tape these bar magnets form a chain with like poles adjacent to each other. When a recorded tape moves across the playback head, there is a

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change in flux linkages with the head and hence a voltage is developed across its coil terminals. This technique of recording and reproducing audio signals is illustrated in Fig. 10.4.
Tape width Gap length Bar magnets H
N N
S

Output

Tape head
S
N

v v

One wave Recording length head

N

S Tape motion v0

N N

S S

N

0

t

0

t

Input signal

Output signal

Fig. 10.4(a) Electrical signal recorded in the form of bar magnets on the magnetic tape.

Fig. 10.4(b) Development of signal during playback.

Audio Range Each head has a certain gap length. As the recording frequency increases, the length of the bar magnets decreases. A limiting frequency is reached when at a given tape speed, total length of the two adjacent bar magnets becomes equal to the gap length. At this frequency, output of the playback head will be zero since each bar magnet will produce equal and opposite voltage in the coil, with the result that no net flux passes through core of the head. For a tape speed of 19 cm (7.5″) per second and with a gap length of 6.3 microns (0.00025″) the usable frequency comes to about 15 KHz and is enough for audio recording. Audio Signal Dynamic Range Since output voltage from a playback head is directly proportional to the rate of change of flux, for every doubling of frequency the output voltage will become twice. In other words, every time the frequency gets halved (one octave lower), the output falls by 6 db. Assuming that the entire audio range occupies 9 octaves, output at frequencies that lie in the lowest octave will be 54 db below the output in the highest octave. This discrepancy is got over by providing equalizing circuits in the playback amplifier. The equalizing network is designed to have characteristics where the output voltage falls by 6 db per octave to allow for the rising response at the playback head. AC Bias If the signal to be recorded is applied directly to the record head, the output will be highly distorted on account of non-linearity of B-H curve of the core material around its zero axis. This difficulty is solved by superimposing the recording signal on a high frequency ac voltage. The amplitude of the high frequency bias is so chosen that its positive and negative peaks lie within the linear portions of the B-H curve. As shown in Fig. 10.5, the two outputs (marked X and Y) add up to give linear output with improved signal to noise ratio. The ac bias frequency

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is kept fairly high so that the beat signal between the highest signal frequency and bias frequency does not fall in the audio range. An ac bias frequency close to 60 KHz is considered adequate and is normally employed in audio tape recorders.

Video Recording The above introduction to audio recording will enable a better appreciation of the special problems of recording video signals on a magnetic tape. Video Frequencies For recording up to the highest video frequency of 5 MHz if the head gap is kept at 0.00025″ (6.3 microns) a tape speed of the order of 39 metres (1300 inches) per second would be necessary. With the head fixed as in audio recording and moving the tape at such a high speed would result in excessive wear and tear besides mechanical instability. Decreasing the gap below about 6 microns to lower the tape speed is not possible because of technological problems. However, it is not essential that only tape should move and head remains stationary. It is the relative speed of tape and head which is responsible for the output voltage. Hence the relative speed is increased by moving the tape head in opposite direction to the tape movement. In practice tape heads are mounted on the periphery of a drum which rotates around an axis relative to the direction of tape movement. This enables reduction in tape speed to as low as 7.5 inches (≈ 20 cm) per second. Video Signal Dynamic Range As described earlier, output voltage from a playback head is directly proportional to the rate of change of flux. In case of video frequencies consisting of dc to 5 MHz, theoretically there would be infinite number of octaves. Even if frequencies down to 25 Hz are required to be retained (dc can be recovered by clamping), the band between 25 Hz to about 5 MHz would occupy nearly 17 octaves. This means that the average output on account of frequencies in the lowest octave will be about 100 db below the output in the highest octave. It is very difficult to handle such a large dynamic range where low frequencies run into noise levels. The octave problem could be solved by translating video frequencies to higher frequencies by amplitude modulation. For example, a carrier frequency of 10 MHz would provide sidebands in the range of 5 to 15 MHz. This covers only two octaves. However, amplitude modulation is not used because of the following reasons:

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(i) At the speeds used in video recording, it is difficult to keep good contact with the head and this results in amplitude variations. Such amplitude variations are reproduced as noise. (ii) In order to avoid distortion it is necessary to use an ac bias having a frequency atleast four times the carrier frequency. However, such a high frequency (4 × 10 MHz = 40 MHz) is not desirable on account of high frequency heating and other such losses. Keeping in view the above drawbacks FM is used for video recording. A carrier frequency of about 6 MHz is often employed. Its amplitude is kept quite large thus making use of any ac bias unnecessary. FM is insensitive to small amplitude variations. However, if present due to improper contact between the tape and head, such variations can be removed by the use of amplitude limiters. The dynamic range with FM recording is also quite small despite the fact that it has a wider bandwidth. This is so because FM sideband frequencies which have significant amplitudes occupy only a few octaves. An additional advantage of FM is that it becomes possible to record and transmit even the dc component of the video signal. Audio signals which accompany the scene are recorded on the same tape by a separate head and played back by normal audio tape recording and playback techniques. Scanning Methods The two methods commonly used for video recording are called transverse scanning and helical scanning. In transverse scanning four quadruplex heads are used for recording and reproduction while in helical scanning either one or two heads are employed. Transverse (Quadruplex-head) Scanning. Transverse scanning, though expensive is superior, and is therefore preferred in professional video tape recorders. Figure 10.6 (a) illustrates
Audio erase head Video heads Record head

the unidirectional path of the tape past a rotating head wheel which has 4 record/reproduce heads affixed to it at accurately spaced 90 degree intervals. Thus the heads rotate transversely across the tape while it is pulled slowly from one spool to another. Each head comes in contact with the tape as the previous one leaves it. The net result is a large relative motion between the head gaps and tape surface. In order to achieve good contact between the tape and head (see Fig. l0.6 (b)) the tape is made to move in a curvature. The curvature is of the shape of head travel and this is given to the tape by a vacuum pump arrangement to ensure that each head maintains a proper fixed pressure contact with the oxide surface. Because of tape movement and relative arc of the recording heads, the recorded track is somewhat slanted (see Fig. 10.6(c)) in the direction of tape travel. At the top and bottom edges of the tape, sync and audio signals are recorded respectively. A switching arrangement transfers signal to the active head (the head which is in contact with the tape) at the appropriate moment. A small guard band appears between any two slanted recording tracks on the tape. A full track erase head is used before the tape goes to the drum that carries the recording heads. Synchronization. While recording, the completion of one video track must correspond to one field of picture scanning. Similarly during replay the video head must track it accurately otherwise reproduced picture will get severely distorted. Therefore speed regulation of the tape mechanism is very critical. The control signals recorded at the edges of the tape should be synchronized and thus locked with head rotation. This is done by servo control methods. The television signal to be recorded has its own pulse train but this cannot be used directly on the tape since it needs drastic changes before use for synchronizing and control of drive speeds. Servo Control System. A signal in the form of pulses is generated for each rotation of the video head. This as shown in Fig. 10.7 is done either by a bulb and a photocell or by a small magnet and coil combination. The drum has a small slit through which light from a bulb falls
Head motor shaft

on a photocell once during each revolution of the drum. In the second method, a small magnet fixed on the drum, induces a small voltage pulse as it passes over a fixed coil once during each revolution. The drum pulses thus generated are compared with the incoming field sync pulses obtained during recording. The comparator produces an error signal corresponding to the difference in phase and frequency between the two signals. This is amplified and applied to eddy current brakes provided on the head motor. The brakes adjust the motor speed to provide necessary synchronization. On replay the drum pulses are compared with 50 Hz pulses derived from supply mains. The error signal thus derived from the comparator is used for controlling speed of the head motor. In addition to this the drum pulses are also used to ensure that the head runs accurately at the centre of the recorded track. Maximum video output is the indication of correct video head tracking. The use of four recording heads together with switching arrangements, a vacuum system for good tape contact with recording heads and elaborate synchronizing facilities contribute to the high cost of this system. However, it is justified because professional VTRS which employ quadruplex system of recording with a full bandwidth of 5 MHz providc such good quality pictures that they look like a live programme. Helical Scan Recording. Smaller and low-priced video tape recorders employ 0.5″ or 1″ tape and provide a bandwidth up to about 2.5 MHz. Such recorders normally use one head and a relatively simple drive mechanism. In this system of recording the tape is wrapped around a drum inside which the head rotates. The video head protrudes through a horizontal slit in the drum to come in contact with the tape. Figure 10.8(a) shows the mechanical layout of this method of recording. The video head rotates at 50 revolutions per second such that one field of picture information is recorded in one revolution. As shown in the figure, the tape comes in contact at the upper edge of the drum and leaves it at its bottom edge. Thus the recorded track is in the form of a helix and this gives it the name of helical scanning. While a single video head is used for both recording and playback, the tape passes before a full track erase head before it goes around the drum. A typical one inch tape format is shown in Fig. 10.8(b).
Takeup spool Rotating drum Video head Rotating drum Tape head

It consists of an audio track and guard band at the top and audio cue, guard and control tracks at the bottom. The video tracks across the tape are at an angle of 5° to the tape length axis separated by 4 mil (0.004″) guard bands. Quadruplex Head Recording and Playback Circuits. Figure 10.9 shows the basic block schematic arrangements of recording and playback in a quadruplex head VTR. In record mode the composite video signal of about 1 V P-P amplitude, as obtained from a camera set up, feeds into a two stage wide-band video amplifier in the video tape recorder. The output from this amplifier is fed to the FM modulator via a pre-emphasis network and driver.
Video input Video amp Sync separator and shaper Servo control cct Audio input Head drum motor drive Ultrasonic biasing network Pre-emphasis circuit Driver amp FM modulator Recording head amp Slipring brushes Four video heads

Fig. 10.9(b) Simplified block diagram of a quadruplex head video tape system in the playback mode.

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The modulator output is amplified by the recording head amplifier and fed to the record heads through slip-ring brushes. The accompanying audio signal is amplified, given an ac bias, and then fed to the audio head. As shown in Fig. 10.9 (a) a sync separator circuit is also connected across the output of the wide-band video amplifier. The sync pulses, after separation, are suitably shaped and used to synchronize and control the head drum motor speed by servo control techniques. During playback (see Fig. 10.9 (b)) the video head outputs are collected through slipring brushes. These are then fed to an electronic switcher, which selects and amplifies the signal from the head that is in contact with the tape. The selection takes place during horizontal retrace blanking intervals so that the switching transients are not visible. The selected output is fed to the FM demodulator through an equalizing network and several limiter stages. The detected output is amplified by a four stage video amplifier before feeding it to a monitor or an amplitude modulator. After modulation with the carrier frequency of any one of the band I channels, the incoming programme can be viewed on a TV receiver. The speed control signals are recovered from the corresponding tape tracks and processed for driving the capstan motor and synchronizing the head drum motor drive. The audio signal, as obtained from the audio playback head, is amplified and fed to the monitor or a modulator for use in a TV receiver. Video Disc Recorder. Optical video disc recording is a recent development. It uses a laser beam in its pick up system. The video disc is 12″ (301.6 mm) in diameter and is made of transparent plastic. Since its reverse surface is coated with aluminium to reflect the laser beam, it has an appearance of a metallic disc. Sound and image signals are stored in tiny pits located in a substrate l.l mm from the surface. There are close to 14 billion pits on one surface of a disc. The reflection of the laser beam from the disc is intermittently interrupted according to the distribution and width of these pits and the reflected laser beam is converted into electrical signals. Since there is no friction from a stylus as in the case of conventional audio pickups, the sound and image quality of the laser beam type video disc is almost permanent. The rotational speed of the disc is so controlled that the relative beam velocity is constant from the outermost edge to the innermost end. One revolution of each track forms one frame of picture and one side of the disc can record up to 54000 frames. Both, one hour and half an hour duration dises are now available. In the later type of video disc, sound is FM modulated and recorded in two separate channels. Therefore, the two channels may be used for recording and reproduction of bilingual programmes or high fidelity stereo music.

10.7 TELEVISION VIA SATELLITE
The conventional methods for extended coverage of TV by microwave space communication and coaxial cable links are relatively expensive. Geostationary communication satellites launched into synchronous orbits around the earth in recent years have enabled not only national but also international television programmes to be relayed between a number of ground stations around the world. Three artificial satellites placed in equatorial orbits at 120° from each other cover practically the whole populated land area of the world. High power, highly directive land based transmitters transmit wideband microwave signals to the geostationary satellite above the transmitter. Each microwave channel has a bandwidth of several tens of megahertz and can accommodate many TV signals and thousands

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of telephone channels or suitable combinations of these. The satellites usually powered by solar batteries receive the transmission, demodulate and amplify it and remodulate it on a different carrier before transmitting again. The transmitting antenna on the satellite, by the use of a suitable reflector, can direct the radiated beam to a narrow region on the earth and economize on power to provide a satisfactory service in the desired area. Higher power satellites can provide large power flux densities so that smaller size antennas can be used for reception. For national distribution the transmission is downwards from a wide angle antenna so that the whole national area is ‘illuminated’ by the transmission if possible. For international distribution the transmission is also towards the other one or two satellites (which are in line of sight direction) from highly directive antennas. The demodulation-amplification-remodulation transmission process is repeated in the second satellite. The final ‘down channel’ transmission is received (in the same or a different country) by a large cross-section antenna and processed in low noise receivers and finally reradiated from the regular TV transmitters. There are a number of ‘INTELSAT’* satellites over the Atlantic, Pacific and Indian Oceans operating as relay stations to some 40 ground stations around the world. The international system of satellite communication caters to the continental 625/50 and the American 525/60 systems. As television standards differ from country to country, the transmitting station adopts the standards of the originating country. The ground station converts the received signal with the help of digital international conversion equipment to the local standards before relaying it. Frequency modulation is used for both ‘up channel’ and ‘down channel’ transmission. FM, though it needs a larger bandwidth, offers good immunity from interference and requires less power in the satellite transmitter. Frequency Allocation The frequency bands recommended for satellite broadcasting are 620 to 790 MHz, 2.5 to 2.69 GHz, and 11.7 to 12.2 GHz on a shared basis with other fixed and mobile services. The satellite antenna size and the RF power naturally depend upon the frequency of operation. Space erectable antennas are used for the 620-790 MHz band, with the size limited to about 15 metres, while rigid antennas are used both for 2.5 and 20 GHz bands, the size being limited to about 3 metres. For the ground terminal, the maximum diameter of the antenna is restricted by the allowable beamwidth and frequency. The cost and complexity of the receiver increases with increase in frequency. Extended Coverage of Television Besides the use of satellites for international TV relaying, satellites can be used for distributing national programmes over extended regions in large countries because of their ability to cover large areas. For this, satellites can be used in three following ways. (i) Rebroadcast System. In this system emission from a low power satellite is received with the help of a high sensitivity medium size (about 9 m) antenna on a high sensitivity low noise earth station. The received satellite programme is rerelayed over the high power terrestial transmitter for reception on conventional TV receivers. This method is suitable for metropolitan
*INTELSAT stands for INternational TELecommunication SATellite-consortium.

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areas where a large number of TV receivers are in operation. This method also enables national hook-up of television programmes on all distantly located television transmitters. For a large country like India, to do so by microwave or coaxial cable links would be expensive. (ii) Limited Rebroadcast System. In rural areas where clusters of villages and towns exist, and the receiver density is moderate, low power transmitters can be used to cover the limited area. This reduces the ground segment cost by eliminating the need for special frontend equipment, dish antennas and convertors for each receiver. The SITE (Satellite Instructional Television Experiment) programme conducted by India in cooperation with NASA of USA in 1975-76 was a limited rebroadcast system. A high power satellite ATS-6 (Application Technology Satellite) positioned at a height of 36,000 km, in a geostationary synchronous orbit with sub-satellite longitude of 33° East was used for beaming TV programmes over most parts of the country. TV programmes from the earth station at Ahmedabad were transmitted to ATS-6 at 6 GHz FM carrier with the help of a 14 m parabolic dish antenna. The FM carrier had a bandwidth of 40 MHz. The ‘down transmission’ from the satellite was done from a 80 W FM transmitter at 860 MHz. The transmitted signal consisted of a video band of 5 MHz and two audio signals frequency modulated on two audio subcarriers of 5.5 MHz and 6 MHz. This enabled transmission in two different languages and reception of any one of these. A block diagram of this system is shown in Fig. 10.10.

(iii) Direct Transmission. Direct reception of broadcast programme is the only possibility in areas remote from terrestial broadcast stations. In this system the cost of reception is very high even with high power satellite transmissions. This is so because of the need for a special antenna to receive the signals and front-end convertor unit to modify the signals into conventional broadcast standards. Advances in technology reducing the cost of low noise frontend for rcceivers may make direct individual reception feasible in the near future. Japan was the first country to launch a medium scale satellite (‘YURI’) in April 1978 for experimental purposes towards direct reception. It radiates two colour channels in the 12 GHz band. The additional receiver equipment consists of 1 to 1.6 metres parabolic dish antenna and a frontend convertor to feed UHF-AM TV signals to the conventional receiver. Cost of Satellite Communication The cost of satellite communication would be very much lower than it is but for the limited life of the satellite. The life is limited because a geosynchronous satellite using high gain antenna requires close control of both its position and altitude in ‘Orbit’. The position and altitude control rockets require fuel that has to be put in once for all before launch. Thus for a given payload, the longer the life the heavier is the satellite and correspondingly expensive. All communication satellites are therefore designed for a maximum operating life limited by its positioning fuel capacity. This of course has an advantage too. Successive generations of communication satellites can incorporate the latest developments in electronics and communication technology, packing much more capacity into satellites of comparable size. It is noteworthy that the cost per channel of one hop satellite communication has decreased over the last decade by a factor of more than ten. We can therefore hope that with advances in technology direct reception at reasonable costs will become a reality in a not too distant future.

10.8 TV GAMES
Television games is a relatively new application of digital electronics and IC technology to TV products. The first of the solid-state games used Transistor-Transistor Logic (TTL). The earlier set-ups provided logic for playing question answer games on the television screen. Later paddle type games were developed which included generation of sounds to give a touch of reality to the game being played. Then colour was added to the display and the challenge of contest amongst players increased by programming the game electronics to adapt to the player’s skills. Now with the development of microprocessors (µP) further sophistication has become possible, where, for example in card games, players can compete against the computer and against each other. Though logic can be developed to play almost any game but most common and commercially available games include tennis, soccer, squash and rifle shooting. The receiver screen shows the game in progress and also displays its current score. As the game proceeds the score display is updated properly. As an illustration, if the game being played is tennis, the screen (see Fig. 10.11) will show the court lines, the net, the rackets and the ball. The movement of the ball is fully shown as it is hit from one side to the other. In addition, individual scores are displayed on both sides of the court. Suitable sounds are generated when any contact occurs and these are reproduced on a loudspeaker in synchronism with the action. In some games like football, players are also shown along with movements of the ball.

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TV screen

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Court boundary

Ball

Net

Racket 4 2 Racket

Fig. 10.11. Tennis game display on a receiver screen.

Functional Organization of TV Games Any television game consists of two parts-the game unit and TV receiver. The game unit contains complete electronic circuitry necessary for generating signals, which when fed to the television receiver display the current status of the game on the screen. Functionally a TV game unit consists of three sub-units or blocks. These are: (i) player or user’s control unit, (ii) game cum control circuit logic unit and (iii) RF oscillator and modulator section. Figure 10.12 shows these blocks or units and also the manner in which the modulated game video signal is fed to the receiver.

TV receiver

Antenna switch

TV game unit Player(s) or user(s) controls Game and control logic RF osc and modulator

Fig. 10.12. Functional blocks of a TV game system.

The player control block contains various controls available to players for playing the game. In addition, it has the associated circuitry for generating corresponding command signals to initiate various actions. The game and control logic section is the heart of any TV game. It produces video signals necessary for displaying game characters and game field on the screen. The characters may be simple paddles, bats, rackets or complex figures representing men, women and other objects necessary for the game. The interface circuitry for both player and game-action control forms part of the character and field video generation circuits. This section also provides logic circuits for game-playing rules, score display and totalling during the game. The control and logic section also inserts sound signals at appropriate points in the game by pulse detection gating. 1n addition to all this it has a sync generation section which develops both horizontal and vertical sync pulses needed to time the composite video signals correctly.

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The composite video signal which contains full game and sync information feeds into the VHF oscillator-modulator block. The oscillator is set for channels 3 or 4 and its output after modulation provides input signal for the TV receiver. It is normally fed to the receiver input terminals through a special antenna isolation switch. The output signal level from the oscillatormodulator unit is kept low to avoid any interference to other television sets operating in the vicinity. Development of TV Game Circuits Earlier TV game units, built with TTL consisted of different general purpose chips interconnected on a printed circuit board. These games being simple in nature needed a limited number of printed circuit boards. However, with time, the number of games offered by the same game unit increased and this made the units more complex. The corresponding complexity of electronic circuits required to implement these games made the TTL hardware too bulky and unmanageable. Accordingly manufacturers of TV games have now switched over to the use of dedicated (special purpose) ICs and microprocessors for designing all types of complex games and their logic circuits. Dedicated ICs for TV Games Single customer built LSI chips are now available which contain almost all the circuitry which goes into the making of a game unit. Besides a modulator, a clock-generator and a regulated power supply, such ICs need only a few discrete components like resistors and capacitors to produce multi game units. TV game industry has made big strides during the past decade and a large number of dedicated n-channel MOS chips, both for 625/50 scan (PAL-Colour) and 525/ 60 scan (NTSC-Colour) systems are now available. These include a choice of ball and paddle games with true game rules, realistic courts, and individual player identification. The battle games offer all the thrilis and excitements of real battle scenes. The organization of a TV game unit employing a specially built IC is illustrated in Fig. 10.13. As shown there, all outputs (command signals) from the user’s panel and clock generator chip feed into this IC. The various outputs from the IC are combined in a video summer unit to form a composite video signal for feeding onto the modulator unit.
Clock generator User’s panel and command circuits Control logic and circuits in the game chip (IC)

Fig. 10.13. Block diagram of a TV game system employing a dedicated IC chip.

In order to illustrate various functions performed by such ICs, a 28 lead dual-in-line dedicated IC package is shown in Fig. 10.14. It contains logic and controls for six selectable games which can be played by one or two persons with vertical paddle motion. The games

The ball video signal is output at this pin. (i) Input set to logic ‘1’ selects a low speed of ball motion (ii) Input set to logic ‘0’ selects a higher speed of ball motion

8.

Manual service

With input logic at ‘1’ the game stops after each service. However, when logic is switched to ‘0’, the circuitry changes to automatic service mode Right Player (RP) and Left Player (LP) video outputs are available at these pins As shown in the figure an R-C network connected to each of these pins enables vertical position control of the paddle/ball through a 10 K potentiometer. (i) Logic 1—large paddle/bat size (ii) Logic 0—small paddle/bat size No connection Standard horizontal and vertical sync and blanking pulses aie available at this pin. The output of the master clock (chip) is fed at this pin (Frequency = 2 MHz) To select a particular game the corresponding switch (see figure) is set for logie ‘0’ i.e. connected to VSS. Other game selection switches remain at logic ‘1’ i.e. open circuit. The score and field output video signals are available at this pin. The input switch is momentarily connected to Vss (logic 0) to reset the score counters and to start a new game. Normally this pin connection stays at logic ‘1’. This input is driven by a positive pulse indirectly obtained from the user’s panel to indicate a ‘shot’ This pin is also driven by a positive pulse triggered by the shot input if the target is hit. No connection

9, 10 11, 12

Player outputs Right and Left Paddle/ ball position (location)

13 14, 15 16 17 18 to 23

Bat size NC Sync output Clock input Game Selection

24 25

Score and field output Game Reset

26 27 28

Shot input Hit input NC

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which can be played are tennis, soccer, squash, practice (one man squash) and two rifle shooting games. It features automatic on-screen score display from 0 to 15, sound generation for hit, boundary and service, selectable paddle size, ball speed, two different rebound angles, automatic or manual ball service and visually defined areas for all ball games. The video signal output is suitable for black and white display on a standard domestic TV receiver. The functions of various pin connections on this IC (see Fig. 10.14) are as follows.
1 11 10K pot 25 Right paddle/ball location Game reset 14 NC 15 28 Clock input 17 From clock generator

Fig. 10.14. Block diagram of a TV game dedicated IC. (Note: The pin numbers and their locations are arbitrary).

TV Games with Colour Display Initially, designers of TV games were hesitant to provide colour display because of the complications of colour signal generation and modulation. However, now with the availabilityof separate ICs for such purposes, addition of colour needs little more than adding one or two

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integrated circuits to the schematic. National Semiconductor’s LMI 889 is one such IC which accepts luminance (brightness), syne, chrominance (colour) and audio inputs and produces an RF modulated composite video signal. This IC includes two RF oscillators which are tuned to VHF low-band channels 3 and 4. Either output can be selected by applying a voltage to the external R-L-C tank circuit. The sound oscillator is isolated from the rest of the IC, and can be externally frequency modulated with a varactor diode or by switching a capacitor across the tank circuit. The crystal controlled colour subcarrier oscillator feeds two chroma modulators with quadrature signals for generating (B-Y) and (R-Y) colour-difference signals. Two RF modulators then add video, chroma and sound to the selected carrier frequencies. Microprocessor (µP) Controlled TV Games Some of the available games have a µP as the basic control element and use plug in ROM (Read Only Memory) cartridges to store game sets. Many side benefits are accrued by incorporating a µP into a particular video game. Most obvious of these are the versatility of the design, the multiplicity of games available and the ease with which a particular game design can be modified. A µP controlled game may include, as its functional components, a microprocessor, Random Access Memory (RAM), Read Only Memory (ROM), Cassettes or other secondary magnetic storage medium besides a key board or other player control blocks, a video interface, a modulator and course a receiver as the display unit. A block schematic of such a system is shown in Fig. 10.15. A brief description of the various blocks follows: (i) Display Unit. The television receiver is used as the display unit. The receiver handles the signal fed to it in the usual and displays appropriate patterns at desired positions on its screen. The rate of display is made fast enough to maintain the illusion of continuity as is the normal practice in television broadcasts. The display on the screen is organised by the games system in a standard format which is usually 150 rows × 250 columns. (ii) Player Controls/Keyboard and Interface. This unit provides a link between the system and players. The knobs on the keyboard are moved to initiate various actions. For instance in a tennis gams the rate and direction of displacement of the control knob will decide how quickly and in which direction the bat will move for hitting the ball. The outputs from the keyboard are analog in nature. The associated interface accepts the serial analog inputs and converts them into parallel digital form for processing by the µP (microprocessor). Similarly other command knobs generate appropriate analog signals which are necessary for a particular game. (iii) Memory—RAM (Random Access Memory) and ROM (Read Only Memory). As shown in the figure the memory consists of two blocks—RAMs and ROMs. The RAMs store information temporarily which continuously updated or changed on receipt of commands from the players. The ROMs store fixed instructions for the µP for its processing the received data and sending out command signals to associated units. The ROMs also contain instructions for generating video signals for fixed patterns. For example in a football game, ROM stores the dot pattern of a football. When the game is in progress, it feeds this data continuously to the µP along with the information it receives about speed and direction of motion of the ball.

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(iv) Row and Column Counters. The receiver screen is divided into 150 rows and 250 columns. This is done to determine the position of the object on the screen. The row and column counters are used to determine the row and column position of any object. To display a particular object at a certain place on the screen, the counters are suitably set by the µP. The counters in turn feed digital signals to the PVI (Programmable Video Interface) for generating corresponding video signals. For example, to give the impression of a horizontal movement of any object, the µP on receipt of such information from the memory, continuously changes the column counter to indicate the next horizontal position. Accordingly the column counter continuously feeds digital command signals to the PVI for generation of varying video signals to flash the object at its correct location on the screen.
Memory Cassette and Interface ROM and Interface RAM and Interface

(v) Programmable Video Interface (PVI). This unit generates video and audio signals on receipt of digital commands from the µP both directly and indirectly. The µP continuously feeds the PVI with dot pattern output of various objects and indicates their motion and location through Row and Column counters. The PVI also receives input from the clock generator to produce audio signals to indicate various sound outputs at appropriate moment. The sync pulses are also added here to form a composite video signal. (vi) Sound Generator. In order to make playing of TV games more realistic, suitable sounds are generated for various events Iike service, rebounds, shots, etc. The audio signals are processed within the PVI. The µP is programmed to control a square wave generator (clock generator) whose fundamental frequency, f0, is usually about 7800 Hz. For each different occurrence a special bit (= 1) is set on. An 8-bit number (n) in the PVI determines the type of sound to be generated like a rattle, collision, a shrill whistle etc. This as stated above is determined by the µP and depends on the nature of game and the occurrence of different sounds while the game is in progress. The PVI on receipt of a particular 8-bit number sends an

206 audio signal whose frequency is given by f=

MONOCHROME AND COLOUR TELEVISION

7800 f0 = Hz (n + 1) (n + 1)

The sound output can have a frequency as low as 30 Hz. The audio signal is either fed directly to a separate loudspeaker or is reproduced on the receiver loudspeaker in the usual way. (vii) Microprocessor (µP). The heart of the TV games system is the microprocessor. It processes the data fed to it and generates digital output which is used by the PVI to generate proper video and audio signals continuously. The memory unit (RAMs and ROMs) inform the µP how to process the data and send appropriate command signals. For example in a tennis game, as the player moves the control knob in an effort to hit the ball it generates a particular control signal. On receipt of this information via the memory unit, the µP decides whether row and column locations of the racket and ball overlap or not (for a contact) at the same instance. lf contact is made the direction of the ball is reversed. The ball’s dot structure remains the same while its address location changes continuously to show the ball in motion. However, if no contact is made the µP decides whether the point where the ball landed (X, Y co-ordinates) are within the acceptable areas, that is, inside the court or outside it and accordingly gives credit. The output signal is also used to update the score on the screen. Similarly the µP decides the nature of’ the sound to be produced and sends a command signal for its generation. The above explanation is a simplified view of what actually goes on in the system. The actual process involved is more complex and requires detailed and complicated programming. (viii) Cassette Recorder and Interface. A large number of games can be stored in cassettes. These are fed to the memory via a suitable interface. The cassette output is analog and the interface converts it into digital form. This allows the RAMs to readily store the contents of the required ‘game’ from the cassette player. When a different game is desired, the tape is advanced or rewound till the desired game appears on the tape head. Information about one or two games is permanently stored in the memory unit and thus these can be played without any input from an external cassette receiver. (ix) VHF Modulator. The video and audio outputs from the PVl together with sync pulses can be fed directly to the receiver through a cable. However, in modern systems these signals modulate VHF carriers of a particular channel (usually 2 or 3) as is normally done in a TV transmitter. The modulated output is fed to the receiver input terminals via a coaxial cable. In some designs the modulated signal is radiated through a small antenna. The receiver antenna intercepts the radiated signal and processes it in the usual way to reproduce visual display on the screen and sound in the loudspeaker. References
1. 2. Riley, M.P. ‘Video Tape Recording’, Television August 75, Vol. 25 No. 10. Sept. 75, Vol 25, No. 11. Oct. 75, Vol. 25, No. 12. Nov., 75 Vol. 27, No. 1. Wentworth, John W., ‘The Technology of Program Production and Recording’ Proc. 1RE, 50 (No. 5) (May 1962) 830-836.

Review Questions
1. Draw the block diagram of a MATV system and explain how television signals are picked up from several stations and distributed to various locations in an apartment building or hotel. How is the impedance match maintained at different subscriber tap points ? Describe the main merits and applications of a CATV system. Draw a typical layout of this system of signal distribution and label all the blocks. Why are amplifiers and equalizers required along trunk distribution lines ? How is a CCTV system different from regular TV broadcasts ? Enumerate various applications of this system of television. Describe with suitable block diagrams various methods employed to feed/transmit video signal to different monitors/receivers. Discuss special problems of video tape recording and explain haw these are overcome for recording video signals on a magnetic tape. What is a basic difference between transverse and helical scan recording ? Explain with suitable diagrams the basic difference between these two methods of video recording. Draw block diagrams illustrating record and playback modes of a quadruplex head VTR system. Label all the blocks and explain sequence of operations both for recording and playback. How is scanning and speed synchronization achieved in such a recording and reproduction system ? Describe briefly various systems which can be employed for distributing national television programmes by a satellite over extended regions in large countries like India. What is the function of a special front-end used along with TV receivers for direct reception from a satellite ? Describe with a block diagram the functional organization of a TV game set-up and explain the use of dedicated ICs for processing and control of analog signals generated at the user’s panels. Draw a schematic block diagram of a TV games set-up which employs a microprocessor for processing input data and generating digital outputs for the PVI.

2.

3.

4.

5.

6.

7. 8.

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11
Video Detector

11
Video Detector
From antenna to the input of video detector of a television receiver, it is all radio frequency circuitry, and is similar to a superhetrodyne AM radio receiver. It is at the video detector that picture signal is extracted from the modulated intermediate carrier (IF) frequency. This, after detection is amplified and fed to the picture tube cathode or grid circuit for reproduction of the picture. The sound signal which is frequency modulated with the sound IF carrier frequency is translated in the detector to another carrier frequency, which is the difference of the picture IF and sound IF, i.e., 38.9 – 33.4 = 5.5 MHz. The intercarrier FM sound signal thus obtained is separated at this stage or after the first video amplifier. It is then amplified and detected before feeding to the audio section of the receiver.

11.1 VIDEO (PICTURE) SIGNAL DETECTION
The video detector is essentially a rectifier cum high frequency filter circuit to recover video signal from the modulated carrier. Semiconductor diodes are used exclusively for detection and need about 2 volts or more of IF signal for linear detection without distortion. The signal to the detector is fed from the output of last IF amplifier stage. Either polarity of this signal can be rectified by suitably connecting the diode, since both sides of the modulated envelope have the same amplitude variations. This choice depends on the number of video amplifier stages used and the manner in which the vidoe signal is injected in the picture tube circuit. It should be noted, however, that polarity is not important in an audio system because the phase of ac audio signal for the loudspeaker does not matter in reproduction of sound, but a polarity inversion of video signal driving the picture tube would produce a negative picture. The detector may use either series circuit or shunt circuit, the basic forms of which are shown in Fig. 11.1. The series circuit arrangement is preferred because it is more suited for impedance match between the last IF amplifier output and input of the video amplifier.
Diode Demodulated video output C

Video IF input

C

R

IF input

D

R

Video output

(a)

(b)

Fig. 11.1. Basic detector circuits (a) series (b) shunt.

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VIDEO DETECTOR

211

In most television receivers a positive going signal is obtained at the output of the detector because this is the correct polarity for cathode injection in the picture tube after one stage of video amplification. The choice of polarity is also influenced by the type of AGC used. In some transistor receivers a negative going signal is developed, where a two stage video amplification is employed for feeding the picture signal to the cathode of picture tube. A schematic representation of the two types is illustrated in Fig. 11.2.
+ v0 +v Black level D v0 0 t Last IF stage D v0 –v Negatively modulated video signal Load and filter network Black level

0

t

Negative going signal 0 – v0 Black level Positive going signal t

Fig. 11.2. Production of negative and positive going video signals from a negatively modulated video signal.

11.2 BASIC VIDEO DETECTOR
The basic circuit of a video detector employing a diode is shown in Fig. 11.3 where a parallel combination of C, a small capacitor and RL, a large resistance constitutes the load across which
Cd vs

D Last IF vs – +

0

t

C

RL v0 v0 0 t D.C. level Video signal output

Fig. 11.3. A simple diode detector and filter circuit.

rectified output voltage v0 is developed. Note that the load is connected to anode of the diode to develop a negative output voltage with respect to ground. The diode conducts during negative half cycle of the input to charge ‘C’ up to a potential almost equal to the peak signal voltage vS.

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The difference is due to diode trop, since the forward resistance of the diode though small is not zero. On the downward and positive half of each carrier cycle the diode becomes nonconducting, and during this interval some of the charge on capacitor ‘C’ decays through RL, to be replenished at the next negative peak. The time constant of RLC network is kept large compared with the time period of the applied IF signal. Then the circuit settles down to a condition, in which short current pulses flow through the diode only during the tip of each IF cycle to replenish the charge lost. The time constant of the network must, however, allow the capacitor voltage to follow the comparatively slow variations of the envelope of the modulated carrier. This condition can be shown to occur approximately when
1 = ωm × RL C m 1 − m2

where ωm is the highest modulating frequency and m is the modulation index.With the highest modulating frequency of 5 MHz and assuming an average modulation index of about 0.4, the time constant (RLC) comes to nearly 0.08 µs. The period tC of the carrier (IF = 38.9 MHz) frequency = 0.025 µs and that of the highest modulating frequency (5 MHz) = 0.2 µs. In practice a time constant close to half the period of the highest modulating frequency is chosen for effective and distortion free detection. This, in our case, is 0.1 µs and is nearly equal to the calculated value of 0.08 µs. This time constant (0.1 µs) is seen to be much higher than the IF time period and thus an output voltage that is very nearly equal to that of the envelope of the AM wave is ensured. If the RLC time constant is made too small, the output waveform will have a large IF ripple content which is not desirable. However, if the RLC product is kept too large it will not affect the negative going half-cycle of the envelope waveform, but will cause distortion of the positive going movements of the modulated envelope. This is known as positive peak clipping. Choice of RL and C When the diode is conducting, it is obvious that some part of the output voltage is dropped across Rd, the series diode forward resistance. The detector effeciency then depends on the ratio of RL/Rd, where RL is the load resistance. Thus, for higher efficiency, a diode with a small forward resistance must be chosen and RL should be kept as high as possible. As RL is increased, C has to be reduced to maintain the correct time constant. The smaller the value of C the more significant becomes Cd, the anode to cathode capacitance of the diode. A reference to Fig. 11.3 will show that C and Cd form a potential divider across the input circuit. During positive half cycles of the applied carrier voltage, when the diode does not conduct, the entire signal voltage splits across these two capacitors. This results in some unwanted positive half of the applied voltage being developed across the load capacitor C. This reduces the net negative-going output and hence the efficiency. Therefore in an effort to increase RL, C cannot be reduced too much because then most of the unwanted positive going voltage would develop across RL and reduce the net output voltage greatly.Since the voltage divides itself in inverse proportion to the capacitances, and the larger voltage appears across the smaller of the two capacitors in series, the ratio C/Cd should be as large as possible. If C is to be decreased to use a high value of RL, then Cd must be very small.

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213

Choice of the Diode For the time constant fixed at 0.1 µs, RL varying between 2.7 K and 5 K in parallel with C, between 15 pF and 30 pF are commonly employed as load network for the detector in television receivers. Since RL is only a few kilo ohm, the forward resistance of the diode must be as small as possible. The diode capacitance Cd must also be very small because the value of C chosen for the network is only around 20 pF. Some of the semiconductor diodes that meet the above requirements are 0A79, IN69 and IN64, and are often used in video detector circuits. It would be instructive to compare the values of RL and C chosen for the TV receiver detector with that of a broadcast radio receiver. The corresponding values for a radio receiver detector are : RL = 500 K and C = 100 pF (see Fig. 11.4) and are based on the same considerations as explained for the television detector.

D vs

Filter circuit Rf 50 K

100 pF

C

100 pF

Cf

RL 500 K

v0

IF = 455 KHz fm = 5 KHz

C = 100 pF RL = 500 K

RL C = 50 ms

Fig. 11.4. Detector and filter circuit of a radio receiver.

11.3 IF FILTER
The detector output voltage, v0 consists of three components—(i) the required video signal, (ii) an IF ripple voltage superposed on the video waveform and (iii) a dc component of amplitude almost equal to the average amplitude of the AM wave. The IF carrier component is removed by passing the signal through an IF filter. The dc component, if not required, is blocked by inserting a coupling capacitor in series with the signal path. However, the dc component forms a useful source of AGC voltage and represents the average brightness of the scene. A brief survey of the filter circuit used along with a radio receiver detector will be helpful before looking into the special problems involved in detection of video signals. Such a detector circuit with provision to filter out IF ripple frequency is shown in Fig. 11.4 Rf is chosen to be very much greater than Xcf (reactance of Cf) at the intermediate frequency of 455 KHz. Most of the IF voltage gets dropped across Rf and v0 is practically ripple free. Cf being small acts as an open circuit for the audio signal. Rf and RL then form a potential divider for the desired audio signal, and a part of this signal is also lost across Rf. However, since Rf << RL (nearly 1/10th) the loss of audio signal is very small. The filter configuration shown is standard in radio receivers. The RC filter circuit used in radio receivers is not practicable for video detectors because of very low values of load resistance. A suitable value of Rf to attenuate IF would seriously attenuate the video signal as well, because of small value of RL across which the video output voltage develops. A series inductor is therefore used in place of Rf and the RC filter is thus

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replaced by an LC filter (see Fig. 11.5). At the IF frequency (38.9 MHz), the inductor Lf has much higher reactance than that of the shunt capacitor Cf but at the video frequencies the
D Lf 100 mH Last IF vs C Cf 10 pF RL v0

Fig. 11.5. Basic video detector and fitter circuit.

reactance of Lf is much lower as compared to that of Cf. As labelled in Fig. 11.5, Lf = 100 µH and Cf = 10 pF have been chosen to illustrate the filter action. At IF frequency the ratio of the reactance of series inductor to that of shunt capacitor (XLf /XCf) is nearly equal to 60. This means an attenuation of about 35 db for the ripple voltage and is considered adequate. However, the attenuation of the video signal will be different for different frequency components of the composite signal. The higher video frequencies suffer more attenuation than the lower ones. In order to overcome this discrepancy the load is modified to include an inductor in series with RL. The reactance of this compensating coil (LC) increases with frequency and thus the magnitude of the complex load increases to counteract the additional drop across Lf, the series inductor. A typical value of such a compensating coil is of the order of 200 µH. Such an arrangement also takes care of the input capacitance of the following video amplifier which would otherwise tend to attenuate high frequency components of the video signal. The filter circuit thus modified, ensures low, and almost constant attenuation over the entire frequency range. This is known as video bandwith compensation. The bandwidth must extend up to 6 MHz to include sound intercarrier frequency of 5.5 MHz and its FM sidebands besides the video signal. Figure 11.6, shows a practical video detector circuit. It employs a load resistance = 3.9 K and a compensating coil = 250 µH. Its output is dc coupled to the video driver. The resistors R1 and R2 form a voltage divider across the 12 V dc supply to fix necessary forward bias at the base of the emitter follower (driver). The capacitor C1 provides effective ac bypass across R2.
To driver of video amplifier

From last video IF

D C2

Lf

Lc 250 µH

RL = 3.9 K 100Ω + 12 V C1 R2 15 K R1 = 4.7 K + 12 V

Fig. 11.6. A practical video detector circuit.

VIDEO DETECTOR

215

Filter Circuit Modifications A somewhat non-linear behaviour of the diode results in production of a series of harmonics and beat frequencies at output of the detector. It is possible for these unwanted products to get coupled to the tuner and go through a zero beat as the tuner is tuned (through the corresponding RF range) to produce what are known as ‘tweets’. Such interference is usually confined to channels that lie between 80 to about 180 MHz. To eliminate these interferences, the diode and part of the filter circuitry are enclosed in the ‘can’ of last picture IF stage coupling transformer for effective screening (see Fig. 11.6). In some detector designs self-resonant chokes that are tuned to suppress specific troubling frequencies are inserted in series with the signal path. Figure 11.7 shows such a circuit configuration. Note that a major part of filter capacitors is provided by the stray and wiring capacitances and the ‘wired in’ (physical) capacitors are much less in value and range from 5 pF to 10 pF. Both the series inductors Lfa and Lfb are made to resonate at the desired frequencies by their self-capacitance and no physical capacitors are actually needed. In some cases the filter capacitor Cf across the load is provided by the input capacitance of the video amplifier stage so that no ‘wired in’ Cf appears in the circuit.
Ca D Last IF Lfa vs Cs C1 C2 Lfb Cf Lc RL Cin v0 Cb

Fig. 11.7. A modified video detector circuit.

11.4 DC COMPONENT OF THE VIDEO SIGNAL
The video detector output includes a dc component which must be preserved for a true representation of the transmitted picture. Therefore with dc coupling employed between the detector and the video amplifier and video amplifier to the picture tube, all shades from white to grey get correctly reproduced. However, in some receivers ac coupling is used between the detector and video amplifier. The insertion of a coupling capacitor in series with the signal path completely removes the dc component. The video signal waveform then settles down with equal areas on either side of the zero voltage line. This is illustrated in the waveforms drawn in Fig. 11.8. Note that this results in lesser contrast between two lines having different brightness levels. Similarly an increase in the average brightness of the transmitted scene results at the receiver in a depression of the black level, so that the reproduced range in brightness is less than that of the original scene. Notwithstanding this disadvantage ac coupling is sometimes used because of certain other merits. However, the dc level of the video signal

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can be reinserted by what is known as ‘dc reinsertion’ technique before injecting it at the grid or cathode of the picture tube. This is fully explained in Chapter 14.
0 Peak white level t C 0 Grey line v0 White line Black level Black level v0

Fig. 11.8 (a). Video detector output for two different lines, one grey and the other white. Note the black level is same for both the lines.

Fig. 11.8 (b). Effect of a.c. coupling. The black level is now different and thus the d.c. component is lost.

11.5 INTERCARRIER SOUND
In addition to recovering the composite video signal, rectifying action of the diode in the video detector also results in frequency translation of the sound IF signal. The strong picture IF carrier at 38.9 MHz acts as a local oscillator and heterodynes with the attenuated sound IF carrier at 33.4 MHz to produce a difference frequency of 5.5 MHz. The resulting new IF together with its FM sidebands is known as intercarrier sound signal. The video detector circuit is modified (see Fig. 11.9) to extract the sound signal. As shown in the figure, a parallel tuned circuit, commonly known as sound IF trap is inserted in the signal path. It is tuned somewhat broadly with a centre frequency of 5.5 MHz. This offers a high impedance to the sound component of the detected signal to remove it effectively from the video signal path. A tuned secondary circuit delivers the intercarrier IF to sound section of the receiver. In some TV receivers the intercarrier sound signal is allowed one stage of amplification in the video amplifier and then separated through a trap circuit. It may be noted that in colour receivers two separate diode circuits are used, one as a 5.5 MHz sound convertor and the other for video signal detection. This done to reduce interference in colour pictures due to the beat note produced at the difference frequency of sound carrier and colour subcarrier frequencies.
To sound IF Compensating network

D

Lfa

Lfb

4.7 pF

4.7 pF

RL

3.9 4.7 K K Video amplifier

Sound trap circuit

Fig. 11.9. Video detector circuit with intercarrier sound trap.

VIDEO DETECTOR

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11.6 VIDEO DETECTOR REQUIREMENTS
It is now obvious that in practice the video detector diode load is more complex than a simple shunt combination of resistance and capacitance because of the many functions it is required to perform. The requirements are as follows: (a) The detector load must provide a suitable impedance as seen through the diode, at input of the detector to tune and damp secondary of the last IF coupling circuit correctly. (b) The detector load must remove from the output, the IF content in the signal as much as possible. For this purpose the load usually includes one or two low-pass filter sections. (c) The detector load should have a trap circuit (a series rejector circuit) for separating the intercarrier sound signal. (d) The detector load must also include a provision to boost the higher video frequencies to compensate for the loss due to input capacitance of the video amplifier. It is obvious from these requirements that rigorous theoretical design of such a diode detector is very complex and therefore, in practice the design is usully reached by empirical methods based on filter circuit theory.

11.7 FUNCTIONS OF THE COMPOSITE VIDEO SIGNAL
Figure 11.10 illustrates various paths for the composite video signal as obtained from the video detector. We can consider that the signal is coupled to several parallel branches for different functions. Therefore, each circuit can be operated independently of the others. For instance clipping the sync pulses in the sync-separator stage does not interfere with the video amplifier supplying signal to the picture tube. Similarly, the AGC circuit rectifies the video signal for developing AGC bias. With the same video signal the video amplifier provides complete video signal to the picture tube for reconstruction of the televised scene. In some receiver designs a cathode or emitter follower is used to isolate the video detector from these circuits. In many receivers the video signal for the sync-separation circuit is tapped after one stage of video amplification.
To video amplifier For picture tube

From IF amplifier

Video detector and filter circuit

To colour amplifiers (in colour receivers) To AGC circuit

To sync separator

Fig. 11.10. Functions of the composite video signal.

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Review Questions
1. 2. 3. 4. 5. Draw the basic circuit of a video detector and explain design criteria for the choice of time constant of the load circuit. What determines the polarity of the diode in the detector circuit ? Describe the factors that influence the choice of RL and C in the load circuit. Suggest suitable values of RL and C for the 525 line system where IF = 45.75 MHz and fm = 4 MHz. Explain why is it necessary to employ and L-C instead of R-C filter to remove IF ripple from the detected output. Give typical values of L and C and justify them. Why is the filter circuit generally modified to include self-resonant inductors in the signal path ? What precautions are taken to prevent undesired harmonics from reaching the tuner ? Explain how compensation is provided in the detector load circuit to extend its bandwidth. Why is it necessary to have a bandwidth of nearly 6 MHz ? Sketch the complete circuit and label all components. How is the FM sound signal at 33.4 MHz translated to a new carrier frequency of 5.5 MHz and separated from the composite video signal ? Give a practical video detector circuit imcorporating the following features: (i) effective IF filtering, (ii) suppression of harmonics, (iii) separation of the intercarrier sound signal, and (iv) frequency compensation. Give typical values of all the circuit components.

6. 7.

12
Video Section Fundamentals

12
Video Section Fundamentals
The amplitude of composite video signal at the output of video detector is not large enough to drive the picture tube directly. Hence, further amplification is necessary, and this is provided by the video amplifier. The manner in which video signal is applied to the picture tube (cathode or control grid) decides the type of video section circuitry. The video signal on application to the picture tube varies the intensity of its beam as it is swept across the screen by deflection circuits. The gain control of the video amplifier constitutes the contrast control, whereas the brightness control forms part of the picture tube circuit.

12.1 PICTURE REPRODUCTION
Figure 12.1 shows how the input video signal voltage for one line, impressed between grid and cathode of the picture tube results in reproduction of picture elements for that line. It should be noted that the results are the same when reversed video signal is applied between cathode and control grid. In fact the video signal should so align itself that its black level drives the grid voltage to cut-off. Any grid voltage more negative than that is called blacker than black and this part of the video signal corresponds to sync-voltage amplitude. At the grid of picture tube sync pulses really have no function, but these are used in the synchronizing section of the receiver to time deflection circuits for vertical and horizontal scanning.

12.2 VIDEO AMPLIFIER REQUIREMENTS
In order to produce a suitable image on the screen of picture tube, the video amplifier must meet the following requirements. (i) Gain The video signal must be strog enough to vary the intensity of the picture tube scanning beam to produce a full range of bright and dark values on the screen. This is illustrated in Fig. 12.1, where the signal amplitude is large enough to provide the desired contrast between white and dark parts of the scene being televised. However, with a video signal having smaller peak-topeak variations, the brightness extends from dark, at cutoff bias, to some shade of grey, with the result that there is less contrast between dark and light areas. Figure 12.2 illustrates the light variations produced when the signal amplitude is reduced to about half as compared to the signal amplitude required for full contrast. Any further reduction in the video signal amplitude will result in a washed-out picture, and there will be little difference between dark and light areas of the picture. A video signal amplitude of about 75 volts peak-to-peak is needed 220

to obtain a picture with full contrast. Some colour picture tubes need higher signal amplitudes of the order of 150 V peak-to-peak, for proper reproduction of the picture. With a detector output betwen 2 to 4 volts in all tube receivers, a gain between 25 to 50, at the video amplifier is considered adequate. A single pentode valve can develop this gain and most TV receivers

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using tubes have one stage of video amplification between video detector and picture tube. However, in transistor receivers where detector output seldom exceeds 2 volts, a two to four stage video amplifier becomes necessary to fully modulate the beam of the picture tube. (ii) Bandwidth As explained in an earlier chapter higher frequencies are needed to reproduce horizontal information of the picture. The lowest frequency for picture information in the horizontal direction can be considered as 10 KHz when the camera beam scans all white and all black lines alternately. Note that the active line period has been taken as 50 µs instead of the actual period of 52 µs. Similarly when the beam scans half white and half black lines in succession, the video frequency generated is 20 KMz. This is illustrated in Fig. 12.3 where different alternate black and white widths have been closen to demonstrate the generation of high frequency signal. Thus, for reproducing very minute details, a very high video frequency would be necessary, but keeping in view the limitations of channel bandwidth, the upper limit has been fixed at 5 MHz.
Width of picture or » 50 ms 100 KHz = 5 ms » 200 KHz = 2.5 ms » ½ cycle of 500 KHz = 1 ms » 5 MHz = 0.1 ms » (a) Horizontal information (b) Vertical information Height of the picture » 18720 ms = ½ cycle of 26.7Hz » 25 Hz » 50 ms

Fig. 12.3. Relationship between picture size and video frequencies.

The signal frequencies corresponding to picture information scanned in the vertical direction are much lower compared with those for reproduction within a line. If the video voltage it taken from top to bottom through all the horizontal lines in a field, the variation will correspond to a half-cycle of a signal with a frequency of approximately 25 Hz. When the brightness of the picture varies from frame to frame the resultant signal frequency is lower than 25 Hz. However, this is considered as a change in dc level corresponding to a change in the brightness of the scene; and this can become almost zero Hz (i.e., dc) when the average brightness does not change over a long period of time. Ideally then, the video amplifier response should be linear from dc to the highest modulation frequency of 5 MHz. This is possible only when the video amplifier is direct coupled. (iii) Fequency Distortion The gain at high frequencies falls-off because of shunting effect of the device’s output capacitance, stray capacitances and input capacitance of the picture tube. When ac coupling is employed the gain decreases at low frequencies on account of increasing reactance of the coupling capacitor. This inequality in gain at different frequency components of the signal is called frequency distortion. Excessive frequency distortion cannot be tolerated because it changes picture information. If high frequency content of the video signal is lost due to poor high

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frequency response the rapid changes between black and white for small adjacent picture elements in the horizontal line cannot be reproduced. This results in loss of horizontal detail. For example in the test pattern shown in Fig. 12.4, the individual black lines close to the centre will loose their identity and instead appear as an illdefined black patch. Similarly small details such as individual hairs of a person’s eyebrows do not appear clearly. However, in a close up of the same face, where each area of the picture gets enlarged, the sharpness of each detail improves because a relatively low frequency range is required for proper reproduction of details. Frequencies from about 100 KHz down to 25 Hz represent the main parts of the picture information, out of which frequencies from 100 KHz to about 10 KHz correspond to black-andwhite information of most details in the horizontal direction and frequencies from 10 KMz down to 25 Hz represent changes of shading in the vertical direction. If the low frequency response is poor, the picture as a whole is weak with poor contrast. The lettering, if any, is not solid and the average brightness appears to be changing gradually from top to bottom of the raster, instead of complete change of brightness in the actual scene.

(iv) Phase Distortion Phase distortion is not important in audio amplifiers, because the ear does not detect changes in relative phases of the various frequency components present in a given sound signal. However, it is important in video amplifiers, since phase shift implies time shift, which in turn means position shift in the reproduced visual image. The resultant shift in relative positions of the various picture elements is detected by the eye as distortion. Therefore relative phases of all the frequency components present in the video signal must be preserved. The time delay due to phase shift is not harmfull if all frequency components have the same amount of delay. The only effect of such uniform delay would be to shift the entire signal to a later time. No distortion results because all components would be in their proper place in the video signal waveshape and so also in the reproduced picture. Therefore, the phase angle delay should be directly proportional to the frequency or all frequency components must have the same time delay. This is illustrated in Fig. 12.5 (a) and (b). It should be noted that the signal inversion of exactly 180° in any one stage of the video amplifier does not mean phase distortion. There is no time delay, but only a polarity reversal.

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Phase angle delay q° Time delay l(ms)

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q

l

Frequency (a)

Frequency (b)

Fig. 12.5. Phase response of the amplifier: (a) Phase angles proportional to frequency. (b) Corresponding time delay which is constant.

Figure 12.6 shows frequency response of an RC coupled amplifier where fL and fH are the lower and upper 3 db down frequencies. The corresponding phase shift angles at these frequencies are +45° and –45° relative to midband frequencies. Phase distortion is very important

at low video frequencies because here even a small phase delay is equivalent to a relatively large time delay. As an illustration consider an amplifier designed to have fL = 2.5 Hz. The phase shift at 2.5 Hz = 45° and that at 10 fL (25 Hz) it is nearly 6°. The relative time delay
6 10 6 × ≈ 660 µs, which in turn 360 25 would mean that the picture information due to the two frequency components (2.5 Hz and

between this frequency and midband frequencies would be

660 ~ − 10 lines. To correct this 64 discrepancy even if phase shift at 25 Hz is made = 1°, the corresponding time delay would be 120 µs and the picture information will get displaced by about 1.5 lines on the raster. The eyes are very sensitive to time delay errors and see this as ‘smear’ on the picture. At very high video frequencies the effects of phase distortion are not as evident on the screen because the time delay at these frequencies is relatively small. For example, if fH, the upper corner frequency of the amplifier is set at 5 MHz, the corresponding time delay with respect to midband frequencies
25 Hz) would get displaced with respect to each other by nearly is only

too small to be detected. Thus, a video amplifier with flat frequency response up to the highest useful frequency, has negligible time delay distortion for very high video frequencies.

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225

(v) Amplitude Distortion of Nonlinear Distortion If the operating point on the transfer characteristics of a device for a given load and signal amplitude is not carefully chosen, amplitude distortion occurs where different amplitudes of the signal receive different amplification. This can result in limiting and clipping of the signal or in weak signal output. If sync pulse voltage gets compressed, synchronization may be lost, because the video amplifier usually provides composite video signal for the sync separator. Very often some gain has to be sacrificed to avoid amplitude distortion. (vi) Manual Contrast Control It should be possible to vary amplitude of the video signal for optimum setting of contrast between white and black parts of the picture. Any control that varies the amount of ac video signal will operate as a contrast adjustment. Therefore contrast varies when gain is varied in either picture IF section or video amplifier. However, such a control is not possible in the IF section because all receivers employ automatic gain control circuits to maintain almost a constant voltage at the output of the video detector. Also, with intercarrier sound, any change of gain in the IF section would affect the sound volume. Therefore, contrast control is provided in the video amplifier, and this in effect, is the gain control of video amplifier.

12.3 VIDEO AMPLIFIERS
It is obvious from the preceding discussion that video amplifiers must meet several exacting demands and this calls for careful and rigorous design considerations. The wide-band requirement starting from almost dc to several MHz with minimum phase distortion is perhaps the most stringent requirement. Both direct-coupled and RC coupled configurations are used and each type has its own merits and demerits. Both types need high frequency compensation, and this is met by shunt-peaking and series-peaking techniques. Though a dc amplifier does not need any low frequency compensation, the RC coupled amplifier employs special low frequency boost techniques to extend the bandwidth at the lower end of its response. A basic RC coupled amplifier configuration, which applies to both tubes and transistors, is shown in Fig. 12.7. It is designed to work under class ‘A’ operation. The gain of this amplifier
Cc v0 0.1 mF vin Video amplifier Ct 18PF RL = 2.5 K B+ B– Rg = 250 K

falls off rapidly at high frequencies because of shunting effect of inter-electrode, input, and stray capacitances in parallel with the load resistance RL. Video stages use low value of RL compared with audio amplifiers because of large bandwidth requirements. Typical values are 2 KΩ to 8 KΩ in tube versions. Although gain is reduced, lowering RL extends the high frequency response. The effect of Ct does not become important until its reactance is low enough to become comparable with the resistance RL. Then, the shunt reactance Xct lowers the impedance ZL, and this causes a fall in the gain at higher frequencies. The lower the RL, and smaller the shunt capacitance Ct, better is the high frequency response. However, practical video amplifiers generally use a relatively higher value of RL with peaking coils to boost the gain at higher frequencies. A typical arrangement known as ‘shunt peaking’ is shown in Fig. 12.8. Here a small inductor, L0 is connected in series with RL. This peaking coil resonates with Ct to boost the gain at high frequencies (see Fig. 12.8 (b)), where the response of the uncompensated RC coupled amplifier would normally drop off. The resistance of the coil is very small and so it does not effect dc voltages and response at the middle frequencies. A peaking coil is effective for frequencies above about 400 KHz.
Cc v0 Q vfn Bias + – RE + – (a) RL Vcc 0 RB 0 v0 Without shunt peaking (b) Ct L0 0 v¢0 f The response curve of Ct and L0 in parallel f With shunt peaking f vin

Another arrangement to extend the high frequency response is known as ‘series peaking’ compensation (Fig. 12.9). In this circuit, LC is in series with the two main components of Ct. At
RD Cc LC Q vfn + – RE Cout RL + 0 (a) Video amplifier with series peaking Cin v0 To picture tube input circuit Rin Without series peaking (b) Frequency response vin

one side of LC is Cout of the video amplifier and on the other side is Cin of the next stage or input capacitance of the picture tube. This arrangement reduces shunting capacitance across RL which results in more gain, while LC resonates with Cin to provide a rise in voltage across Cin at high frequencies. A series peaking coil usually has a shunt damping resistance such as RD, the function of which is to prevent oscillations or ringing in the coil with abrupt changes in signal. The circuit of Fig. 12.10 combines shunt and series peaking which results in more gain, extended high frequency response and improved transient behaviour.
Lc = 145 mH Cc v0 0.1 mF 5 PF vfn Video amplifier C0 L0 RD = 10 K 35 mH RL 4K Contrast control R1 C2 .005 mF Cf 10 mF Rf 10 K B+ B– Cln 12 PF Rin 400 K To picture tube circuit

Fig. 12.10. Video amplifier employing both shunt and series peaking for high frequency compensation and a special decoupling circuit to boost low frequency response. Note that the component values shown in the circuit are for a tube amplifier.

The low frequency response is affected by increased reactance of the coupling and bypass capacitors. This can be improved by using largest possible values of these capacitors. In adition, the decoupling filter RfCf in the B+ supply line (Fig. 12.10) can be used to boost gain and reduce phase shift distortion at very low frequencies. The capacitor Cf offers large reactance at low frequencies and then Rf in series with RL becomes the effective load. The increased load results in higher gain for low frequencies. This rise in gain compensates for the reduction in gain caused by reactance of the coupling capacitor CC. Furthermore, phase shift caused by shunt capacitor Cf is opposite to phase shift caused by series capacitor CC. As a result Cf tends to correct phase distortion introduced by the coupling cpacitor. A small paper or ceramic capacitor C2 is normally provided across Cf to bypass very high video frequencies because Cf, being a large value electrolytic capacitor, has a non-negligible inductance and fails to provide a bypass at these frequencies. Gain Control The change in gain of the video amplifier to provide contrast control can be effected in different ways. A common method is to vary the cathode/emitter resistance R1. This resistance is left unbypassed and its variation alters the negative feedback which in turn changes the gain of the amplifier. The change in the value of R1 also varies the bias and the consequent shift in the operating point can introduce amplitude distortion. To overcome this problem, in many video amplifier designs, a potentiometer is provided at the output of the video amplifier to alters the

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magnitude of the video signal applied to the picture tube. This method is the same as volume control in an audio amplifier. 5.5 MHz Sound Trap The video amplifiers usually have a trap circuit, tuned to the intercarrier sound frequency of 5.5 MHz to keep the sound signal out of the picture signal. If sound signal is not separated at the video detector, the trap circuit can be modified to deliver the intercarrier sound signal to the sound IF amplifier. This is illustrated in Fig. 12.11 (a) where L1 and C1 constitute the trap circuit and winding L2 coupled to L1 serves as take-off point for the 5.5 MHz sound signal.
To sound IF amp L1 C1 5.5 MHz trap cct vin Video amplifier L2 Lc v0 Cc RL RD RD Video amplifier RL L0 B+ C 5.5 MHz trap cct Lc Cc v0

L0 B+ – (a)

vin

L

(b)

Fig. 12.11. Video amplifier circuits (a) with parallel resonant trap in series with the output. The trap circuit also serves as the sound take-off circuit (b) with series resonant 5.5 MHz trap in shunt with the load.

As shown in the figure, the trap circuit is in series with the output and when tuned to resonance at 5.5 MHz, offers maximum impedance. Therefore, the sound signal is removed because maximum voltage across the trap is developed at this frequency. In Fig. 12.11 (b) another trap circuit arrangement is shown. Here L and C form a series resonant circuit tuned to 5.5 MHz. This trap circuit is in shunt with the load, and at resonance, provides practically a short circuit to the intercarrier sound signal. This prevents it from appearing across the picture tube input. If the 5.5 MHz sound signal together with its sidebands is not fully suppressed it causes beat interference which results in diagonal lines on the picture having small wiggles. The weave in the lines is the result of frequency variations in the FM sound signal. This effect is also called ‘wormy’ picture. The interference disappears when there is no voice or music, leaving just the straight lines corresponding to the 5.5 MHz carrier without modulation. The trap circuits are tuned for minimum interference in the picture.

12.4 BASIC VIDEO AMPLIFIER OPERATION
Before attempting to discuss complete video section circuits it is desirable to recapitulate the operation of a basic RC coupled amplifier, which after adding compensating elements serves as the video amplifier. The circuit of such an amplifier employing a pentode is drawn in

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Fig. 12.12 (a). The voltage drop across RK provides grid bias and ensures class ‘A’ operation. The load resistance RL provides output voltage between the plate and ground. While Rs is the screen voltage dropping resistance. The bypass capacitor CS, connected between screen grid and ground provides an effective short at all frequencies of interest. CK connected across RK is a bypass for the ac component of the current to prevent degeneration of the input signal. The input signal waveform and corresponding plate current and plate voltage waveforms are shown in Figs. 12.12 (b) to (d). When the input signal swings to the extreme positive, maximum current flows and the plate voltage swings to its minimum value. Again, when the input signal attains its maximum negative value, the plate current becomes minimum and plate voltage attains its maximum positive value. Thus, the output voltage is 180° out of phase with respect to the input voltage. The dc component of the plate voltage is blocked by coupling capacitor (CC) and the amplified ac component then appears across the output terminals of the amplifier. An RC coupled amplifier employing a transistor in the common emitter configuration performs in the same way as its tube counterpart, but with a difference, that instead of voltage it needs current drive for its operation. Normally potentiometer biasing is provided for setting the operating point. The voltage and current polarities for an n-p-n transistor are the same as with a vacuum tube. However, with a p-n-p transistor all signal polarities are negative with respect to ground.
Cc P Cc vin Rg G1 G2 RS 25 K K RK CS 4 mF 250 v CK RL 5K + Vpp – VGK P–P 4V (vin) v0 0 t

(a) Amplifier circuit volts iP mA 40 P–P value = 31 mA 185 150 225

(b) Input voltage

VP0

20 IP0 (average value) 0 (c) Plate current t

75

P–P value = 140 V t (d) Plate voltage

0

Fig. 12.12. R.C. coupled amplifier.

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12.5 COMPARISON OF VIDEO SIGNAL POLARITIES IN TUBE AND TRANSISTOR CIRCUITS
As explained earlier a negative going signal is a must when the video signal is injected at the cathode. Similarly positive going signal is required for feeding at the grid of the picture tube. If opposite polarity is used the result will be a negative picture, in the same sense as a photographic negative. Besides this, the polarity of video signal with respect to ground also affects the biasing of the device employed in the video amplifier. These aspects are examined in greater detail both for tube and transistor amplifiers using grid and cathode modulation of the picture tube. Grid Modulation of the Picture Tube (i) Transistor Circuits. The circuit arrangement shown in Fig. 12.13 is of a direct coupled video amplifier employing a n-p-n transistor to grid modulate the picture tube. The resistors R1 and R2 together with RE, that is bypassed by CE, provide forward biasing at the base-emitter junction. This biasing arrangement provides good stability of the operating point against device replacement, temperature variations, dc supply changes and ageing of circuit components.
B+ Brightness control

Since a positive going signal is needed at the collector, a negative going video signal is required to the base. The value of emitter resistance RE is chosen to set the operating point ‘Q’ for minimum collector current in the absence of input signal. In fact the base-emitter bias is so set that when video signal is applied, the sync tips make the base most positive. Thus the collector current becomes maximum for sync tip levels and the collector acquires minimum positive potential. This is the correct polarity for obtaining minimum beam current of the picture tube. Since the operating point of the transistor is set close to cut-off bias, the collector voltage, in the absence of any video signal is highest, which makes the grid of the picture tube least negative with respect to cathode. Therefore, with no video voltage drive, though the transistor draws a minimum current, the picture tube beam current is maximum.

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Notice that the cathode of picture tube is returned to the brightness control which sets the cathode at a voltage which is more positive with respect to ground than the grid. Brightness control is achieved by varying net negative voltage between grid and cathode. The resistance RX fixes the minimum limit of bias and prevents the possibility of a net positive voltage on the grid. If a p-n-p transistor is employed the base-emitter junction must be enough forward biased, with no input signal, so that the positive going sync tips drive the bias backwards towards the Ib = 0 point. Note that the input signal is all positive and the least positive level is reached on peak-white and the transistor then passes maximum current. The maximum collector current results in least negative collector voltage which in turn makes the picture tube grid less negative with respect to its cathode and the beam current increases to reproduce the peak-white values of the picture. It may also be noted that in the absence of any video signal the transistor collector is at a minimum negative potential and the corresponding beam current is large. Similarly a p-n-p transistor needs a dc source of opposite polarity than a n-p-n configuration. This necessitates interchanging of the locations of orightness control and resistor RX to make sure that the picture tube grid never attains a positive potential. (ii) Vacuum Tube Circuit. In the case of a vacuum tube a positive supply voltage is needed and a positive drive to the grid is necessary to increase the plate current. Therefore, the working conditions of a vacuum tube video amplifier are exactly similar to those of a n-p-n transistor. However, the potentials needed are much higher and the biasing technique is somewhat different. Thus in a vacuum tube amplifier, with no input signal the tube draws a minimum current, and the picture tube beam current is maximum. Cathode Modulation of the Picture Tube When the video signal is injected at the cathode of picture tube a negative going signal is needed at the anode/collector of the video amplifier and this necessitates a positive going signal at the grid/base of the amplifier. (i) Tube Circuit. The necessary circuit details of the video section and the signal waveforms at the input and output of the video amplifier are illustrated in Fig. 12.14. The tube is biased close to VGK = 0 point so that maximum current corresponds to peak-white and minimum plate current occurs on sync pulse tips. The anode voltage waveform is then negative going as required for correct cathode modulation. The picture tube beam current is again high with no signal at input of the amplifier. (ii) Transistor Circuits. Fig. 12.15 shows a p-n-p configuration and associated waveforms. The transistor input circuit is back biased towards Ib = 0 µA so that on peak-whites the base current and hence the collector current is very small. Minimum collector current results in maximum negative voltage at the collector, so that once again the necessary negative going collector signal is derived for the picture tube cathode. In the absence of any input video signal the collector voltage is more negative and hence the signal beam current is maximum. If a n-p-n transistor is employed in the video amplifier designed for cathode modulation, the signal polarities both at the input and output terminals are exactly the same as shown in Fig. 12.14 for a vacuum tube configuration. Therefore the picture tube beam current will be high with no input signal to the amplifier.

These results may be summarized as follows: (i) The sense of the video signal relative to black level seen either at the input or output terminals is the same in tube and transistor circuits. (ii) When the tube or n-p-n transistor is called upon to deliver maximum plate/collector current the p-n-p transistor has to pass minimum collector current and vice versa. (iii) The picture tube beam current is maximum with no input video signal both for grid modulation and cathode modulation when dc coupling is used.

0

t

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233

12.6 RELATIVE MERITS OF GRID AND CATHODE MODULATION OF THE PICTURE TUBE
The main points to be considered in assessing relative merits of the two systems are as follows: (i) Picture Tube Input Characteristics This is a measure of the beam current change for a given change in the video signal drive voltage. The advantage lies with cathode modulation. The beam current is determined by the grid-cathode voltage and by the positive voltage on the first anode with respect to cathode. With grid modulation, the only factor which determines the change in beam current results from a given change in video signal voltage between the grid and the cathode. No other electrode voltage is effected with the applied voltage. However, when cathode modulation is applied a second factor influences the beam current. This is the voltage between the first anode and cathode. Thus as the video signal input voltage moves, from black-level towards peak-white the cathode moves more negative, not only to the control grid but also to the first anode. As stated above the voltage on the first anode and cathode itself have a marked influence upon the beam current of the picture tube. Thus the change in beam current brought about by effective reduction in grid-to-cathode. negative bias, is further augmented due to increase in positive voltage between the first anode and cathode. However, with grid modulation the voltage between the first anode and cathode remains constant. If follows then, that for a given change in the input signal voltage, the change in beam current is less with grid modulation than with cathode modulation. For the same input, the beam current is about 20% greater on peak-white with cathode modulation, or the video amplifier gain can be smaller for the same beam current. (ii) Feed to the Sync Separator Most commonly used sync separator circuits employ either a tube or a transistor, which is cutoff during the picture information part of the video signal but is driven into conduction by the sync pulses. A negative going video signal is then needed at the grid of the tube or base of the n-p-n transistor being used as a sync separator. With cathode modulation the right polarity is available at the anode/collector of the tube/transistor. It is not so when grid modulation is employed and thus cathode modulation has the added advantage over grid modulation of automatically providing the right polarity of the video signal needed to drive the sync separator. However, if a p-n-p transistor is used for sync separation, the grid modulation will provide signal with the correct polarity. With transistor receivers employing a two stage video amplifier, the question of feeding the sync separator is no longer important since both positive and negative polarities of the video signal are always available. (iii) Safety of Picture Tube in the Event of Video Amplifier Failure With tube receivers or the ones employing an n-p-n transistor for video signal amplification the advantage lies with cathode modulation if direct coupling is used. Should the emission of the video amplifier tube fail or the n-p-n transistor stop conducting, the plate/collector current drops to zero and the cathode of the picture tube attains a positive potential equal to B+ supply. This immediately cuts-off the beam current and thus no damage is caused to the picture tube. With grid modulation such a fault will make the grid highly positive causing excessive beam current. The shorts between plate and cathode (collector and emitter) are rare and need not be considered. When a p-n-p transistor is employed in the video amplifier the effects would be opposite.

234 (iv) Cathode to Heater Voltage Stress

MONOCHROME AND COLOUR TELEVISION

In tube receivers using series heater arrangement the picture tube is placed at or near the ground end of the chain to give it. maximum protection against possibility of damage due to short circuits across the heater line. Thus the heater is at negative end of the dc supply. With grid modulation the cathode voltage of the picture tube is at a higher potential than when cathode modulation is used. The stress is therefore more in the case of grid modulation. However, in modern picture tubes the likelihood of cathode heater breakdown has been minimized. Also with capacitive coupling the voltages get reduced, and therefore this point does not very much influence the choice of modulation method. In conclusion, it may be said, that the video amplifier is one of the most important section of the receiver. It not only amplifies the video signal which extends from almost dc to a very high video frequency of 5 MHz, it also acts as the source for feeding the video signal to the sync separator and automatic gain control circuits. In most cases the sound signal is also separated at this stage after amplification. It would then be desirable to take up a detailed consideration of the video section design and circuitry. The next two chapters are devoted to these aspects of video amplifiers.

Review Questions
1. 2. 3. What are the essential requirements that a video amplifier must meet for faithful reproduction of picture details ? How does phase distortion in the video signal affect the quality of the picture ? What causes ‘smear’ in the picture and how can this be minimized ? Draw the circuit diagram of an RC coupled amplifier employing an n-p-n transistor in common emitter configuration and explain its operation as a voltage amplifier. Sketch its frequency response and explain why the gain falls-off both at very high and low frequencies. Describe briefly with circuit diagrams the techniques employed to extend the bandwidth of an RC coupled amplifier to accommodate full range of the video signal. Why are trap circuits provided in video amplifiers to attenuate frequency spectrum occupied by the FM sound signal ? What is the undesired effect of sound signal on the reproduced picture ? Draw simple circuit diagram of a dc coupled video amplifier that feeds the grid of the picture tube. Sketch suitable input and output voltage waveforms and justify that the chosen polarity of the video signal will result in correct reproduction of the picture on the screen. Identify the location of the brightness control in the circuit drawn by you. Discuss relative merits of cathode and grid modulation of the picture tube. Explain why cathode modulation is considered superior to grid modulation.

4. 5. 6.

7.

13
Video AmplifiersDesign Principles

13
Video AmplifiersDesign Principles
The choice of basic amplifier that can be modified to meet the requirements of a video amplifier is restricted to direct coupled and RC coupled configurations. The techniques employed to achieve broad-band characteristics are same for tube and transistor amplifiers. However, their mode of operation and impedance levels are quite different from each other. Therefore, while explaining design fundamentals, video amplifiers employing tubes and transistors are considered separately.

13.1 VACUUM TUBE AMPLIFIER
The basic circuit of a video amplifier employing RC coupling, together with its ac equivalent circuits valid for medium, high and low frequency regions are shown in Fig. 13.1. The capacitor C0 represents output capacitance of the tube, Cs stray shunt and wiring capacitance and Ci input capacitance of the picture tube circuitry. All the three capacitances are effectively in parallel, and when added constitute Ct = C0 + Cs + Ci. The total shunt cpacitance seldom exceeds 20 pF and thus acts as an open circuit to frequencies in the low and midband ranges. The coupling capacitor CC is chosen to be quite large to provide nearly a complete ac short, even at very low frequencies. Gain Expressions It is an easy matter to deduce gain expressions for the three frequency regions from the corresponding equivalent circuits of the basic amplifier configuration (Fig. 13.1). The results are summarized below as a starting point for explaining wide-banding techniques. (i) Gain at midband (Amid) (see Fig. 13.1 (b)) A(mid) = – gm R || ≈ – gm RL ...(13.1) where R || = RL || Rg || rp ≈ RL, because load resistance in video amplifiers seldom exceeds 10 K-ohms. The minus sign signifies phase reversal of 180°. (ii) Gain at high frequencies (see Fig. 13.1 (c)) A(HF) = – gmZt where Zt = RL || X C . t Substituting for X C = t

1 , Zt = 0.707 of its midband value and thus the gain at this 2πCt RL frequency falls to become – 3 db with respect to the midband gain. (iii) Gain at Low Frequencies (A(LF)) (see Fig. 13.1 (d)). Proceeding in the same way as for the high frequency gain expression, and after a little manipulation, the gain in the low frequency region can be expressed as
Note that at f = fH =

frequency again falls to –3 db with respect to midband gain. The frequencies fH and fL are known as corner frequencies and the gain at these frequencies is 70.7% of the midband value. Bandwidth The bandwidth of an amplifier is defined as BW = (fH – fL) ≈ fH =

1 2πCt RL

...(13.4)

In an RC coupled amplifier, even when RL is made as low as 0.5 K-ohms, fH seldom exceeds 3 MHz. However, it is not possible to make the load resistance (RL), too small, because the gain requirement (gain = gmRL) of the video amplifier is not fully met. Therefore, some other means have to be devised to extend the frequency range upto 5 MHz without unduly reducing RL.

13.2 HIGH FREQUENCY COMPENSATION
The bandwidth is normally extended by making the plate load complex in such a way that its magnitude increases with increase in frequency. Thus, the compensation technique is aimed at pushing up the upper –3 db frequency fH, which normally would occur at a relatively low frequency due to the presence of shunt capacitance Ct. Negative feedback is also applied to increase the bandwidth but this results in some loss of gain. The various HF compensation techniques are as follows: (a) Shunt Inductance Peaking A small inductor of the order of 50 to 250 µH is added in series with the load resistor RL. Though connected in series with RL, the coil is in fact a part of the shunt plate circuit. This is llustrated in Fig. 13.2, where the compensated amplifier configuration together with its high frequency equivalent circuit is drawn. As shown there, the effective circuit, in shunt with the signal path, consists of Ct in parallel with series combination of RL and inductor Lpx. The inductor increases the net plate circuit impedance at high frequency end of the frequency

As stated earlier the gain at fH where XCt = RL, falls to 70.7% of the midband value. Therefore, to extend the midband range, Zt should increase at this frequency to yield a gain equal to the midband gain. This can be readily achieved by setting X L px = fH) in equation (13.5). This, on substitution, yields

Thus, the midband gain extends to fH, where, without compensation it was down by

Procedure for fixing RL and Lpx. It is necessary to first determine the total shunting capacitance (Ct) and the highest frequency (f) up to which flat response is desired before fixing the values of RL and Lpx. The highest frequency of interest in the 625 line system is 5 MHz. The value of Ct can be estimated from the data of the tube chosen for the amplifier, and by measuring stray capacitances if necessary. With both f and Ct known, the values of RL and Lpx can be found as under. RL =
1 2πfCt

This equation can be written in a general form as where ‘n’ can be made to have any value of < 1, in order to vary frequency response in the region close to the new value of fH. It may be noted that the circuit will resonate if ‘n’ exceets unity. Effect of varying Lpx. If gain versus frequency plot of such an amplifier is drawn for different values of ‘n’, it is revealing to note, that for values of ‘n’ greater than 0.5, the response has a peak which becomes more pronounced as ‘n’ increases. Furthermore, increasing the value of inductance (i.e., n) increases the amplitude of the hump and also steepens the rate at which the gain falls off above fH. It is characteristic of compensating coils that while they lift the response curve in the desired region, the subsequent fall-off of gain is more rapid than in the uncompensated circuits. Too steep an edge in the response curve is not a desirable characteristic, since it can give rise to a tendency to produce overshoot or transients. In fact no single value of ‘n’ can give (i) constant gain throughout the pass-band, (ii) linear phase response, and (iii) fast transient response without overshoot, all at the same time. It can be shown, that for optimum frequency response, a value of n = 0.414, for least phase distortion; n = 0.322 and for critical damping n = 0.25 is necessary. In the practical development of a particular circuit, it is usual to start-off with an inductor of value Lpx = 0.5 RL2Ct and then experiment with larger or smaller inductors until the desired response is obtained. It may be noted that time delay due to phase shift at high frequencies is very small and if linear phase characteristics are not obtainable while satisfying other requirements, it will not cause any problems. (b) Series Inductance Peaking In this arrangement the compensating coil is inserted in series with CC, which means that the inductor is in series with the signal path, rather than in shunt with it. Figure 13.3 shows this circuit arrangement with its equivalent circuit. In practice, the coil is fixed very close to the plate pin of the tube, and in this position it effectively separates the total shunt capacitance Ct, into two parts, with C0 on the tube side and (Cs + Ci) on the other side of the coil. As seen in the equivalent circuit, this arrangement takes the form of a low-pass filter. The value of Lpy is so chosen, that the filter passes all frequencies within the required video band, but offers a rising attenuation above the upper limit of this frequency hand.

Since the total shunt capacitance gets divided into two parts, it is possible to choose a higher value of RL, because C0 is only across RL and not Ct as was the case in shunt compensation. A 50% increase in RL becomes possible, i.e., Eqn. (13.7) can be modified to become RL =
1.5 2πf H Ct

. ..(13.10)

This results in higher gain of the amplifier. Choice of Lpy. It is obvious that the behaviour of the filter will be affected by the disposition of the total shunt capacitance across the input (shown as C1) and across the output (shown as C2) of the resultant filter configuration. A typical value of C2/C1 is 0.75. A useful basic design formula for the inductor is given by Lpy = nRL2Ct ...(13.11) where n varies between 0.5 and 1. With a ratio of C2/C1 = p = 0.75, a value of n = 0.67 is commonly used, since it gives optimum frequency response. Though with series compensation more gain is possible and a better rise time performance results, but the fall-off in gain just beyond and upper edge of the required band is much steeper. This can cause excessive overshoot and even oscillations. This tendency towards ringing can be reduced by connecting a resistance in parallel with Lpy. A typical practical value of such a damping resistance is 5RL and varies between 15 and 20 K-ohms. (c) Combined Shunt and Series Peaking Coils Shunt and series inductance compensation can be combined to get a peformance slightly superior to that of the series peaking circuit. The corresponding amplifier configuration with its equivalent circuit is shown in Fig. 13.4. The following formulae may be used as a guide to establish approximate values of Lpx and Lpy Lpx = nxRL2Ct and Lpy = nyRL2Ct ...(13.12) where nx and ny are the corresponding values of n and are dictated by the value of p, i.e., (C2/C1). As a starting point, approximate values can be determined with the help of the following chart:

It may be noted that in any case it would be necessary to test experimentally with various inductors to achieve the desired response. This is best done by using a visual display system*. In addition, the transient response may be checked by feeding a square-wave signal to the amplifier, and measuring the rise-time of the output waveform with a cathode-ray oscilloscope. Cathode Compensation. The basic principle of this method is to apply negative feedback over the low and middle frequency regions, but arrange to remove it progressively in the HF region. Because of negative feedback the overall gain gets reduced but it results in a considerable increase in bandwidth. The simplest of the various possible circuit arrangements is shown in Fig. 13.5 (a), where the value of Ck has been so chosen, that it completely bypasses Rk at high video frequencies, but at medium and low frequencies, its reactance becomes comparable with Rk. This results in negative feedback which increases as the frequency decreases. In fact at medium and low frequencies, the reactance of Ct is large compared with RL, and that of Ck is large compared with Rk, thus the amplifier effectively performs as one with a plate load of RL and an unbypassed cathode resistance Rk. However, at higher frequencies when the shunting effect of Ct on RL becomes appreciable, the reactance of Ck becomes comparable with Rk, and this reduces feedback to improve gain and maintain the frequency response. Condition for Maximum Flatness of Frequency Response. Gain of the above amplifier that employs cathode degeneration can be expressed in the form

A(mid) A(HF)

= 1 + jωCtRL + gmRk

1 + jωCt RL 1 + jωCk Rk

...(13.13)

*Necessary details of visual display system are given in Chapter 28.

VIDEO AMPLIFIERS—DESIGN PRINCIPLES

243
B+ RL RL B+

CC vin Rg RK CK C1 vin

CC

Rg RK LK CK

C1

(a)

(b)

Fig. 13.5. Widebanding by cathode compensation (a) Reactance of CK comparable to RK at medium and low frequencies, (b) LK and CK resonate at the upper edge of the video frequency band.

Differentiating Eqn. (13.13) with respect to ω and equating this equal to zero yields RLCt = RkCk ...(13.14) This is the condition that must be met for maximum flatness of the frequency response. As already stated the improvement in bandwidth occurs at the expense of overall gain. However, this sacrifice is worth it, because with negative feedback the amplifier gain becomes more stable and in addition there is a considerable reduction of distortion in the output of the amplifier. Another cathode compensation method is shown in Fig. 13.5 (b), where an inductor Lk in series with Ck shunts the cathode resistor. The values of Lk and Ck are so chosen, that the combination exhibits series resonance at the upper edge of the video frequency band. At resonance the very low impedance of the series tuned circuit effectively short circuits the feedback resistor and negative feedback is virtually reduced to zero. Typically, the ratio of cathode to plate circuit time constants falls in the range of 0.5 to 2.0.

13.3 LOW FREQUENCY COMPENSATION
In video amplifiers that employ ac coupling a large coupling capacitor is normally used to obtain a fairly low value of fL. No special low frequency compensation is thought necessary because of the annoying effects of too good a low frequency response. This aspect is fully explained in the next chapter. Direct Coupled Video Amplifier When direct coupling is used the question of low frequency compensation does not arise but the problems of high frequency response are the same as with RC coupled amplifiers. Though such amplifiers can amplify changes in dc level, they have other inherent problems of drift, need for a highly regulated power supply and the high voltage dc source for adjustments of voltages at the grid and cathode in the absense of ac coupling. All this adds to cost and therefore in many cases partial dc coupling is preferred. Selection of Tubes for Video Amplifiers The ability to provide high gain and to handle signals up to 5 MHz are the primary considerations in selection of tubes for use as video amplifiers. To achieve a high gain and large signal swings

244

MONOCHROME AND COLOUR TELEVISION

without excessive distortion, pentodes and beam power tetrodes having high current and large power dissipation ratings are preferred. Figure of Merit The figure of merit of a high frequency tube is defined as the product of gain and bandwidth. This can be expressed as: Gain × bandwidth = A(mid) × (fH – fL) Substituting for A(mid) = gmRL and setting (fH – fL) ≈ fH = Figure of Merit = gm × RL ×

1 we get, 2πCt RL
...(13.15)

1 gm = 2πCt RL 2πCt

Obviously, larger the value of this expression, better is the tube for use as a videoamplifier. However, because of large power needs, and the consequent large electrode structure, the output capacitance (C0) of such tubes cannot be made very small. This reduces the figure of merit of such tubes (fH reduces) and it becomes necessary to provide HF compensation to achieve the desired bandwidth. PCL84 is one such tube which has been specially designed for use in TV receivers. The pentode section of this tube is used as a video amplifier whereas the triode section is connected as a cathode follower for feeding video signal to AGC and sync circuits. Video Amplifier Circuit Based on the design criteria developed in the earlier sections of this chapter, video amplifier design employing tube PCL84 has been more or less standardized. A typical circuit is drawn in Fig. 13.6 with component values labelled on it. The tube employs a load resistance to the order of 4 K-ohms and with a steady plate current close to 18 mA, the amplifier delivers enough peak-to-peak video signal to produce a full contrast picture.
+ 200 V 4K RL Lp×1 Lpy To cathode of picture tube Lp×2 L.C. 4.7 K 5.6 K 100 V Ry 22 K To triode section of PCL84 + 200 V 0.7 V 47W PCL84 + CK 3 KPF 4 mF 4 KPF 0.1 mF 20 K pot contrast control 56 K

The main design features of this video amplifier are summarized below : (i) The plate circuit contains both shunt and series compensating coils, that are mutually coupled to provide adequate frequency broadbanding. (ii) Cathode compensation is also provided by using a small cathode bypass capacitor. (iii) The amplifier is designed for a full gain of about 30. The screen grid voltage is varied to contol the gain and this serves as the contrast control. (iv) The amplifier is dc coupled and has excellent low frequency response. In many designs partial dc coupling is used for optimum results.

13.4 TRANSISTOR VIDEO AMPLIFIER
Gain requirement from both tube and transistor video amplifiers is usually the same, and varies between 25 to 60, depending on the video signal amplitude available at the output of video detector and the transfer characteristics of the picture tube. Transistor video amplifiers are almost always direct coupled. This not only solves the gain and phase shift problems at low frequencies, but also makes the use of large coupling capacitors unnecessary. Direct coupling in transistor circuits does not present any serious problems so far as dc supply is concerned, because the magnitudes of voltage needed are much less than in tube circuits. However, the output transistor must have a VCC supply of the order of about 150 volts for delivering a video signal of nearly 75 volts peak-to-peak to modulate the picture tube. Transistors for Video Amplifiers The output capacitance of transistors is comparable with that of tubes, but because transistors operate at lower impedances, the frequency and phase response in the collector circuit of an RC coupled transistor amplifier remains unaffected up to a higher frequency than in the tube plate circuits. The input capacitance of bipolar transistors is in general, much greater than that of tubes, but its effect on the previous collector circuit, can be nullified by using interstage emitter followers. In a transistor amplifier the upper frequency limit is determined not by stray and shunt capacitances, but by the reduction in current gain, as the cut-off frequency of the transistor is approached. Low power transistors amplify up to very high frequencies and the problem is one of limiting the bandwidth rather than extending it. However, this remark is not applicable to power transistors. The high signal voltage, that the last video amplifier stage is expected to deliver with restricted load resistance, needs large collector current operation. This in turn needs a transistor of about 2 watt rating with a high breakdown voltage. When the above two conditions are met, the desired gain at high frequencies cannot be easily achieved, because it is difficult to make transistors with lesser collector to base capacitance and high junction breakdown ratings. Therefore, it becomes necessary to use peaking coils in the collector circuits of transistor video amplifiers to extend the high frequency range. Amplifier Configuration One high gain, high frequency transistor could by itself provide the required gain and bandwidth, but input and output impedance requirements make a single stage transtor amplifier difficult

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to design. Therefore, all video amplifier configurations are preceded by a driver stage, connected as an emitter follower. The driver connects the video detector to the output stage and meets the following requirements: (i) It presents a high input impedance to allow the use of a high detector load (about 5 KΩ). This ensures higher detector efficiency and more output voltage. (ii) It has low output impedance which facilitates matching to the input of the video amplifier transistor. (iii) Since the gain of the driver is less than one, it has a large bandwidth to ensure full transmission of video and intercarrier sound signals. (iv) The driver output is in phase with its input, and thus provides the correct polarity of video signal for cathode modulation of the picture tube after one stage of amplification.

13.5 TRANSISTOR CIRCUIT ANALYSIS
Common emitter circuit arrangement is the best suited configuration as a video amplifier, because of its moderate input and output impedances, high voltage and current gains besides a large power output. Figure 13.7 shows hybrid-pi model of a transistor in the common emitter mode. The equivalent circuit (model) has been simplified by reflecting, collector to base junction capacitance (Ccb or Cµ), to the input loop and neglecting the high collector to base resistance (rb′ c, or rµ).
rx B vbe vb¢e B C

rp

Cp

Cm gmvb¢e E

rce

RL

Cin = Cp + Cm = Cp + Cm (1 + gmRL)

Fig. 13.7. Hybrid-pi(π) model of a transistor in common emitter configuration.

In the equivalent circuit (See Fig. 13.7): (i) rx is the base spreading resistance expressed as a lumped parameter. Its value varies between 50 to 150 ohms. (ii) rπ is base to emitter junction resistance. Note that be common emitter input resistance in the ‘h’ parameter model, i.e., hie = rx + rπ. (iii) gm (mA/V) is a constant of proportionality between collector current and base-emitter voltage. It varies with collector current and is governed by the relation KT I gm = C , where VT = (VT = 0.026 V at room temperature.) V VT (iv) Cπ is the sum of diffusion capacitance and emitter to base junction capacitance.

(v) β is the short circuit current gain of the transistor at low frequencies. It varies with collector current and falls-off rapidly at collector currents beyond 10 mA.

VIDEO AMPLIFIERS—DESIGN PRINCIPLES

247 β = gm × rπ ...(13.16)

when β decreases, both gm and rπ are effected, and get reduced. (vi) rce is collector to emitter resistance. (vii) Cm (reflected Miller capacitance) = Cµ(1 + Av) where Av is the voltage gain of the stage. (viii) Cin (total input capacitance) = Cπ + Cµ(1 + Av) (ix) RL is external load resistance. (x) v′be is the effective voltage between base and emitter. fβ (Half Power Frequency) The circuit of Fig. 13.7 takes the form shown in Fig. 13.8 (a) when RL is set equal to zero. Note that rce disappears from the circuit once RL is made zero. The input side of the circuit has a single time constant, consisting of rπ in parallel with Cin. From this, 3 db down frequency fH = fβ =
RS B rx B¢

When f is set equal to fβ, the short circuit current gain drops by 3 db (see Fig. 13.8 (b)). This frequency fβ, at which the short circuit current gain becomes 70.7% of its maximum value is the half-power or—3 db frequency. The frequency range up to fβ is then referred to as the bandwidth of the circuit. fT (Unity Current Gain Frequency) fT is defined as the frequency at which Ais attains a magnitude equal to unity, that is: 1= or

βfβ fT

(from Eqn. 13.19) ...(13.20)

fT = β fβ Substituting values of β and fβ from (13.16) and (13.18) in (13.20), we get fT =
gm 2πCin

The above expression shows that fT is a function of the transistor parameters only. Since fT controls the gain at high frequencies it is also known as ‘Figure of Merit’ of the transistor. Most of the above parameters are listed in transistor manuals. However, some parameters, if not given, can either be measured or calculated from the various relations given above. Voltage Gain of the Basic Amplifier The circuit of the basic amplifier employing a BJT (transistor) in common emitter configuration is shown in Fig. 13.9 (a). In its equivalent circuit (Fig. 13.9 (b)), biasing resistance RB has not been included, because its shunting effect on input impedance of the transistor is negligible. Similarly rce being very large, in comparison with RL has been neglected.
VCC RB RS vS CT RL v0 vS

If follows from this expression that | A(mid) | = Gain Bandwidth Product The corner frequency occurs when the reactance of Cin (= Cπ × D) equals Rth, where Rth is the Thevenin’s equivalent of the circuit (Fig. 13.9 (b)) to the left of points a and b. Rth = where RS is the source resistance. On equating X Cin = Rth and some manipulation we get :
rπ ( RS + rx ) rπ + RS + rx

....(13.26)

...(13.27)

B 1 where ωβ = and the other factor rπ Cπ
B=

fH =

βfβ

or ωH =

βω β B

rx + rπ + RS RS + rx

...(13.28)

fH (= bandwidth) can also be expressed as

or

f B × T (since fT = fβ × β) β D B ωT × ωH = D β
fH =

...(13.29)

250 Finally, Gain × Bandwidth = A(mid) × ωH =

MONOCHROME AND COLOUR TELEVISION

βRL B ωT × × (from Eqs. (13.26) and (13.29)) rπ + rx + RS β D

Substituting for B from Eqn. (13.28) G×B.W=

ωT RL × D RS + rx

...(13.30)

13.6 GUIDELINES FOR BROAD-BANDING
Equation (13.30) serves as a guideline for explaining the means to extend high frequency response of the amplifier with or without sacrificing midband gain. This is explained by considering separately all the constituents of the Gain-Bandwidth expression. (a) The gain-bandwidth product increases with decrease of source resistance RS. Thus a reasonable first step while designing a video amplifier is to choose the lowest possible value of RS. This requirement is readily met, since the driver stage is an emitter follower, and its output resistance can be made very low without any appreciable loss in gain. (b) ωT (fT = βfβ) does not stay constant and drops-off both at very low and high values of emitter current. However, there is a range of IE (emitter current) over which it stays high and substantially constant. Therefore, it is advantageous to fix the transistor operation, in this region, as far as possible. (c) Once the transistor and its operating point (IE or IC) have been chosen, rx gets fixed and cannot be varied. (d) If Cµ, that forms part of factor D is decreased, the bandwidth will increase with no corresponding loss in gain. However, the value of Cµ is dictated by the VCC supply chosen or the maximum collector voltage rating of the transistor. Therefore, Cµ is more or less fixed and cannot be changed for extending high frequency region of the amplifier. (e) The last variable is RL, that can be changed to control bandwidth. But, this too cannot be varied much because of large peak-to-peak output voltage required to modulate the picture tube, and the maximum permissible dissipation of the transistor. Output Circuit Corner Frequency Input capacitance of the picture tube, together with transistor output and wiring capacitances easily add up to about 15 pF and very much limit the value of RL. In fact the net output capacitance (CT) of the video amplifier in parallel with RL provides another corner frequency which turns out to be lower than the input circuit corner frequency. This, then controls the bandwidth of the amplifier. As stated above, decreasing RL will push up the output corner frequency, but the load resistance cannot be made too small because it will increase the power dissipation of the device.

VIDEO AMPLIFIERS—DESIGN PRINCIPLES

251

13.7 FREQUENCY COMPENSATION
It is clear from the above discussion that a high fT and high power dissipation transistor is necessary for video amplifiers. These two requirements, though necessary, are mutually contradictory to a large extent. In the past these requirements were met by cascoding, where the power dissipation was shared equally by the two transistors employed in such a configuration. With advances in technology, transistors with high power dissipation and reasonably high fT have now become available. However, the configuration still requires some high frequency compensation and is normally provided by a shunt or peaking coil in the collector circuit. This and other relevant details are explained by a design example. Video Amplifier Design Data Output voltage Bandwidth Detector output voltage Voltage gain Configuration Coupling VCC supply Transistor 75 V (p-p) 5 MHz 2 V (p-p) 40 Common Emitter Direct 150 V BF 178

Transistor parameters at IC = 15 mA are : rx = 50 ohm, rπ = 200 ohms, Cπ = 100 pF, β = 20, fT = 120 MHz, Cµ at 150 V = 1.25 pF, max collector dissipation = 1.7 watts, minimum collector emitter breakdown voltage = 145 V. Choice of RL and Operating Point To avoid non-linear distortion due to saturation and cut-off, the best course for such a large output is to draw several load lines on the characteristics of the chosen transistor and calculate distortion for each load by the usual three or five point analysis. This would help to decide the optimum value of RL. Note that too small and too large a value of load resistance is not acceptable for reasons already explained. Such an exercise on the characteristics of BF178 led to the following results: IC ≈ IE = 15 mA RL = 4.9 K VCE (min) inclusive of drop across RE = 30 V (RE is the emitter resistance) This leaves 120 V to accmmodate the output signal with enough margin for the blanking excursion. For class ‘A’ operation the following relations are valid: Pmax =
2 VCC 4( RL + RE )

...(13.31)

252 IE =
VCC 2( RL + RE )

MONOCHROME AND COLOUR TELEVISION

...(13.32)

Choosing RE = 100 ohms and allowing a 10% limit in VCC variations, i.e., VCC max = 165 V, Pmax from eqn. (13.31) = 1.36 watts and IC ≈ IE from eqn. (13.32) = 15 mA. The calculated value of IC checks with that found graphically. Similarly the calculated value of Pmax is within the max. permissible dissipation. However, the transistor would need a suitable heat sink and this is always provided. Amplifier Circuit The circuit of the amplifier is drawn in Fig. 13.10 (a) and its equivalent circuit valid at high frequencies is shown in Fig. 13.10 (b). Besides other circuit elements the two corner frequencies (break points) are labelled as f, 3 db (in) and f, 3 db (out) in the equivalent circuit.
+ VCC

Voltage Gain Equation (13.26) can be modified to include the effects of biasing network and the inadequately bypassed RE. Putting Rb = R1 || R2 (biasing network resistors) and adding (1 + β)RE ≈ βRE (reflected emitter resistance) to rπ, the new voltage gain can be calculated. With the given design values even if Rb is taken as 7 K it can be neglected in comparison with the other shunting resistances. With this assumption the new voltage gain βRL | Av(mid) | = ...(13.33) Rs + rx + rπ + βRE

This result shows that the input circuit does not need any compensation and would safely transmit up to the highest modulating frequency of 5 MHz. Output Circuit Corner Frequency (f3 db (out)) The total collector network capacitance in a well laid out receiver would by typically as follows: Picture tube cathode and leads Heat dissipator BF 178 output capacitance Total capacitance CT ∴ f3 db (out) (uncompensated) = Frequency Compensation The above result shows that the output circuit has a lower corner frequency and hence would determine the extent of compensation needed to push this corner frequency to about 5 MHz. The design criteria for calculating the values of peaking coils are the same as used in tube circuits. For shunt compensation: From equation 13.9 we have Lpx = 0.5 CTRL2 On substituting RL = 4.9 kΩ and CT = 13 pF Lpx = 0.5 × 13 × 10–12 × (4.9 × 103)2 ≈ 158 × 10–6 H = 158 µH An inductor of this value will form part of the collector load to provide necessary compensation. = 7 pF = 3 pF = 3 pF = 13 pF

1 1 = ≈ 2.6 MHz. 2π × 4.9 K × 13 pF 2πRL CT

254 Broad-Banding by Negative Feedback

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In addition to shunt compensation the emitter resistance RE is bypassed by a very small (300 pF) capacitor CE (see Fig. 13.10 (a)) to provide negative feedback at medium and low video frequencies. This not only improves the bandwidth but also ensures stable operation of the amplifier. The value of CE is determined by the consideration that emitter and collector network time constants should be approximately equal.

13.8 VIDEO DRIVER
The output resistance of the driver together with the input resistance and capacitance of the video transistor form a network whose time constant should be compatible with the required bandwidth. Assuming 3 db point at 5 MHz and with Cin = 180 pF R(out) ≤

10 12 ≈ 150 ohms 2π × 5 × 10 6 × 180

Therefore, ideally, ignoring Rin, the output resistance of the driver stage should not exceed this value. This checks with the value of RS used while determining midband gain and input corner frequency. However, in most practical designs. a value of 500 ohms can be used, because of increase in bandwith available by partially decoupling the emitter resistance of the output transistor. The emitter follower will have a gain nearly equal to 0.9. This will feed about 1.8 V video signal to the output transistor which in turn will deliver ≈ 75 (p-p) (gain ≈ 40) for the picture tube. Video Detector Loading A high frequency transistor like BF 184 having β = 75 and fT = 300 MHz, if employed as an emitter follower with RE = 470 ohms will have Rin = 75 × 470 = 35 KΩ, and Cin = 5 pF. This is acceptable to a video detector circuit having a 3.9 KΩ load resistance. Video Driver Biasing The bias in the driver stage must be carefully set to permit maximum collector voltage swing. With a high input signal, the output will clip if the transistor cuts off or if VCE reaches zero. The result is loss of detail in dark grey or white parts of the picture and a buzzing tone in the sound output. For this reason the upper biasing resistance is often a pre-set variable resistor. The video detector diode is invariably direct coupled to the driver and thus held at the same steady bias voltage as the base of the emitter follower. The biasing network is often designed to provide a small forward bias on the diode to reduce distortion on small input signals.

13.9 CONTRAST CONTROL METHODS
Contrast control is a manual control for setting level of the video signal fed to control grid or cathode of the picture tube. Its setting determines the ratio of light to dark in the picture.

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255

Contrast Control in Vacuum Tube Video Amplifiers (a) Cathode Network Control. In this method negative feedback is applied in one form or the other, taking into account the biasing requirements of the tube. Figure 13.11 (a) shows one such method, where the adjustment of the contrast control does not change the operating point of the amplifier. This maintains a constant black level of the video signal. (b) Plate Network Control. In this arrangement (Fig. 13.11 (b)) the contrast control potentiometer regulates the magnitude of the video signal to the picture tube. The stray capacitance of the potentiometer and its connecting leads can reduce high frequency response of the amplifier. To minimize shunt capacitance the control is usually mounted close to the video amplifier with the shaft mechanically coupled to the front panel of the receiver. In addition, the capacitors shown along with the potentiometer, provide frequency compensation to maintain the same frequency response, at different settings of the contrast control.
CC To plate circuit Video input Rg Contrast control C 1 RK1 RK RK2 (b) (a) B+ Video input Rg CK B+ LPX RL B+ B– C1 C2 C3 To picture tube circuit R3

Contrast Control in Transistor Circuits (a) Base Network control. A contrast control technique that maintains a constant black level is shown in Fig. 13.12 (a). If the values of R3 and R4 are chosen to give a voltage that is equal to the black level of the video signal at the emitter of Q1, the black level of the signal fed to Q2 will remain constant over the contrast control range. Because of bandwidth requirements, R2 should not be higher than 1 K-ohm. The parallel value of R3, R4 should be about one quarter of the value of R2 thus giving a contrast range of about 5 : 1. (b) Emitter Network Control. Fig. 13.12 (b) shows one type of emitter network contrast control. It is a degeneration control. When the arm of potentiometer R4 is at ground, its resistance is unbypassed causing maximum feedback. This results in small video output. Any variation of the arm towards the emitter reduces feedback to deliver more output. This provides the desired contrast control. C1R1, and C2R2 are video peaking networks which cause higher gain at low contrast settings for high frequencies, making the picture sharper. (c) Collector Network Control. As shown in Fig. 13.12 (c) the 25-K frequency-compensated potentiometer operates like a volume control. Setting of contrast control determines the peakto-peak amplitude of the video signal taken from the collector of the video amplifier and

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coupled to the cathode of picture tube. Because of the high impedance level of the collector network, stray capacitances place severe restrictions on the circuit layout. The compensation should be so provided that for any setting of the contrast control almost equal time constants are obtained in the two arms of the connecting network.
+ VCC2 + VCC1 Lpx RL Q2 a¢ R2 R4 CE RE To picture tube circuit

13.10 SCREEN SIZE AND VIDEO AMPLIFIER BANDIWDTH
It may not be immediately apparent but the size of the screen on which the picture is reproduced will also govern how much finer details the picture should possess. On a small scrren the number of active lines that carry video information are very close to each other. This disables the eye to distinguish fine details, unless the viewer comes very close to the screen. The reason stems from the fact, that unless the two adjacent objects subtend an angle of one minute or more at the observer’s eye, they cannot be seen as distinct units. With small screens the distance necessary for the eye to resolve details is so short that the viewer normally never comes that close to the screen. Manufacturers of small screen TV receivers take advantage of this fact and design video amplifiers with a bandwidth that is much less than 5MHz. This reduction, in the highest modulating frequency to be reproduced, also makes full bandwidth in the RF and IF tuned amplifiers unnecessary. All this results in considerable reduction in the overall cost of the receiver which is a big factor in a highly competitive consumer goods industry.

14
Video Amplifier Circuits
The essential requirements of video section circuitry and general wideband techniques were explained in the previous two chapters. The merits of cathode over grid modulation of the picture tube were brought out in an earlier chapter. This is now the most preferred method of feeding video signal to the picture tube unless there are strong reasons in favour of grid modulation. However, there are several methods of coupling the video amplifier to the picture tube. This, together with other relevant circuit details, is discussed in this chapter. Various Coupling Methods. Though it cannot be denied that for near perfect reproduction of the transmitted picture, dc link has to be maintained between the video detector and picture tube, but dc coupling has its own problems which when fully taken care of add to the cost of the receiver. Therefore in many video amplifier designs full dc coupling is dispensed with yet maintaining optimum reception which is subjectively acceptable. The various possible coupling arrangements between the video amplifier and picture tube may be classified as: (a) DC coupling (b) Partial dc coupling (c) AC coupling with dc restoration (d) AC coupling Though the circuit details differ from chassis to chassis, typical circuits of each type are examined to identify merits and demerits of the various types of coupling.

14.1 DIRECT COUPLED VIDEO AMPLIFIER
A commonly used video amplifier employing PCL 84 (pentode-triode) is shown in Fig. 14.1. The video signal is directly coupled from video detector to cathode of the picture tube. The main features of this circuit are as follows: (i) Frequency Compensation The plate circuit contains both shunt and series peaking coils to provide enhanced high frequency respouse. Additional broadbanding is achieved by using a small (0.003 µF) cathode bypass (Ck) capacitor. The network L1R1 in the grid circuit of the tube provides frequency compensation to offset its input capacitance. (ii) Contrast Control Gain of the amplifier is controlled by varying dc voltage (potentiometer R6) at screen grid of the pentode. This becomes the contrast control. The need for two decoupling capacitors at the screen grid arises from the fact that electrolytic capacitors have a small self-inductance which 260

(iii) Sync and AGC Take-off Points The triode section of PCL 84 is connected as a cathode follower. The resistors R4 and R5 form a potential divider at the cathode of the triode to feed necessary video voltage to the sync separator circuit. AGC circuit is fed directly from output of the cathode follower. The use of cathode follower avoids any loading effects from sync separator and AGC circuits and thus fully isolates the video amplifier from these circuits. In the absence of such a provision some additional capacitance appears across the output of the amplifier and tends to lower its high frequency response. (iv) Flyback Suppression Pulses Field and line flyback suppression pulses are injected at the control grid and first anode of the picture tube through isolating networks. These negative going pulses are of sufficient amplitude to cut-off the beam current during flyback intervals. (v) Brightness Control and Switch-off Spot Brightness control is achieved by varying positive voltage at the grid (G1) of the picture tube with potentiometer R8 that is connected in series with a VDR (voltage dependent resistance) across B+ supply. The VDR has a special function to perform. When the receiver is switched-off the time-base circuits stop immediately and the picture tube spot assumes central position on

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the screen. The cathode and grid potentials rapidly becomes equal to chassis potential since the B+ voltage disappears. However, the picture tube’s cathode remains hot for some time and keeps emitting electrons. At the same time the EHT capacitance formed by the aquadag coating of the tube does not immediately discharge since the resistance of the EHT circuit is very high. For a few moments, therefore, beam current continues to flow with zero grid-cathode bias and no deflection fields. A bright spot known as ‘switch-off spot’ appears on the screen centre, which can in due course burn a small portion of the phosphor coating on the screen. Suppression of the switch-off spot is brought about in this circuit by the VDR which forms the lower arm of the brightness control potential divider. When the receiver is switched off and the B+ voltage disappears, the resistance of the VDR immediately becomes very high. This allows the charge on the associated capacitor C3 to remain for a short time so that the grid is momentarily positive with respect to cathode. The result of this is that a high beam current passes for a brief instant and this discharges the EHT smoothing capacitor. This happens as the normal B+ voltage is decreasing and before the raster finally collapses. Thus the heavy beam current is spread over the screen face and not concentrated on a central spot.

14.2 PROBLEMS OF DC COUPLING
Direct coupling, though very desirable has the following stringent requirements, which, when provided for add very much to the cost of the receiver. (a) Regulated EHT Supply Regulation of normal type of EHT systems used in most television receivers is not good. Full contrast range to be handled by the picture tube with dc coupling puts a heavy demand on the high voltage supply. On signal levels that correspond to peak whites, excessive beam current flows and this results in a drop of voltage. In turn this tends to an instantaneous increase in deflection sensitivity so that the picture expands (blooms) as the voltage falls. The increase in deflection sensitivity is due to the fact that as EHT voltage falls, forward velocity of the electrons, that constitute beam current, decreases. The electrons then spend a longer time under the deflection field, and are deflected more by a given field than they would normally be. Therefore, either a well regulated EHT system should be provided, or the natural range of brightness levels, which occur in the original transmitted picture, should be artificially restricted at the receiver. The latter, that is, partial loss of dc component of the video signal is lesser of the two evils and involves a commercial compromise. This is explained in another section of this chapter. However, as stated earlier, in terms of absolute picture fidelity, both the complete retention of dc component and a well regulated EHT system are necessary. (b) Beam Current Limiting In a dc coupled video stage, for cathode injection, the picture tube can be driven to a high brightness level if the input signal is removed. This increase in beam current is expensive both in high voltage supply source and the life of the picture tube. A simple circuit which limits the mean beam current to a pre-determined value (without limiting the contrast range) is shown in Fig. 14.2 (a). With this circuit arrangement, dc coupling is maintained only on low key scenes, where it is most important. In this circuit, V3 = (Id + Ib)R1 = IdR1 + IbR1,

At the threshold of limiting, Ib × R1 = V2 = V3 Therefore Id = 0 Beyond this threshold V2 < V3 and the capacitor C1 charges to V3 – V2, that is, the picture tube drive is now ac coupled. The picture tube, therefore, receives an additional backbias proportional to the excess mean drive. It may be noted that it is not possible to establish precise limiting threshold because the mean value of V2 varies with the picture content. However, by a suitable choice of the value of R1, excess of picture tube EHT current and the consequent blooming (or breathing) of the picture are prevented. In video circuits, that employ partial dc coupling, beam current is automatically reduced, making use of a diode unnecessary. (c) Other Direct Coupling Problems Besides the need for a regulated EHT supply and beam current limiting, the complexity in direct coupled amplifiers arises on account of the following : (i) It is necessary to have a stable (regulated) B+ source to avoid any drift in the output of the amplifier. (ii) The reflections from any passing aeroplanes result in a steep rise and fall of input signal at the antenna of the receiver. This, despite an efficient fast acting AGC, causes a momentary flutter of the reproduced picture. It occurs because of very good low frequency response of the amplifier that extends down to zero Hertz. (iii) There is a possibility of heater to cathode insulation breakdown during picture highlights because of high dc voltage, equal to the plate voltage that appears on the cathode of the picture tube. (iv) For cathode injection, if the detector output is directly coupled to the video amplifier, the latter must be biased to conduct heavily when no signal is present. This is expensive both in B+ current and life of the device.

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14.3 PARTIAL DC COUPLING
As mentioned earlier the solution to direct coupling problems, as a compromise, lies in attenuating the dc voltage before applying it to picture tube and providing a low frequency bypass to reduce some of its annoying effects. This arrangement is a common feature in most television receivers and is known as ‘partial dc coupling’. The relevant portion of a transistor video amplifier is shown in Fig. 14.2 (b). In this circuit (i) C1, R1 and R2 constitute dc attenuator and low frequency filter circuit. The long time constant circuit R1, C1 in series with video signal path to the picture tube cathode makes the aeroplane flutter effect less annoying. The capacitor C1 (0.1 µF) fails to bypass very low frequencies, with the result, that R1 provides series attenuation in the signal path to offending low frequency pulsations. As obvious, low frequency components of the video signal get attenuated by a factor R2/(R1 + R2), and this is what gives the circuit the name ‘partial dc coupling’. (ii) The values of R1 and R2 are chosen so as to considerably reduce the dc voltage that reaches the cathode of picture tube. This not only attenuates the low frequency content of the
2 video signal but the consequent reduction in the working dc voltge VCO × R + R at the cath1 2

FG H

R

IJ K

ode reduces the magnitude of dc voltages required at the accelerating and focusing anode of the picture tube. This in turn results in a saving in the cost of power supply circuit. (iii) The reduction in the cathode voltage reduces any possibility of heater cathode breakdown of the picture tube. (iv) Since the dc component is partly removed by potential divider action of R1 and R2, the difference in the mean level brightness from scene to scene is reduced. This restricts the overall range to be handled by the tube and hence limits the maximum demand that is made on the EHT system. Video Amplifier Circuit A transistorised video amplifier circuit with emitter follower drive and partial dc coupling is shown in Fig. 14.3. The salient features of this circuit are : (i) Signal from the video detector is dc coupled to the base of Q1. This transistor combines the functions of an emitter follower and CE amplifier. The high input impedance of emitter follower minimizes loading of the video detector. The sync circuit is fed from the collector of this transistor, where as signal for the sound section and AGC circuit is taken from the output of the emitter follower. (ii) The output from the emitter follower is dc coupled to the base of Q2. This is a 5 W power transistor, with a heat-sink mounted on the case. The collector supply is 140 V, to provide enough voltage swing for the 80 V P-P video signal output. (iii) In the output circuit of Q2, contrast control forms part of the collector load. The video output signal is coupled by the 0.22 µF (C2) capacitor to the cathode of picture tube. The partial dc coupling is provided by the 1 M (R2) resistor connected at the collector of Q2. (iv) The parallel combination of L1 and C1 is tuned to resonate at 5.5 MHz to provide maximum negative feedback to the sound signal. This prevents appearance of sound signal at the output of video amplifier.

(v) The neon bulb in the grid circuit provides protection of a spark-gap since the neon bulb ionizes and shorts to ground with excessive voltage. The ‘spark gaps’ are employed to protect external receiver circuitry from ‘flash overs’ within the tube. The accumulation of charge at the various electrodes of the picture tube results in the appearance of high voltates at the electrodes, which if not discharged to ground, will do so through sections of the receiver circuitry and cause damage. (vi) Note that dc voltages at the base and emitter of the two transistors have been suitably set to give desired forward bias. (vii) Vertical retrace blanking pulses are fed at the grid of the picture tube through C3, and the grid-return to ground is provided by R3. (viii) Brightness control. The adjustement of average brightness of the reproduced scene is carried out by varying the bias potential between cathode and control grid of the picture tube. In the circuit being considered a 100 KΩ potentiometer is provided to adjust dc voltage at the cathode. This bias sets correct operating point for the tube and in conjunction with the video blanking pulses cuts-off the electron beam at appropriate moments. The setting of grid bias depends upon the strength of signal being received. A signal of small amplitude, say from a distant station, requires more fixed negative bias on the grid than a strong signal. The dependency of picture tube grid bias on the strength of the arriving signal is illustrated in Fig. 14.4. For a weak signal, the bias must be advanced to the point where combination of the relatively negative blanking voltage plus the tube bias drives the tube into cut-off. However, with a strong signal the negative grid bias must be reduced, otherwise some of the picture details are lost. Since the bias of the picture tube may required an adjustment for different stations, or under certain conditions from the same station, the brightness control is provided at the front panel of the receiver.

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Picture tube transfer characteristics

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Average brightness

Black –90 –60 –30 DC bias 0

VGK

Weak signal

Reduced bias

Fig. 14.4. Optimum setting of contrast control for different amplitudes of the video signal.

The effects of brightness and contrast controls described earlier overlap to some extent. If setting of the contrast control is increased so that the video signal becomes stronger, then the brightness control must be adjusted to meet the new condition, so that no retrace lines are visible and the picture does not look milky or washed out. Too small a value of the negative grid bias allows average illumination of the scene to increase thus making part of the retrace visible. In addition, the picture assumes a washed out appearance. Too low a setting of the brightness control, which results in a high negative bias on the picture tube grid, will cause some of the darker portions of the image to be eliminated. Besides this overall illumination of the scenes will also decrease. To correct this latter condition, either the brightness control can be adjusted or the contrast control setting can be advanced until correct illumination is obtained. If the brightness control is varied over a wide range the focus of the picture tube may be affected. However, in the normal range of brightness setting made by the viewer, changes in focus do not present any problem. It is now apparent that despite the fact that video signal, as received from any television station, contains all the information about the background shadings of the scene being televised, an optimum setting of both contrast control and brightness control by the viewer is a must to achieve desired results. Many viewers do not get the best out of their receivers because of incorrect settings of these controls. However, to ensure that retrace lines are not seen on the screen due to incorrect setting of either contrast or brightness control, all television receivers provide blanking pulses on the grid electrode of the picture tube.

Brightness
Strong signal

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267

14.4 CONSEQUENCES OF AC COUPLING
As already explained, dc coupling though desirable, adds to the cost of receiver. Partial dc coupling does not reduce fully the circuit complexity and other side effects of dc coupling. This suggests the use of ac coupling from video detector to picture tube. However, before doing so, it would be desirable to review the consequences of ac coupling. Figure 14.5 (a) shows video signals for two lines taken at different moments from a television broadcast. One signal represents a line from a predominently white picture while the other belongs to a black background. As they come out of the video detector, their sync pulses are aligned to the same level. When amplified by a dc coupled amplifier, they get inverted but retain their common blanking level (Fig. 14.5 (b)). At the picture tube, with a suitable fixed bias, black levels of the two signals automatically align themselves along the beam current cut-off point. This happens because of different dc contents in the two signals. Thus dc components of video signals enable scenes with different background shadings to be correctly reproduced on the raster without having to change the setting of brightness control.
0 t

Now consider that these signals are passed through a capacitor as would be the case if ac coupling were employed. This is illustrated in Fig. 14.6 by an equivalent circuit and associated waveshapes. In the equivalent circuit (see Fig. 14.6 (a)) dc component of each signal has been represented by a battery and the ac component by a generator. The combined signal feeds into an RC circuit. On application of any composite signal the coupling capacitor will charge to a value equal to the battery voltage. However, the ac video content will cause the capacitor to

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charge and discharge as the applied voltage exceeds and falls around the dc voltage across the capacitor. Thus, while the dc component is blocked by the capacitor, the current which continuously flows through the load resistance develops an ac voltage drop across it. The resulting output waveshapes and their locations along with corresponding input waveshapes are shown in Figs. 14.6 (b) and (c). Note that while the output waveshapes are almost identical to their input counterparts, their sync and blanking levels no longer align with each other. Each signal has set itself around the zero axis as a consequence of ac coupling. This leads to the following undesired effects. (i) Visible Retrace Lines The retrace lines become visible because the blanking pulses do not remain at a constant level and may not have enough amplitude to cause retrace blanking. Most modern receivers, irrespective of the coupling employed, incorporate special vertical and horizontal retrace blanking circuits as a means of preventing retrace lines from becoming visible.
+ vc C Video signal’s Input ac component voltages and dc component v0 Output voltages –

(ii) Possible Loss of Sync This occurs due to loss of dc component. The sync pulse amplitudes now vary with picture content and this can lead to inadequate amplitude of sync pulses at the output of sync separator.

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269

Therefore, if synchronization is to remain stable, the dc level must be restored (sync pulses must line up) before the video signal is fed to the sync separator circuitry. (iii) Loss of Average Brightness of the Scene This means that bright and dark scenes may not be easily distinguishable. With loss of dc component the average brightness information is lost. Thus signals from different brightness backgrounds will lose this identity and reproduce pictures with a background of some grey shade. (iv) Poor Colour Reproduction In colour television a change in the luminance (brightness) signal will cause a change in the brightness of a colour. Therefore loss of dc component of the video signal will result in poor colour reproduction.

14.5 DC REINSERTION
As explained earlier, relative relationship of ac signal to the blanking and sync pulses remains same with or without the dc component. Furthermore, brighter the line, greater is the separation between the picture information variations and the associated pulses. As the scene becomes darker, the two components move closer to each other. It is from these relationships that a variable bias can be developed to return the pulses to the same level which existed before the signal was applied to the RC network. DC Restoration with a Diode A simple transistor-diode clamp circuit for lining up sync pulses is shown in Fig. 14.7 (a). The VCC supply is set for a quiescent voltage of 15 V. In the absence of any input signal the coupling capacitor ‘C’ charges to 15 V and so the voltage across the parallel combination of resistor R and diode D will be zero. Assume that on application of a video signal, the collector voltage swings by 8 V peak to peak. The corresponding variations in collector to emitter voltage are illustrated in Fig. 14.7 (b). The positive increase in collector voltage is coupled through C to the anode of diode D, turning it on. Since a forward biased diode may be considered to be a short, it effectively ties (clamps) the output circuit to ground (zero level). In effect, each positive sync pulse tip will be clamped to zero level, thereby lining them up and restoring the dc level of the video signal.
+ VCC RL + 15V + vin VCE C – R – 4V + D v0 19V 15V 11V 0 (a) (b) VCO t (c) VCE + v0 0 –4 –8 DC level t Clamp level

In the case under consideration the diode will cause the coupling capacitor to charge to a peak value of 19 V. However, during negative excursion of the collector voltage the capacitor fails to discharge appreciably, because the diode is now reverse biased and the value of R has been chosen to be quite large. The average reverse bias across the diode is – 4 V, being the difference between the quiescent collector voltage and peak value across the capacitor. Note that as the input video signal varies in amplitude a corresponding video signal voltage appears across the resistor R and it varies in amplitude from 0 to – 8 V (peak to peak). This, as shown in Fig. 14.7 (c), is the composite video signal clamped to zero. Similarly as and when the average brightness of the scene varies the capacitor C charges to another peak value thereby keeping the sync tips clamped to zero level. Reversing the diode in the restorer circuit will result in negative peak of the input signal being clamped to zero. This would mean that the dc output voltage of the circuit will be positive. The video signal can also be clamped to any other off-set voltage by placing a dc voltage of suitable polarity in series with the clamping diode. Limitations of Diode Clamping It was assumed while explaining the mechanism of dc restoration that charge across the coupling capacitor C does not change during negative swings of the collector voltage. However, it is not so because of the finite value of RC. The voltage across C does change somewhat when the condenser tends to discharge through the resistor R. Another aspect that merits attention is the fact that whenever average brightness of the picture changes suddenly the dc restorer is not capable of instant response because of inherent delay in the charge and discharge of the capacitor. Some receivers employ special dc restoration techniques but cost prohibits their use in average priced sets.

14.6 AC COUPLING WITH DC REINSERTION
Figure 14.8 is the circuit of a video amplifier employing ac coupling with dc restoration. The coupling capacitor C1 offers negligible reactance to the high frequency content of the video signal and it gets coupled to the picture tube grid directly. With no input signal, C2 charges to the steady dc potential existing at ‘X’ through R4, R6 and R7. With arrival of video signal the potential at ‘X’ falls during negative swing of the collector voltage. This causes C2 to discharge through R4, R3, VCC source, R7 and diode D1. However during positive voltage swing at the collector, C2 fails to regain its charge because during this interval the diode is reverse biased and R6 has been chosen to be too large. Thus the reduced voltage across C2 is maintained at this level during intervals between sync tips because of the relatively large value of C2 and associated resistors. The resultant difference of potential between ‘X’ and VC2, that effectively appears between ‘Y’ and ground (see Fig. 14.8) gets applied to the grid via isolating resistance R5. This amounts to restoring dc component of the video signal, which otherwise is blocked by the coupling capacitor. When the average brightness of the scene increases the video sync tips move further away from the picture signal content and the point ‘X’ then attains a new less positive potential during sync tip intervals. This further lowers the potential across the capacitor C2. The enhanced difference in potential between the point ‘X’ and the new value of VC2 gets

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applied to the grid of the tube. This, being positive, reduces the net negative voltage between the grid and cathode and the scene then moves to a brighter area on the picture tube characteristics. Any decrease in average brightness of the scene being televised will have the opposite effect and net grid bias will become more negative to reduce background illumination of the picture on the raster. Thus the diode with the associated components serves to restore the dc content of the picture signal and the difference in potential between ‘X’ and ‘Y’ serves as a variable dc bias to change the average brightness of the scene.
v0

Last video amp vin

R1

C1

0

t v0 To grid of picture tube

L1 X

L2 R4 –

C2 + VC2

R5 Y R6

R3 R2 + – VCC

D1

R7

Fig. 14.8. Practical dc restorer circuit.

14.7 THE AC COUPLING
Cost is a strong determining factor in the design of commercial television receivers. If it is possible to reduce the cost of a set without impairing the picture quality too much, then a sacrifice in quality for cost is justifiable and is made in some receiver designs. Therefore many receivers use only ac coupling. In other words the dc component is removed from the signal and never reinserted. The AC Coupled Video Amplifier Figure 14.9 shows an ac coupled video amplifier. The coupling capacitor (0.22 µF) and resistance of the brightness control network constitute the ac coupling network. The contrast control is located in the emitter circuit of the first video amplifier. It is also ac coupled to the output video amplifier. The amplifier employs the usual broadbanding techniques. It has a sound trap (resonant) circuit in the emitter lead of the output transistor. An interesting feature of this circuit is the provision of a spot-killer switch. This swith opens when the receiver is switched off. Its operation removes dc voltage at the cathode of picture tube reducing grid-cathode voltage to zero. The residual beam current increases and quickly discharges the EHT smoothing capacitor thereby reducing intensity of the switch-off spot.

(i) First Video Amplifier The output from video detector feeds at pin 9 and this is also the sound take-off point. The video preamplifier employs atleast two stages of differential amplifiers and is preceded by a

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driver stage to provide impedance matching. The voltage gain from this stage is about 70 db. Input signal from the detector is clamped at 3 V and the video output is obtained at pin 11 of the IC. The output drives video output transistor 2N 3501 through a contrast control network as shown in the figure. The VCC supply to the IC is a stabilized + 16 V derived/from the low voltage (LV) rectifier and filter network. (ii) Sync Separator The sync separator receives input from the video preamplifier and is suitably biased to deliver clean sync pulses. The circuit also employs a noise suppression circuit. Integrated vertical sync pulses are fed to the vertical oscillator through a capacitor from pin number 14 on the IC. The amplitude of the vertical sync output is around 11 V. (iii) AFC Circuit The horizontal AFC circuit employs a single ended discriminator. It derives sync pulse input from the sync separator and flyback pulses of opposite polarity from the horizontal output transformer at pins 4 and 10. The vertical blanking pulses are also added at pin 10 from the vertical output transformer. The AFC control voltage is available at pin 2 and is fed to the input of horizontal oscillator through an anti-hunt filter circuit. AFC output voltage ranges from 2 to 10 volts. (iv) AGC Circuit The IC includes a keyed AGC circuit and receives flyback pulses through pin 10 along with the AFC circuit. The AGC output voltage varies from 1 to 12 V and is fed to the IF section from pin 7 as shown in the figure. Delayed AGC voltage for the tuner is available at pin 6 and its amplitude varies from 0.3 to 12 V.

Review Questions
1. 2. Enumerate the various coupling methods employed between video detector and picture tube. Why does dc coupling add to the cost of the receiver ? Describe the main features of the dc coupled video amplifier shown in Fig. 14.1. What is a ‘switch-off ’ spot ? Explain how its undesirable effect on the screen is minimized by using a VDR in the brightness control circuit. 3. Explain briefly the essential requirements, which must be met, while providing dc coupling in the video section of the receiver. Describe with a suitable circuit diagram how a diode can be used to limit beam current of the picture tube to a safe upper limit. What do you understand by partial dc coupling ? Explain with a circuit diagram how some of the annoying features of dc coupling are almost eliminated by partial dc coupling. Justify that it is a reasonable compromise between cost and quality. Describe the consequences of ac coupling. Show with a suitable circuit diagram 1and illustrations how dc component of the video signal is restored back by diode clamping in an otherwise ac coupled video amplifier. Explain how the removal of dc component from the video signal affects the overall contrast range and necessitates repeated adjustments of the brightness control. State the arguments that are advanced in favour of employing ac coupling in the video section of a TV receiver. Draw the circuit diagram of a typical coupled video amplifier and explain its main features.

4.

5.

6. 7.

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15
Automatic Gain Control and Noise Cancelling Circuits

15
Automatic Gain Control and Noise Cancelling Circuits
Automatic gain control (AGC) circuit varies the gain of a receiver according to the strength of signal picked up by the antenna. The idea is the same as automatic volume control (AVC) in radio receivers. Useful signal strength at the receiver input terminals may vary from 50 µV to 0.1 V or more, depending on the channel being received and distance between the receiver and transmitter. The AGC bias is a dc voltage proportional to the input signal strength. It is obtained by rectifying the video signal as available after the video detector. The AGC bias is used to control the gain of RF and IF stages in the receiver to keep the output at the video detector almost constant despite changes in the input signal to the tuner.

15.1 ADVANTAGES OF AGC
The advantages of AGC are: (a) Intensity and contrast of the picture, once set with manual controls, remain almost constant despite changes in the input signal strength, since the AGC circuit reduces gain of the receiver with increase in input signal strength. (b) Contrast in the reproduced picture does not change much when the receiver is switched from one station to another. (c) Amplitude and cross modulation distortion on strong signals is avoided due to reduction in gain. (d) AGC also permits increase in gain for weak signals. This is achieved by delaying the application of AGC to the RF amplifier until the signal strength exceeds 150 µV or so. Therefore the signal to noise ratio remains large even for distant stations. This reduces snow effect in the reproduced picture. (e) Flutter in the picture due to passing aeroplanes and other fading effects is reduced. (f) Sound signal, being a part of the composite video signal, is also controlled by AGC and thus stays constant at the set level. (g) Separation of sync pulses becomes easy since a constant amplitude video signal becomes available for the sync separator. AGC does not change the gain in a strictly linear fashion with change in signal strength, but overall control is quite good. For example, with an antenna signal of 200 µV the combined RF and IF section gain will be 10,000 to deliver 2 V of video signal at the detector output, 276

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whereas with an input of 2000 µV, the gain instead of falling to 1000 to deliver the same output, might attain a value of 1500 to deliver 3 V at the video detector. Basic AGC Circuit The circuit of Fig. 15.1 illustrates how AGC bias is developed and fed to RF and IF amplifiers. The video signal on rectification develops a unidirectional voltage across RL. This voltage must be filtered since a steady dc voltage is needed for bias. R1 and C1, with a time constant of about 0.2 seconds, constitute the AGC filter. A smaller time constant, will fail to remove low frequency variations in the rectified signal, whereas, too large a time constant will not allow the AGC bias to change fast enough when the receiver is tuned to stations having different signal strengths. In addition, a large time constant will fail to suppress flutter in the picture which occurs on account of unequal signal picked up by the antenna after reflection from the wings of an aeroplane flying nearby. With tubes, a typical AGC filter has 0.1 µF for C1 and 2 M for R1.
IF amplifier RF amp Video signal input

For transistors, typical values are 20 kΩ for R1 and 10 µF for C1. The filtered output voltage across C1 is the source of AGC bias to be distributed by the AGC line. Each stage controlled by AGC has a return path to the AGC line for bias, and thus the voltage on the AGC line varies the bias of the controlled stages.

15.2 GAIN CONTROL OF VT AND FET AMPLIFIERS
The gain of a vacuum tube or FET amplifier can be determined from the equation AV = gmZL, where gm is the transconductance of the device and ZL the impedance of the load. The impedance is determined by the components used in the tuned circuit and does not lend itself to simple manipulation. However, gm of both tubes and field-effect transistors can be controlled by varying their bias. Hence all AGC systems vary the bias of RF and IF stages of the receiver to control their gain. The variation of gm with control grid voltage for a vacuum tube is illustrated in

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Fig. 15.2(a). The transconductance is smaller near cut-off but increases as the bias decreases towards zero. It also decreases if operation of the tube is brought close to saturation. The region near cut-off bias is used for AGC operation. The self-bias is chosen to fix the operating point for high gain. The AGC voltage which is negative gets added to it and shifts the operating point to change the gain. The resulting change in gain compensates for variations in the input signal thereby maintaining almost constant signal amplitude at the output of video detector. Figure 15.2 (b) shows how the control grid is returned to the AGC line for negative bias. In tube circuits a negative bias varying between – 2V and – 20V is developed by the AGC circuit depending on the strength of incoming signal. In order to obtain high gain and minimum cross modulation effects, pentodes with remote cut-off characteristics are preferred in video IF amplifier circuits.
IF amp gm IF transformer

15.3 GAIN CONTROL OF TRANSISTOR AMPLIFIERS
The transistor is a current controlled device. Therefore, in transistor amplifiers it is desirable to consider power gain instead of voltage gain for exploring means of their gain control by AGC techniques. The power gain (G) of a transistor amplifier may be determined by the equation, G≈

β 2 RL Rin

where β and Rin are the short circuit current gain and input resistance of the transistor respectively. Here again it is convenient to vary one of the parameters of the transistor in order to control overall amplification of the receiver. The magnitude of β depends on the operating point of a transistor which is established by the base to emitter (VBE) forward bias. Shifting the operating point, both towards collector current cut-off and collector current saturation causes a decrease in β which in turn reduces the power gain. Figure 15.3 shows the effect of change in VBE on collector current and power gain of an amplifier employing a silicon transistor. A number of conclusions may be drawn from the curves shown in the figure. First,

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the amount of change in VBE which is necessary to shift the operating point of the transistor from cut-off to saturation is small, only 0.4 to 0.5 V. This is much smaller as compared to about 30 V in a vacuum tube. Second, at some optimum value of forward bias (0.7 V in this case) power gain of the amplifier is maximum and does not change much for small variations in the bias voltage. However, the gain decreases as the bias is either increased (shifting the operating point towards saturation) or decreased (moving it towards cut-off).
IC Saturation

Cut-off

0 G Reverse AGC

0.4

0.7

1.0 V Forward AGC

VBE

0

0.4 Towards cut-off

0.7

VBE 1.0 V Towards saturation

Fig. 15.3. Variation of collector current (IC) and power gain (G) of a transistor amplifier as its base to emitter voltage (VBE) is varied.

15.4 TYPES OF AGC
If the operating point is shifted towards saturation for controlling the amplifier gain, it is called forward AGC. An AGC system which operates by shifting the operating point towards cut-off is referred to as reverse AGC. In many TV receiver designs, either forward or reverse AGC is exclusively employed for affecting gain control. However, in some receivers both forward and reverse AGC are simultaneously employed in different parts of the RF and IF amplifier chain. It may be noted that receivers which use either reverse or forward AGC do not operate the amplifiers at peak gain but fix the no-signal operating point at such a value that the stage gain may be increased or decreased without having to move to the other side of the power gain peak. Reverse AGC The power gain curve in Fig. 15.3 is not symmetrical, that is, the reverse AGC region of the curve falls off more rapidly than does the forward AGC region. This means that reverse AGC will require a smaller change in voltage for full gain control than will forward AGC. However,

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operation in this region, which is close to cut-off makes the receiver more susceptible to overload and cross modulation distortion on strong signals. The circuit of Fig. 15.4 (a) is of a single stage transistor (n-p-n) IF amplifier employing reverse AGC. The voltage divider formed by R1 and R2 provides a suitable fixed forward bias from the VCC supply. The resistor R3 and capacitor C1 constitute the AGC decoupling network.

+ From mixer C1 + VCC 1K R3 R1 R2 – R4 0.005 mF C2 + VCC

To 2nd IF amp

– AGC bias

Fig. 15.4 (a). Tuned amplifier with reverse AGC.

Forward AGC Forward AGC is often preferred for controlling gain of video IF amplifiers because it is more linear in its control action. Besides a change in β, the input resistance (Rin) of the transistor also decreases with increase in forward bias. This results in a power mismatch between the tuned IF transformer and the transistor, thereby providing an additional control on power gain.

+ From mixer R4 C1 + VCC R3 + AGC bias R1 R2 C3 – C2

To 2nd IF amp

AC ground R5(large resistance) + VCC

Fig. 15.4 (b). Tuned amplifier with forward AGC.

A single stage tuned IF amplifier employing forward AGC is shown in Fig. 15.4 (b). In this case, for any increase in signal strength the base to emitter forward bias must increase to

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shift the operating point of the transistor towards saturation. Similarly a decrease in signal strength would require a decrease in forward bias. To achieve this, the AGC system must deliver a positive going voltage to the base of amplifier. If a p-n-p transistor is used the forward AGC system would develop a negative going voltage proportional to the signal strength. As shown in the figure, amplifiers employing forward AGC often use a large resistor (R5) in series with the collector circuit. When AGC voltage varies to increase collector current, effective VCE decreases, thereby allowing the transistor to approach saturation quickly for faster AGC action. AGC is applied to the tuner and 1st and 2nd IF stages but not to the third or last IF stage because amplitude of the input signal to the third IF amplifier is quite large and any shift in the chosen optimum operating point by the application of AGC would result in amplitude distortion. Another reason for not applying AGC bias to the last IF stage is the fact that the AGC control is proportional to stage gain and this is more suited to RF and first two IF stages because the RF signal amplitude here is quite small and these stages can be designed for more gain without any appreciable distortion.

15.5 VARIOUS AGC SYSTEMS
An ‘average’ or simple AGC system, as used in radio receivers, is not suited for control of gain in TV receivers. The average value of any video signal depends on brightness of the scene besides signal strength and so is not a true representation of the RF signal picked up at the antenna. For example, a dark scene would develop more AGC bias as compared to a white one, the signal strength remaining the same. This, if used to control the gain of the receiver, would tend to make dark scenes more dark and white ones more bright. With the present system of transmission, the carrier is always brought to the same level when synchronizing pulses are inserted irrespective of the average level of the video signal. The amplitude of the sync level would change only if the signal strength changes. The sync amplitude level, then can serve as the true reference level of the strength of the picked up signal. The system based on sampling the sync tip levels is known as ‘Peak’ AGC system. Either the modulated picture IF carrier signal or the detected video signal can be rectified by the AGC stage to supply AGC bias voltage. In most receivers, however, the AGC circuit uses signal from the video amplifier because the higher signal level allows better control by AGC bias. It is necessary to provide dc coupling between the video amplifier and AGC system in order to keep the pedestals of the video signal aligned. The peak rectifier output then will be a true measure of the signal picked up by the antenna. Peak AGC System A typical peak detector circuit is shown in Fig. 15.5, where a separate diode is used to rectify the signal which is fed to it through capacitor C1 from the output of the last IF amplifier. During positive half cycles of the modulated video signal, diode D1 conducts and the capacitor C1 charges to peak value of the input signal with the polarity marked across the capacitor. During periods other than sync pulse intervals the diode is reverse biased and no current flows through it. However, the capacitor tends to discharge through secondary winding of the IF transformer and R1. Time constant of the discharge path is 270 µs and this being much greater than the line period of 64 µs, the capacitor discharges only partially and regains charge

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corresponding to the sync tip (peak) amplitude on each successive sync pulse. Thus the current that flows through R1 is proportional to the peak value of the modulated video signal and the voltage drop across it becomes the source of AGC bias.

D2 Video detector circuit Lp Ls 270 PF + – R2 AGC voltage C2 C1 D1

Last IF stage

Discharge path of C1

560 K – – 0.22 mF R1 + +

1M

Fig. 15.5. Peak AGC system.

Negative voltage drop across R1 is filtered by R2 and C2 to remove 15625 Hz ripple of the horizontal sync pulses. The output AGC voltage thus obtained is fed to IF and RF stages as an AGC control voltage. Drawbacks of non-keyed AGC. The peak AGC system which is also called the non-keyed AGC system suffers from the following drawbacks, though it measures the same signal strength. (a) The AGC voltage developed across the peak rectifier load tends to increase during vertical sync pulse periods because the video signal amplitude remains almost at the peak value every time the vertical sync pulses occur. This results in a 50 Hz ripple over the negative AGC voltage and reduces gain of the receiver during these intervals. The reduced gain results in weak vertical sync pulse which in turn can put the vertical deflection oscillator out of synchronism causing rolling of the picture. To overcome this drawback a large time constant filter would be desirable to filter out the 50 Hz ripple from the AGC bias. But with too large a time constant the AGC voltage fails to respond to fast changes like aeroplane flutter and quick change of stations. (b) In fringe areas noise pulses develop an additional AGC voltage which tends to reduce the overall gain. This effect is more pronounced for dark scenes. The net effect is that S/N ratio further deteriorates and this results in a lot of snow on the picture. (c) Even when the input signal strength is quite low, a small AGC voltage gets developed and this reduces the gain of the receiver, when actually, maximum possible gain is desired for a satisfactory picture and sound output. To overcome these drawbacks special AGC circuits known as ‘keyed’ or ‘gated’ AGC circuits have been developed and are used in almost all present day television receivers. The problem of reduction of gain with weak input signals is resolved by using ‘delayed’ AGC action.

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Keyed AGC System In this system, the AGC rectifier is allowed to conduct only during horizontal sync pulse periods, with the help of flyback pulses derived from the output of the horizontal deflection circuit of the receiver. Video signal is also coupled to AGC rectifier to produce AGC voltage proportional to signal strength. However, AGC tube or transistor is generally biased to cut-off so that it conducts only for the short time the keying pulse is applied. This ensures that the rectifier conducts only when the blanking and sync pulses are on. As shown in Fig. 15.6 (a) the flyback pulses are generated during the retrace period of horizontal sweep circuit. Thus the time of flyback pulses corresponds to the time of sync and blanking, assuming that the picture is in full synchronization. The gating or AND function means that both inputs must be ‘on’ at the same time to produce AGC output.
On Off Flyback pulses from H.O.T.

Signal from video amplifier

Fig. 15.6 (a). Keying pulses at horizontal sync rate for AGC circuit.

+ Delayed RF AGC 0.22 mF R1

70 V 8.2 M

400 V P–P Keying pulses + – Horizontal output transformer (H.O.T.)

C3 R2

2.2 M A R5 390 K

0.001 mF

C1 AGC winding

R3 IF AGC – 2 V to – 20 V 0.01 mF C2 R4 B

+ 115 V – 25 V + 56 K

180 K R6 From video amp 80 V P–P

Discharge path – of C1¢ 230 K +

50 K R7 330 K 140 V AGC control

Fig. 15.6 (b). Typical triode keyed AGC circuit.

The basic circuit of a keyed AGC system employing a triode is shown in Fig. 15.6 (b). Video signal is directly coupled from video amplifier to the grid of AGC tube. Because of dc coupling the grid is at + 115 volts and so the cathode is maintained at + 140 volts to develop a grid bias voltage equal to – 25 V (115 – 140). Without any video signal on the grid the tube stays beyond cut-off. No dc potential is applied to the plate of the triode, but instead, positive

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going flyback pulses derived from the horizontal deflection output transformer are fed through C1, which drive the plate positive during the sync pulse periods. The video signal of large amplitude drives the grid positive and clamps it at almost zero potential when the signal rises towards blanking level, so that the tube can conduct as the plate is pulsed positive at that time. When plate current flows, it charges C1 with the plate side negative. The path for the charging current includes cathode to plate circuit in the tube, C1 and AGC winding on the horizontal output transformer (H.O.T.). Between pulses when the tube does not conduct, C1 partially discharges through the path marked on the diagram to charge C2, the AGC filter capacitor. The time constant of the filter circuit is much larger than 64 µs and therefore a relatively steady dc voltage is developed to serve as the AGC source. Since the tube conducts only during horizontal sync pulse periods, even when the vertical sync pulse train arrives, AGC voltage developed across the points A and B represents true signal strength and has no tendency to vary during vertical sync periods. The potentiometer R7 is adjusted for optimum grid bias. The function of R6 is to isolate the AGC tube form the video amplifier which supplies video signal. The chain of resistors (R1 through R4) is for delayed AGC action which is explained in a subsequent section of this chapter. Transistor Keyed AGC A basic keyed AGC circuit designed to develop a positive AGC voltage is shown in Fig. 15.7. The p-n-p transistor Q1 is biased to cut-off under no signal conditions. Negative going retrace pulses are applied to the collector circuit. The base-emitter junction gets forward biased when the video signal at the base approaches its minimum value. This corresponds to sync pulse periods and it is then that the collector is pulsed ‘on’ by the flyback pulses. The keying pulses are fed to the collector of the keyer transistor via diode D1. When the transistor conducts, current flow through R1, R2, Q1, D1, winding on H.O.T. and C1 thereby charging it with the polarity marked on it. The voltage developed across C1 is the AGC output voltage. Diode D1 prevents discharge of C1 during the time between keying pulses. In the absence of D the charge on C1 will forward bias the collector to base junction of Q1 and allow the capacitor to discharge resulting in loss of AGC voltage. However, the diode has the correct polarity to couple negative flyback pulses from AGC winding to the collector of the transistor. The capacitor C1 and resistors R3 and R4 form a filter to develop a steady dc voltage for controlling overall gain of the receiver.
Flyback pulses AGC voltage + 1.5 V to + 4 V R4 25 V P–P R3 5.6 K 6.8 K 10 mF Winding on D 1 4.7 K + H.O.T. Q1 C1 – + 10 V 3.2 V + 2.8 V to 3.7 V to 3.3 V 4K 3K Pot R 2 AGC level 370 W R1 From video amp 2 V P–P

Fig. 15.7. Typical transistor keyed AGC circuit.

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It may be noted that in comparison to vacuum tube circuits, the peak-to-peak amplitude of the keying pulses and video signal is much smaller. Typical transistor circuit values are 25 V peak-to-peak for flyback pulses and 2, V p-p for the video signal. For an n-p-n transistor the polarity of both the inputs is opposite to that needed for a p-n-p transitor.

15.6 MERITS OF KEYED AGC SYSTEM
(a) A long time constant to filter out 50 Hz ripple is no longer necessary because conduction takes place only during the horizontal retrace periods and no undue build up of voltage occurs during vertical sync intervals. The relatively short time constant filter, used to remove 15625 Hz ripple, enables the AGC bias to respond to flutter and fast change of stations, thereby ensuring a steady picture and sound output. (b) AGC voltage developed is a true representation of the peak of fixed sync level and thus corresponds to the actual incoming signal strength. (c) Noise effects are minimized because conduction is restricted to a small fraction of the total line period.

15.7 DELAYED AGC
The picture produced on the raster should be as noise free as possible. This is achieved by lownoise circuits. However, despite careful circuit design, each stage in the receiver contributes some noise. The cumulative effect of this, if not checked, would be a noisy picture. Noise effect can be overcome by high amplification of the incoming RF signal. The amplified signal will then swamp out effects of stage noise as it is processed by the receiver. It is, therefore, necessary to operate the RF amplifier at maximum gain, particularly for weak RF signals. In the circuits discussed so far, the AGC voltage is fed not only to the first and second IF amplifiers, but also to the RF amplifier. Thus the negative AGC bias would reduce the gain of RF amplifier even for low-level RF signals. This undesirable effect is overcome by delaying the AGC voltage to the RF amplifier for weak RF signals. The technique used which is a delay in voltage, not in time, is called delayed AGC. Delayed AGC Circuit In Fig. 15.6 (b) delay action is achieved through the voltage divider R1, R2, R3 and R4 tied between B + supply and ground. This places the RF AGC take-off point (grid of the RF amplifier) at approximately + 70 V to ground. Since the cathode of the RF amplifier tube returns to ground, the grid to cathode circuit acts like a diode and is forced into conduction. Since a conducting diode may be considered to be a short, the AGC take-off point is clamped to approximately zero volt because the cathode of the amplifier returns to ground. The bias clamp will continue to conduct until the input signal becomes sufficiently strong to provide a negative AGC voltage which is large enough to overcome the forward bias applied to the bias clamp diode and turn it off. This will restore normal AGC action to the RF amplifier for input signal amplitudes higher than a predetermined level. It can be set by varying the potentiometer R7. Another method of obtaining delayed AGC is to use a separate diode for clamping action. This method is explained later while discussing a typical tube AGC circuit.

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15.8 NOISE CANCELLATION
The need for minimizing noise set-up in AGC and sync separator circuits has given rise to several methods of noise cancellation. Three commonly used methods are described. In Fig. 15.8 (a) diode D2 is used as a switch which opens in the presence of noise preventing it from reaching the video amplifier. The necessary forward bias enabling it to pass noise free video signals is set by the potentiometer R3. When a strong noise signal arrives the diode gets reverse biased thereby stopping noise pulses from reaching the video amplifier. Since the video signal to both AGC keyer and sync separator is obtained from the output of video amplifier, noise pulses are prevented from reaching these circuits.
Noise pulse Bias D1 L1 0 D2 Video detector From last video IF amp R1 C1 L2 Noise gate R4 To keyed AGC circuit 0 Video amp To picture tube circuit

To sync separator

+

R2

R3 Noise gate level

Fig. 15.8 (a). Diode noise gate circuit.

In the noise cancellation circuit of Fig. 15.8(b) the signal inputs to the control grid (G1) suppressor grid (G3) and plate (P) of a pentode are applied in such a way that the tube conducts only if all the three inputs are present at the same time. This arrangement is often referred to as a coincidence gate or an AND gate. Noise cancellation is achieved by setting the control grid bias in such a way that it goes to cut-off when a strong noise pulse arrives. Special tubes with sharp cut-off characteristics have been developed for use in TV receivers and 6HA7 is one such tube—a twin pentode, where one section is for AGC and the other for sync separation. As shown in Fig. 15.8 (b) the cathode, control grid and screen grid are common to both the sections. The control grid serves as the noise gate for both the circuits. However, there are two suppressor grids and two plates to separate sync output from one pentode and AGC voltage from the other. Another noise cancellation circuit is shown in Fig. 15.8 (c), where noise is eliminated by using a separate noise gate. It separates noise from the composite video signal, amplifies it and then adds it to the inverted composite video signal. The noise gate is a grounded base amplifier, normally set to cut-off. Any incoming noise pulse of sufficient amplitude counteracts the fixed reverse bias and sets the amplifier into conduction. Thus the noise pulse is amplified without inversion of its polarity. The gain of the noise gate amplifier is set equal to that of the

15.9 TYPICAL AGC CIRCUITS
As already explained there is a considerable difference between the methods of controlling gain of vacuum tube and transistor amplifiers. Transistor keyed AGC circuits are complex than their tube counterparts. In some circuits three or more transistors are used to develop the required AGC voltage. There is no standard approach to such circuits and they may vary from chassis to chassis. Some typical AGC circuits employing tubes and transistors are discussed. Keyed AGC with a Twin-Triode The keyed AGC circuit shown in Fig. 15.9 employs a twin-triode for generating AGC voltage. One section (V1) of the tube is used for developing a dc voltage proportional to the peak value

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of the input video signal. The second triode (V2) is connected for keyed AGC action. The video signal is dc coupled to the grid of V1 through a frequency compensating network. It is connected as a cathode follower with R3 as its load resistance. The time constant of R3 in parallel with C3 is chosen to be quite large as compared to the line period of 64 µs. Therefore, the voltage which develops across this network is a dc voltage proportional to the peak value of sync pulses. This is direct coupled to the grid of V2 which conducts during flyback pulse intervals to charge the capacitor C2. In between sync pulses C2 discharges to develop a negative voltage at point A with respect to ground. The potentiometer R5 is varied to adjust optimum bias voltage for V2 to permit sufficient conduction for developing suitable AGC voltage with a known input signal strength. Note that any strong noise pulse will make G2 positive with respect to K2 causing clamping action. Thus noise pulses are prevented from developing any AGC voltage. The resistors R7 and R8 form a potential divider and the voltage which develops across the filter circuit R8–C5 is fed to the IF section of the receiver.
Flyback pulses From video amp 400 V P–P

The values of resistors R7 through R10 are so chosen that at low RF signal levels D1 remains forward biased from B + supply. Thus point B of the AGC circuit is clamped to ground and no AGC voltage gets applied to the RF amplifier. Any change in the negative potential at A affects the potential at point B through the isolating resistor R9. When the input signal strength increases, say when another strong channel is selected, point A becomes more negative with the result that at a predetermined voltage level the potential at B changes to become negative. This reverse biases diode D1 removing its clamping action. Thus full AGC voltage becomes available on the RF amplifier AGC bias line. This then amounts to a delay in applying AGC to the tuner section of the receiver.

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AGC Circuit for Solid-State Receivers A typical circuit is shown in Fig. 15.10 where transistor Q1 is keyed to develop AGC bias and Q2 serves as the AGC amplifier. The video signal is dc coupled at the base of Q1 and its emitter is fed with a suitable dc voltage to reverse bias the emitter-base junction in the absence of any video signal. The flyback pulses are fed at the collector of this transistor through the diode. The pulsed current completes its path back to the winding on the H.O.T. through C1 which then charges with positive polarity towards ground. R5-C1 acts as the AGC filter and this voltage is dc coupled to the base of Q2. The forward-bias on the emitter-base junction of this transistor varies with the voltage across C1 and thus controls the collector current through load resistance R10. A strong input signal at the antenna will develop more negative voltage across C1, thereby, reducing the forward bais on Q2. This in turn will decrease its collector current making the collector more positive. The receiver employs forward AGC control on the RF and IF amplifiers and the voltage developed at the collector is then of the right polarity to decrease gain of these stages. R7-C3 constitute an additional filter circuit for the RF bias line. R8 is the isolating resistance and R9-C4 acts as filter for IF bias line. Another transistor can be added to this circuit to achieve delay action for the RF amplifier.
Flyback pulses + VCC 8V P–P From video amp Driver + R11 33 K R2 C2 R3 8V P–P + 27 V Winding on H.O.T. Keyer Q1 (BC 147) 2.5 mF 330 K Pot – C + 1 R10 47 K R4 1.5 K Q2 Amplifier (BC 147) R8 R 3.3 K R7 22 K 680 RF amp bias 2 to 5 V + C 25mF – 3 IF amp bias 4 to 12 V

200 V R1 1K 047 mF

6 R5 4.7 K 120

R9 C

4

+ 25mF –

Fig. 15.10. Keyed automatic gain control circuit with AGC amplifier.

Improved AGC Circuit An improved AGC circuit is shown in Fig. 15.11. The bias for the two IF amplifiers is developed by Q2, the AGC ‘keyer’. For the RF amplifier, bias is supplied by Q3, the AGC driver. It will be seen that there are different modes of AGC operation, depending on the level of incoming signal. On very low signals, no AGC bias is developed, and the controlled IF and RF stages are fixed biased. On medium-level signals the AGC keyer turns on and biases the two IF amplifiers. As the signal level increases further the AGC driver is also turned on, biasing the RF amplifier, while the bias on the controlled IF amplifiers continues to increase. A point is reached beyond which the bias of the IF amplifiers is clamped. When the incoming signal level becomes very high, both RF bias and IF bias again increase to reduce the gain of the controlled amplifiers. The circuit operates in the following manner. The negative going (sync) composite video signal from the video detector is fed to the video driver Q1 (p-n-p) which is connected as an

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emitter follower and develops at its emitter a negative-going composite video signal, superimposed on a + 3.8 V dc level. This is dc coupled to the base of Q2 through R15, an isolating resistor. The AGC keyer Q2 is normally at cut-off. This is so because its emitter-base junction is reverse biased and no dc voltage is applied at its collector. The transistor Q3 is also normally gated to cut-off. Therefore, with no or on very low-level RF signals, both Q2 and Q3 remain in cut-off.

Q4 1st video IF R14 R20 C5 C8 R11

Q5

2nd video IF

91

R6 C7

C6 + 3.5 V Negative keying pulses 5.6 K R10

470

R7 4.2 V

D4 R12 R13 1.5 K + VCC 12 V

From video detector

3V

Video driver Q1 3.8 V L1 R15

D1

R18 AGC Q2 keyer 150 W 1.5 V R19 R3

Winding on H.O.T. D2 82 W R2 560 + 3.5 V 100 R1 680 4.5 V D3 + – 4V

1.5 K Delay Adj

0.5 V VDR1

AGC Q3 driver

R17

C4 50 mF

R16

+ 12 V Delayed AGC to tuner R9

R4 + 1.5 V C3 R5

+ VCC 12 V

C1 4 mF

R8

4.7 K

680

C2 3 mF

Decoupling network

Fig. 15.11. Improved AGC circuit. Note that the voltages shown are with no signal applied.

The series dc voltage divider consisting of R1 through R5 develops about + 3.5 V at the junction of R3 and R4 from the 12 V dc supply. This potential is coupled via diode D2 to the base of transistor Q5 in the second IF amplifier. Bias for the base of Q4 (first video IF amplifier) is derived from the junction of R6 and R7 located in the emitter lead of Q5. The operating points of both Q4 and Q5 are chosen for forward AGC control. From the same VCC source another circuit is completed through D3, VDR1 and R8, thereby providing about 4 V across C1 and 0.5 V at the base of Q3. The voltage drop across R5, the

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emitter resistance of Q3 is about 1.5 V on account of current which flows through it from the VCC source. This is enough to keep Q3 in cut-off. The voltage drop across R5 (1.5 V) is fed as bias voltage to the RF amplifier through the decoupling network R9-C3. Thus with very low input signal levels the IF and RF amplifiers operate at a fixed bias chosen to provide maximum gain. When the incoming RF signal level increases sufficiently the negative sync pulse amplitude at the emitter of Q1 overcomes the reverse-bias on the emitter-base junction of Q2. As a result, the negative-going horizontal keying pulses injected in the collector circuit of Q2 from the H.O.T. cause collector current flow during sync pulse intervals. This results in a higher voltage across C1 and thus D2 is reverse biased. The increased potential on C1 is now coupled through R10 and R11 to the base of Q5. The enhanced forward bias on Q5 while increasing its collector current takes it to a region of reduced power gain. Similarly the increased voltage drop across R7 is enough to shift the operating point of Q4 to reduce its gain. Thus the gain of both, 1st and 2nd video IF amplifiers is reduced. Note that the increased positive AGC voltage appearing across C1 is prevented from reaching the emitter of Q2 by diode D3 and will therefore not effect the current flow in this transistor. Similarly diode D1 which couples keying pulses to the collector of Q2 prevents application of positive AGC voltage in its collector circuit. For the signal levels just considered the AGC driver transistor Q3 remains cut-off and no additional AGC bias for the RF amplifier is developed. When the incoming RF signal level increases sufficiently, the large positive AGC voltage developed across C1 decreases the resistance of the voltage dependent resistance VDR1, with the result that a higher dc voltage is applied to the base of Q3. This turns on Q3 and the resultant increased positive voltage across its emitter resistance (R5) furnishes additional RF AGC bias. Potentiometer R12 sets the collector voltage level of Q3 thus determining the condition for conduction of this transistor. R12 therefore acts as the tuner AGC delay control. With still higher signal levels, IF AGC voltage will increase unitl it becomes 0.5 V more positive than the voltage across R12. D4 then conducts clamping the IF AGC voltage at that level. Any added increase in the received RF signal will now cause the RF amplifier AGC voltage to increase rapidly. Finally on very strong signals, when the tuner AGC voltage exceed 7 V, Q3 conducts heavily causing the voltage drop across R5 to rise rapidly. The additional voltage drop across R5 raises the potential at point A to such a level the diode D2 again gets forward biased. Thus the IF AGC will again increase causing further reduction of gain in the IF section.

15.10 AGC ADJUSTMENTS
Some receivers do not have any AGC adjustments. Other receivers have as many as three adjustments that affect the operation of the AGC system. In case a noise cancellation control forms part of the AGC circuit, it should be first advanced till the noise cancellation circuit becomes inoperative. This will ensure that no sync inversion occurs while other adjustments are made. After all other AGC adjustments have been made the noise threshold control should be advanced to the point of most stable sync. Many receivers have a tuner AGC delay control. This control should be adjusted on medium-strength signals. As the tuner delay is increased the signal will become noise free.

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With too much delay, strong signals will cause overloading of amplifiers. Indications of overload are buzz in the sound and bending of the picture. The third type of control is an AGC level control, sometimes called the ‘threshold control’. This adjustment sets the voltage to which the sync tips must rise to give AGC action. The effect of this control is to adjust the detector level. In the absence of service notes the output from the detector must be estimated. This normally lies between 2 to 5 volts. However, manufacturers’ instructions, if available, should be followed for all adjustments. AGC Circuit in an IC Chip Weak signal sections of the receiver which commonly use integrated circuits include video IF, video detector, AGC, AFC, video preamplifier and sound strip. The various functions performed by one such IC (TBA 890) were discussed in the previous chapter. The BEL CA3068 is another IC which has complete video IF subsystem and tuner AGC for monochrome and colour receivers. This integrated circuit consists of nearly 39 transistors, 10 diodes, 67 resistors and 18 capacitors. Figure 15.12 is a simplified block diagram of CA3068. Note that the tuned circuit filters and decoupling resistors and capacitors are connected externally at the corresponding pins of the integrated package.
Tuned filter 9 13 AFT drive 14 2nd and 3rd video IF amp 19 Video output 3 Keying pulses AGC noise gate

The IF output from the tuner is fed at pin 6 to the first video IF amplifier. Horizontal flyback pulses are needed for AGC action. The dynamic range of the IF AGC is 55 db. Delayed AGC for the tuner is available at pin number 7, with a AGC voltage variation from 2.2 V to 4.5 V. Besides noise immunity circuits, the IC employs a zener diode as an RF bias clamp to prevent application of excessive AGC bias. The chip also has a provision for ‘service switch’ which can be used to isolate the AGC stage for fault finding.

Review Questions
1. 2. 3. What are the advantages of using AGC in television receivers ? Describe the basic principle of automatic gain control and show how it is applied to tube and transistor amplifiers. What is meant by ‘forward AGC’ and ‘reverse AGC’ in transistor tuned amplifier circuits ? Why is forward bias control preferred to reverse bias method of gain control ?

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4. 5.

What is the basic principle of peak AGC system ? Explain how the control voltage is developed and applied to IF and RF amplifier stages of the receiver. What is delayed AGC and how is it developed ? Why is delayed AGC applied only to the RF amplifier and somtimes to the first IF amplifier of the receiver ? Why is AGC not applied to the last IF amplifier ? Describe with a simple circuit the basic principle of a ‘keyed AGC’ system. How does it overcome the shortcomings of a ‘non keyed’ (peak AGC) control system ? A keyed AGC system employing tubes is shown in Fig. 15.9. Explain how the AGC voltage is developed and applied to RF section of the receiver. What is the function of diode D1 ? Draw circuit diagram of a keyed AGC system employing transistors and having a noise gate. Explain how the AGC voltage is developed and amplified. Explain briefly various types of noise gates used to suppress noise pulses in the video signal before it is applied to the AGC and sync separator circuits.

6. 7. 8. 9.

10. The circuit of an improved keyed AGC system is shown in Fig. 15.11. It is designed to operate in different modes depending on the level of incoming RF signal. Explain step by step how the control voltage is developed and applied to RF and IF sections of the receiver.

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16
Sync Separation Circuits

16
Sync Separation Circuits
The synchronising pulses generally called ‘sync’ are part of the composite video signal as the top 25 percent of the signal amplitude. The sync pulses include horizontal, vertical and equalizing pulses. There are separated from the video signal by the sync separator. The clipped line (horizontal) and field (vertical) pulses are processed by appropriate line-pulse and fieldpulse circuitry. The sync output thus obtained is fed to the horizontal and vertical deflection oscillators to time the scanning frequencies. As a result, picture information is in correct position on the raster. The sequence of operations is illustrated in Fig. 16.1 by a block schematic diagram.

Fig. 16.1. Block diagram of the sync separator and deflection circuits in a television receiver.

16.1 SYNC SEPARATOR—BASIC PRINCIPLE
The problem of taking off the sync pulses from the video waveform is a comparatively simple one, since the action consists of merely biasing the device used in the circuit, in such a way, that only the top portions of the video signal cause current flow in the device. This is readily achieved by self-biasing the tube or transistor used in the circuit. Two basic circuits, one employing a tube and the other a transistor are shown in Fig. 16.2 to illustrate this method of sync separation. Self-biasing or automatic bias means that the dc 296

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297

bias voltage is produced by the ac signal itself. The requirements are to charge the input capacitor by rectifying the input signal while it approaches its maximum value and have an RC time-constant long enough to maintain the bias on the capacitor between peaks of the ac input signal. The video signal is normally obtained from the video amplifier and coupled to the input of the sync-separator circuit. In Fig. 16.2 (a) video signal is fed to the grid of the triode with sync pulses as the most positive part of the waveform. In the quiescent state there is no
+ VPP RL v0 Video input R1 + + VCC RL C2 – R2

bias on the tube. On the arrival of the signal, first few pulses drive a heavy grid current and the capacitor C1 quickly charges up with the grid side negative. Between peaks of the input signal, C1 discharges slightly through R1. The R1, C1 time constant is made large enough to keep C1 charged to about 90 percent of the peak positive input. The effect is to develop an automatic negative bias so that the operating point sweeps back form VGK = 0 to a point well beyond cut-off. After a few pulses the tube settles down into a steady bias such that it is completely cut-off except during the positive going sync pulses. The tube then operates under ‘class-C’ condition i.e., biased well beyond cut-off. This is illustrated in Fig. 16.2 (c). To maintain a steady bias, the sync tip levels of the video signal make the grid slightly positive every cycle

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to cause a small grid current flow, which quickly replenishes the charge lost by the capacitor through the grid leak resistance R1. The average negative bias developed across the capacitor C1 then controls the plate current during sync pulse periods, which in turn gives rise to corresponding negative-going voltage pulses at the plate. Between pulses, the plate voltage rises to + VPP since there is no voltage drop across the plate load resistor. The corresponding transistor circuit is shown in Fig. 16.2 (b) where C2, R2 coupling provides self-bias between the base and emitter of the transistor. The capacitor charges because of the base current that flows during sync amplitude levels of the composite video signal. The circuit operation is illustrated in Fig. 16.2 (d) where the negative voltage developed across C2 reverse biases the emitter-base junction in such a way that only positive sync voltage drives the transistor into conduction to produce sync output in the collector circuit. The sync amplitude varies between VCC and collector voltage corresponding to the maximum collector current. The bias voltage values in transistor circuits are less than in tube circuits because the base-emitter junction requires only a fraction of a volt as forward bias, to produce collector current output. The RC time constant is the same (about 0.1 second) as in tube circuits but with large C and small R because of the lower input resistance of a transistor.

16.2 SYNC SEPARATOR EMPLOYING A PENTODE
The triodes have relatively large interelectrode capacitances and therefore their performance in sync-separator circuits is inferior to that of pentodes. Accordingly a pentode is preferred to a triode in such circuits. A typical sync separator employing a pentode is shown in Fig. 16.3.
B+
80 V P–P

The components and DC source voltage are so chosen that both plate and screen-grid voltages are low. This limits the dynamic region of the tube to a narrow range, with a cut-off bias of the order of about 5 volts. With the input voltage on, the plate voltage attains a minimum value during sync tip intervals. The actual minimum plate voltage obtained, and hence the sync peak-to-peak amplitude, depends on the value of plate load resistance and peak plate current.

SYNC SEPARATION CIRCUITS

299

It should be noted (see Fig. 16.2 (c)) that the effect of driving the tube into grid current on each sync pulse tip is to clamp the sync pulse tops to zero (ground) potential, so that the sync tips align at the same level despite wide variations in the inter-pulse periods. This in effect amounts to restoration of dc component of the video signal, which is lost due to the presence of series coupling capacitor C1. One may ask as to why the video signal is not dc coupled from video amplifier to syncseparator. The reason is that changes in dc conditions, both because of variations of the average brightness of the scene and signal amplitude, when stations are changed, are bound to take place over a period of time and would make it very difficult to arrange for a constantly efficient sync separator action. The ac coupling, with its own automatic and flexible dc restoration function, provides the signal in the form, where it becomes easy to slice-off sync pulses of equal amplitude. The resistance R3 provides isolation between the sync separator and video amplifier circuits. Because of saturation occurring at the sync tip level of the video signal, any noise resting on the sync tips does not get reproduced in the output.

16.3 TRANSISTOR SYNC SEPARATOR
A sync separator employing an n-p-n transistor is shown in Fig. 16.4. The capacitor C1 tends to charge up to the peak input signal voltage less the base-emitter forward voltage drop. There is a marked difference in the signal voltage amplitude necessary to drive tube and transistor sync separators, since a transistor requires only about 0.6 V or so at the base to produce collector current output. An input video signal of the order of 5 to 10 V is all that is necessary to give rise to output sync pulses whose amplitude approaches about 50 volts peak-to-peak.

As shown in Fig. 16.4 a transistor sync separator often employs a fairly high value of collector load resistance with the result that the transistor bottoms (saturates) at a very low value of base (input) current. Full amplitude sync pulses are thus developed for a wide range of input signal strength levels. In general, a higher value of load resistance reduces the peak drive current needed to bottom the transistor but leads to broadening of the output pulses. This is due to charge storage effect, where the charge carriers stored in the base region take longer to dispel, if the collector load resistance is increased. This necessitates the use of

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transistors having fast switching capability which implies that the transistor must have a high upper cut-off frequency. In sync separators which employ switching transistors, a small forward bias is sometimes given to ensure that tips of the pulses drive the transistor well into the bottoming condition to produce good clean output pulses. The factors which must be kept in view while designing a transistor sync separator can be summarized as follows: (i) To achieve a reasonable voltage gain the β of the transistor should be large. (ii) A transistor with a small output leakage current must be chosen because the leakage current lowers the collector voltage and thus reduces net amplitude of the sync output. (iii) To ensure a steep front edge of sync pulses a high frequency transistor is desirable. (iv) Since the transistor is off most of the time a low power transistor can be employed.

16.4 NOISE IN SYNC PULSES
Noise pulses are produced by ignition interference from automobiles, arcing brushes in motors, and by atmospheric noise. The noise is either radiated to the receiver or coupled through the
Noise pulse Video input

power line. Especially with weak signals the noise can act as false synchronizing pulses. Furthermore, when noise pulses have much higher amplitude than the sync voltage, large grid/base current flows which charges the coupling capacitor to a much higher voltage than is normal and this results in a noise set-up. Because of the long time constant of self-biasing network, the sync separator is held much beyond cut-off for a period which depends on the amplitude and width of the noise pulse and the time constant of the input circuit. As shown in Fig. 16.5 (a) the sync separator gets blocked and there is weak or no sync output till the bias returns to its normal value. During strong noise periods, the picture does not hold still until synchronization is restored again. Thus in order to reduce the effect of noise, sync circuits generally employ one or more of the following techniques : (i) Double Time-constant for Signal Bias The time constant of the grid/base leak-bias circuit, at the input of sync separator, must be long enough, to maintain bias from line to line and through the time of vertical sync pulses in order to maintain a constant clipping level. As stated earlier, a time constant of the order of 0.1 second is adequate for this purpose. But such large values would result in long blocking on strong noise pulses. In a similar way too long a time-constant will tend to increase the negative bias then its normal value during vertical sync intervals when the composite video signal voltage stays close to its peak value. This results in shortening of horizontal signal pulses soon after the vertical pulse train during each field. However, if the time constant is made too short to overcome the above drawbacks, this may not maintain bias between sync pulses, specially during the vertical sync pulse time. The result may be inadequate sync separation during and immediately after the vertical sync pulses. Therefore the problem of reducing the effect of high frequency noise pulses without changing the time constant of the average bias network is solved by providing a double time constant circuit at the input of the sync separator. The circuit configuration is shown in Fig. 16.5 (b). It may be noted that the two sync separator circuits described earlier (Fig. 16.3 and Fig. 16.4) also have double time constant circuits at their inputs. The network R1,C1 provides the normal grid/base leak-bias, with a time constant of 0.1 second for the sync signal. The small capacitance C2 (200 pF) and resistance R2 (270 KΩ) provide a short time-constant. The double time constant thus provided enables the negative bias to change quickly to reduce the effect of noise pulses in the input to the sync separator. The capacitor C2 being small can quickly charge when noise pulses produce grid/base current thus increasing the bias for noise. The change in voltage across C1 and C2 is inversely proportional to their capacitance values. Therefore a noise pulse will charge C2 to a voltage 500 times more than that across C1. Since R2C2 time constant is 50 µs, C2 can discharge through R2 between sync pulses. Thus the bias is maintained at the normal value for sync pulses.

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Another advantage which accrues by the addition of a short time-constant circuit is that it acts like a frequency response compensator and maintains the sharp rise time of input sync pulses. (ii) Sync Clipper after Sync Separator The sync output limited by saturation does not result in sharp sync pulses. The clipping level is also not uniform because of shift in self-bias caused by sharp noise pulses. Many sync-separator circuits have a sync clipper stage where sync output of the separator is clipped and amplified. The purpose is to provide sharp sync-pulses with equal and high amplitude, free from noise pulses and without any camera signal. Clipping in successive stages allows the top and bottom of sync pulses to remain sharp and prevents noise pulses from having higher amplitude than the sync. (iii) Use of Noise Cancellation Circuits The use of a double time constant circuit as a noise suppressor is only suitable for non-recurring noise. If noise pulses are periodic, the small charge on C1 (see Fig. 16.5 (b)) contributed by each noise pulse will be cumulative and noise set-up will still occur. Therefore, in many sync separator circuits some form of noise suppression switch is provided. Several such noise cancellation circuits which were described in the previous chapter along with AGC circuits are also used in sync separator circuits.

16.5 TYPICAL TUBE SYNC SEPARATOR CIRCUIT
The separator circuit shown in Fig. 16.6 employs a pentode as a sync-separator followed by a triode which provides gain and ensures sharp edged sync output. The video signal is coupled to
B+

the separator through a cathode follower from the video amplifier. A double time-constant circuit is provided at the input to reduce the effect of noise pulses on clipping level. The triode also operates on self-bias developed through network R3, C3, which couples the pentode to the triode. Since the amplitude of the sync output at the plate of the pentode is almost steady, the time-constant R3, C3 has been chosen to be quite small (2.5 ms). This is adequate, both for a steady auto-bias generation and suppression of occasional excessive noise pulse amplitudes.

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The load of the triode consists of a small inductor L1 and an 8.2 KΩ resistor in series. Sync pulses to the vertical oscillator are fed from plate of the triode whereas the input of the horizontal automatic frequency contol (AFC) circuit is connected to the junction of L1 and resistance R7. The sharp peaked pulses that develop across the inductor when triode conducts are of the desired shape and amplitude for feeding to the AFC circuit. The coil L1 is shunted by a resistance R8 to suppress onset of self oscillations when shock excited by sharp plate current pulses. In some AFC circuits, discussed in the next chapter, a balanced sync pulse output is necessary. This can be easily obtained by inserting a suitable resistor in the cathode lead of the triode. The two outputs, one at the plate and the other at the cathode of the triode are then 180° out of phase with respect to each other.

16.6 TRANSISTOR NOISE GATE SYNC SEPARATOR
A transistorized sync-separator employing a noise gate is shown in Fig. 16.7. The noise gate transistor Q2 is in series with the emitter of the sync separator Q1. The tansistor Q2 is so biased that it normally stays in saturation. The sync separator then operates normally and its emitter current completes its path through Q2, which has only about 0.6 V between its collector and emitter when saturated. Any sharp noise pulse in the video signal gets coupled to the base of Q2 through diode D2 and this cuts-off the noise gate transistor Q2. In the absence of any collector current through Q2, the emitter of Q1 rises to + 20 volts and this blocks the transistor Q1. Hence no sync separation occurs during the noise pulse interval. It may be noted that normal amplitude of the negative video signal fed at the input of Q2 is not sufficient to turn if off. However, if a noise spike with a negative amplitude beyond the sync tip appears, it turns the noise gate off. However, noise pulses which are not longer than sync tip will not cause the noise gate to turn off.
N 470 K + 340 V 180 K 4.7 K From video amplifier circuit Q1 0.068 mF 0.0068 mF + 20 V 27 K 120 K N From video detector output D2 Q2 27 K 0.05 mF Noise gate transistor D1 39 K 100 K 0.015 mF Vertical sync output 0.056 mF Horz sync output

0.0056 mF

Fig. 16.7. Transistor sync separator with a noise gate.

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16.7 IMPROVED NOISE GATE SYNC SEPARATOR
The circuit of Fig. 16.8 is another example of a typical sync clipper cum noise invertor. Q1, an n-p-n transistor, acts as the sync separator. In the absence of any signal, the base-to-emitter voltage is zero and the transistor is off. When a positive-going signal appears at its input,the transistor is turned on. As shown along the circuit a 4 V (p-p) positive composite video signal is applied at the sync input terminals through the isolating resistor R6. This turns Q1 on and base current flows drawing the vertical sync pulses through C1, D1, emitter-base junction and D2. The resulting current charges the capacitor C1 with the polarity marked across it. Since the sync pulse is the highest component of the composite video signal, the voltage developed across C1 at the end of the vertical sync pulse (which is 160 µs wide) is approximately 4 V. When the vertical sync pulse passes, the base of Q1 is held negative with respect to the emitter and it is cut off again. Capacitor C1 tends to discharge through R1 during the interval of 18.84 ms between vertical sync pulses but the time-constant C1R1 is so large (220 ms) that C1 can discharge only about 8 percent of the voltage across it. When the next vertical sync pulse arrives, its peak is approximately 8 percent more positive than the change on C1. Thus Q1 is again turned on during the sync pulse interval and this part of the input signal gets amplified to develop a 20 V p-p negative going sync pulse at the collector.
22 V + VCC 1K input D3 5.6 K R7 5.6 K R9 15 K D4 + C4 R4 6.8 M Q1 – Sync separator Q2 – Noise inverter R8 5.6 K R6 0.1 mF Q2 C5 R5 C6 R1 1M + C1 100 K R2 D2 0.0056 mF C2 R3 3K To horz AFC circuit Q1 C3 0.01 mF To vertical oscillator

4V P–P

D1

0.22 mF 220 mF

0.47 mF

Fig. 16.8. Improved transistor noise gate sync separator.

After the vertical pulse interval, the horizontal sync pulses which are 4.7 µs wide, arrive. Because the charge on C1 is still high to keep Q1 in cut-off, the horizontal sync pulses are provided another path through C2 to turn it on. Base current again flows during the horizontal sync pulse interval, charging capacitor C2. In the 59 µs interval between consecutive horizontal sync pulses, C2 discharges through R2 (R2C2 = 560 µs) by about 12 percent only and thus keeps Q1 in cut-off state. When the next horizontal sync pulse arrives, it is sufficiently positive to turn Q1 ‘on’ again. The negative horizontal sync pulses developed at the collector of Q1 are again of about 20 V peak-to-peak. Thus there are two separate input paths for the vertical and horizontal sync pulses, ensuring that both are amplified by Q1, while eliminating the video information between sync pulses.

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The fact that the positive horizontal sync pulse is coupled directly to the base of Q1, back biases diode D1 during the horizontal sync pulse interval. This prevents the base current of the transistor from discharging C1. Diode D2 prevents transistor conduction in the reverse direction when the negativegoing signal adds to the charge on the base input capacitors to exceed the base-emitter voltage rating. Noise Invertor. As shown in Fig. 16.8 the noise invertor transistor Q2 together with its associated circuitry is in parallel with the input to the sync separator. This offsets the effects of high amplitude noise pulses in the following manner. The diode D4 rectifies the normal incoming positive going composite video signal and acts as a peak detector on account of the long time constant of R4C4(R4C4 = 3.2 sec). The resulting dc voltage across C4 is equal to peak of the sync pulses. This positive voltage gets applied to the cathode of D3 through resistors R8 and R7. In turn this reverse biases D3 under normal signal conditions and no signal appears through D3 at the base of Q2. It stays in cut-off and does not affect the normal operation of the sync separator. However, when a sudden noise spike appears as shown in the figure, voltage across C4 cannot change quickly, and the diode D3 will couple the spike through C5 to the base of Q2. The transistor Q1 then conducts heavily and effectively shorts its collector circuit to ground. This results in shorting of the entire input signal to the sync separator, including video information and sync pulses, during time of the noise spike. As soon as the noise pulse disappears, however, Q2 cuts off again and the next video signal and sync pulses will be uneffected. Capacitor C6 prevents any dc short across Q2.

16.8 SYNC AMPLIFIER
An amplifier is not always used before a sync separator. It will depend on the type of transistor used as a separator and the polarity of the sync pulses at the output of the video detector. If the video detector output is taken from its anode and its amplitude is sufficiently high, it can be fed directly to a p-n-p type sync separator. This eliminates the need for a sync amplifier before the separator. If an n-p-n type transistor is employed, as in Fig. 16.8, it is necessary to reverse the polarity of a negative-going video signal at the output of the detector before it can be coupled to the separator. In that case the output of a common emitter video amplifier may be used as input to the sync separator. IC Sync Circuit In some receivers sync-separator function is performed in an integrated circuit. This IC is a part of the sync-AGC module that combines noise inversion, sync separation and AGC. Noise cancellation and amplification are first performed in the IC. The resultant noise free video signal emerges at a particular pin of the IC and is applied to a time-constant network that is kept outside the IC. This enables the use of different time-constant network to suit a particular receiver design. The output from the network is fed back to the IC where sync separation takes place. Over 20 volts of separated sync is available, both positive and negative going at different pins. The IC also receives gating pulses from the horizontal output circuit for AGC operation.

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Review Questions
1. Draw the basic circuit diagram of a sync separator employing a triode with grid leak bias. Comment on choice of the time-constant of the biasing circuit. Why is a pentode preferred to a triode in tube sync-separator circuits ? Draw the circuit of a sync separator employing a p-n-p transistor. Show input and output waveforms. Why are high-frequency-transistors with small output leakage current employed in sync circuits ? What are the undesirable effects of high amplitude noise pulses in sync-separator circuits ? Explain how the use of a double time-constant biasing circuit overcomes the effect of noise in input signal to a sync separator. Why is a sync clipper often employed after the sync separator ? Explain the operation of sync separator shown in Fig. 16.6. What is the use of inductor in plate circuit of the triode ? Draw the circuit diagram of a typical transistor noise gate sync separator and explain its operation. The circuit of an improved noise gate sync separator is shown in Fig. 16.8. Describe its operation and in particular explain how the noise invertor transistor Q2. cancels the effect of noise pulses.

2.

3.

4. 5. 6.

17
Sync Processing and AFC Circuits

17
Sync Processing and AFC Circuits
The receiver has two separate scanning circuits, one to deflect the electron beam, of the picture tube, in the vertical direction and the other in the horizontal direction. Each scanning circuit consists of a waveform generator, i.e. oscillator and a power output stage. The synchronizing pulses obtained from the sync-pulse separator are used to control the vertical and horizontal deflection oscillators, so that picture tube is scanned in synchronism with the original picture source at the transmitter. The horizontal sync pulses hold the line structure of the picture together by locking in the frequency of the horizontal oscillator; and the vertical sync pulses hold the picture frames locked-in vertically by triggering the vertical oscillator. The equalizing pulses help the vertical synchronization to be the same in even and odd fields for good interlacing.

17.1 SYNC WAVEFORM SEPARATION
This means separating the vertical and horizontal sync pulses. It is the difference in the pulse time duration of the horizontal (line) and vertical (field) sync pulses which makes it possible to separate them. The horizontal sync pulse with a width of 4.7 µs and repeated at 15625 Hz represents a high frequency signal, whereas the vertical sync pulse with a total width of 160 µs which repeats 50 times in a second, is relatively a very low frequency signal. Therefore, the vertical and horizontal sync pulses can be separated from each other by RC filters. A low-pass filter, connected across the incoming sync pulse train, will develop appropriate trigger pulses
Q1 – Emitter follower Q2 – Sync separator – VCC O H – VCC 0.047 mF 100 K Q2 2K 1K C2 HPF t = 0.5 mS R2 R1 H EE V sync pulses EE Clipping level X Y t

To vertical osc vo1 C1

2K From video amplifier

Q1

33 W

5 mF

LPF t = 50 to 60 mS

vo2 To AFC circuit

Spiked horz sync pulses

Fig. 17.1. Separation of vertical and horizontal sync pulses.

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for synchronizing the vertical oscillator. Similarly, a high-pass filter will deliver sharp differentiated pulses for the horizontal oscillator from the same pulse train. This, as shown in Fig. 17.1, is done simultaneously by feeding output from the sync pulse separator to the lowpass and high-pass filter configurations, connected in parallel. The resistor R1 and capacitor C1 constitute the low-pass filter, also known as integrating circuit. The high-pass filter, also known as differentiating circuit, consists of C2 and R2 and has a very small time constant. The integrated output across C1 that builds up 50 times during one second is used for triggering the vertical oscillator. However, the spiked (differentiated) output that develops across R2 is fed to the automatic frequency control (AFC) circuit, the output of which is employed for holding the horizontal oscillator at the correct frequency. The use of the AFC circuit ensures correct synchronization even in the presence of noise pulses. The serrated vertical sync pulses also develop only a spiked output across R2, because the time constant of the circuit R2C2 is relatively too small to produce any appreciable voltage across R2. The sync separator shown in Fig. 17.1 is preceded by an emitter follower to isolate it from the video amplifier. The small amount of forward bias at the base of Q2 ensures good bottoming and thus clean output sync pulses are fed to the filter circuits. The output waveshapes from the integrating and differentiating circuits are shown alongside corresponding filter configurations.

17.2 VERTICAL SYNC SEPARATION
The time constant of the low-pass filter circuit (see Fig. 17.1) is chosen to be much larger than the width of each serrated vertical pulse. This value is not very critical and a time constant R1C1 of about ten times the serrated pulse width is adequate. When the combined sync waveform, beginning with horizontal and equalizing pulses appears at the input of such a circuit, the capacitor C1 charges along an exponential curve governed by the time constant of the filter circuit. Since this time constant is very large compared with the duration of the horizontal and equalizing pulses, the voltage output across C1 during these intervals of the input wave is a very small fraction of the ultimate value. This is shown in the output waveform of the integrator. When the trailing edges of the horizontal or equalizing pulses appear, the small charge stored on the capacitor discharges along an exponential curve governed by the same time constant. The overall result is a very small toothed voltage across the capacitor, which lasts for a time comparable to the width of each horizontal or equalizing pulse, and thus the amplitude of the filter output voltage is negligible during the horizontal and equalizing sync periods. However, when the vertical sync pulse (serrated) arrives, cumulative charging of the capacitor occurs, because the duration of each serrated vertical sync pulse is long compared with the gaps between serrations. Consequently,the charge accumulated from the first input serration (29.7 µs) has little opportunity to discharge during the following notch (2.3 µs). The next broad pulse adds to the charge already built-up across the capacitor which again is only partially discharged during the next gap. The five broad pulses which constitute the vertical sync pulse thus cause a gradual increase in voltage across the capacitor, with small spikes superimposed on it. At the output of the low-pass filter, therefore, the voltage rise appears to be almost smooth corresponding to the vertical pulse. This pulse amplitude is substantially

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greater than the small spikes caused by the horizontal and equalizing pulses. As soon as the vertical sync pulse has passed the integrated output pulse decays almost to zero during the post-equalizing pulse period, and stays at this level during the horizontal pulse train that follows. The purpose and effectiveness of the equalizing pulses is apparent from the plot shown in the figure. The inclusion of equalizing pulses, before and after the field pulses, ensures identical integrated output despite insertion of field waveform in the middle of one line for one field, and at the end of the line on alternate fields. In many vertical sync control circuits the integrated field pulse waveform is clipped by a reverse biased diode, along the line XY as shown in Fig. 17.1,and thus the voltage changes across C1 due to line pulses are not seen by the field oscillator circuit at all. Cascaded Integrator Sections A very large time constant for the integrating circuit removes horizontal sync pulses, and also reduces the vertical sync amplitude across the integrating capacitor. This rising edge of the sync pulse is then not sharp and this can lead to incorrect triggering of the oscillator.However, when the R1C1 time constant is chosen to be relatively small the horizontal sync pulses cannot be filtered out and serrations in the vertical pulse produce notches in the integrated output. Thus though the vertical sync pulse rise is quite sharp, the output voltage attains the same amplitude at different charging excursions. This can also lead to wrong synchronization and so the notches must be filtered out. The resulting outputs with large and small time constant are illustrated in Fig. 17.2 (a).
Input pulse v Insufficient integration (RC too short) Serration

To overcome the above described discrepancy, most receivers employ a two-section integrating circuit with each RC section having a time constant of about 50 µs. Such a circuit is illustrated in Fig. 17.2 (b). The operation of the circuit can be considered as though the R1C1 section provides integrated voltage across C1 that is applied to the next integrating section R2C2. The overall time constant for both sections together is large enough to filter out horizontal sync pulses while the shorter time constant of each section allows the integrated voltage to rise more sharply because each integration is performed with a time constant of 50 µs. In some designs even a three section integrator is provided and such a configuration is illustrated in Fig. 17.2 (c).

17.3 HORIZONTAL SYNC SEPARATION
The high pass filter circuit and the differentiated output are shown in Fig. 17.1 along with the sync separator circuit. The time constant of this circuit (R2C2) is kept much smaller (normally 1/10th) than the width of the horizontal pulse. A time constant between 0.5 µs to 1 µs is often employed. The physical action of the differentiator is as follows. When a leading edge of the incoming pulse train is applied to the C2,R2 circuit, the initial voltage across the capacitor C2 is zero, and so full amplitude of the leading edge appears across the resistor R2 and the output wave follows almost exactly the shape of the input leading edge. When the flat top of the input rectangular wave is reached, on further charging of the capacitor occurs, and the circuit discharges along an exponential curve governed by the timeconstant of the circuit. Since this time-constant is very short compared to the duration of input pulse, the discharge completes itself before the trailing edge of the pulse arrives. When the trailing edge of the input pulse occurs, it produces another pulsed output (see Fig. 17.1) of opposite polarity to that of the first pulse. Since the trailing edge component extends in opposite direction to the leading edge output, it has no effect on the triggering of the horizontal oscillator. The equalizing and vertical sync pulses (notched) produce two leading edge components when they occur during each line scanning interval of 64 µs. The extra leading-edge pulses occur when the horizontal oscillator is insensitive to sync pulses and hence have no effect on its frequency. As was explained in Chapter 3, the leading edge of each horizontal sync pulse, alternate equalizing pulses, and alternate serrations of the vertical sync pulse are correctly timed to initiate the horizontal retrace periods. Moreover as explained above, the differentiating circuit is insensitive to the flat top portions of the rectangular waves, that is, the amplitude of

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the differentiated output is independent of the duration of the input pulse. There is, therefore, no particular response to the vertical sync pulses, except at their edges and they as such have no effect on the triggering of the horizontal oscillator.

17.4 AUTOMATIC FREQUENCY CONTROL
The direct use of the incoming sync pulses to control vertical and horizontal sweep oscillators, though simple and most economical is normally unsuitable, because of their susceptibility to noise disturbance arising from electrical apparatus and equipment operating in the vicinity of the television receiver. The noise pulses extending in the same direction as the desired sync pulses cause greatest damage when they arrive during interval between the sync pulses. Noise pulses which are most troublesome, possess high amplitude, are of very short duration and tend to trigger the oscillator prior to its proper time. When the vertical oscillator is so triggered the picture moves vertically, either upwards or downwards, until proper sync pulses in the signal again assume control. If the horizontal oscillator is incorrectly triggered a series of lines in a narrow band will be jumbled up, giving the appearance of streaking or tearing across the picture. The horizontal sweep system of the receiver is effected more by noise pulses than the vertical system. To understand this fully, it is necessary to examine the nature of interfering voltages. The energy of the noise pulses is distributed over a wide range of frequencies. For a peak to occur, the phase relationship amongst various frequencies must be such as to permit their addition to form a high amplitude pulse. This condition, however, usually exists only for a brief interval which explains the small width of these pulses. The high frequency noise pulses, when they reach the vertical sync separator, i.e. lowpass filter, get suppressed along with the line sync and equalizing pulses because of large time constant of the circuit. The presence of this low-pass filter (the integrating network) is mainly responsible for greater immunity to noise pulses enjoyed by the vertical system. This explains why no special circuit is used between the sync separator and the vertical oscillator. However, when a wide noise pulse is received, it contains enough energy to cause off-time firing of the vertical oscillator, but the annoyance caused to the viewer on account of occasional rolling of the picture is seldom great. On the other hand, circuit leading to the horizontal oscillator, being a high-pass filter, passes noise pulses readily along with the narrow horizontal sync pulses. This results in serious interference with the normal functioning of the horizontal sweep oscillator which, in turn, results in frequent ‘tearing’ of the picture. In order to ensure that the horizontal oscillator operates at the correct frequency, and is basically immune to noise pulses, all horizontal deflection oscillators are controlled by some form of a circuit known as the automatic frequency control circuit (AFC circuit). The AFC circuit receives sync pulses and output from the horizontal oscillator simultaneously and compares them regarding their phase and frequency. The discriminator in the AFC circuit develops a slowly varying voltage, the magnitude of which depends on deviation of frequency of the horizontal oscillator from its correct frequency. In case the oscillator frequency is equal to the incoming horizontal sync frequency, i.e. 15625 Hz, no output voltage is developed.

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The AFC circuit output is filtered by a low-pass filter to obtain an almost dc voltage, which then controls the frequency of the horizontal sweep oscillator. Thus the use of AFC circuit and a low-pass filter at its output, eliminates the effect of sharp noise pulses. The use of this indirect method of frequency control results in excellent horizontal oscillator stability and immunity from noise interference. AFC Operation The block schematic arrangement of a frequently used AFC circuit for the horizontal deflection oscillator is illustrated in Fig. 17.3. Horizontal sync voltage and a fraction of the horizontal deflection voltage, normally taken from the horizontal output circuit, and suitably processed to form horizontal flyback pulses, are coupled in the sync discriminator. The discriminator consists of two diodes and associated circuitry. It detects the difference in frequency and develops a dc output voltage proportional to the difference in frequency between the two input voltages. The dc control voltage indicates whether the oscillator is ‘on’ or ‘off ’ the sync frequency. The greater the difference between the correct sync frequency and the oscillator frequency, larger is the dc control voltage. This dc control voltage is fed to a large time constant filter, the output of which is used to control the oscillator frequency. The shunt by-pass capacitor of this lowpass filter eliminates the effect of noise pulses. A large time constant filter could not be used directly, in the horizontal system ; because while suppressing noise pulses it would have prevented the desired horizontal sync pulses from reaching the horizontal sweep oscillator. This explains the need and use of the AFC circuit.

The automatic frequency control circuit is generally called flywheel sync, sync lock, stabilized sync or horizontal AFC, based on the technique employed to develop the control voltage. Most present day receivers use either a push-pull or a single-ended phase discriminator AFC circuit for sensing any error in the horizontal oscillator frequency.

17.5 AFC CIRCUIT EMPLOYING PUSH-PULL DISCRIMINATOR
A typical circuit arrangement employing push-pull phase discriminator is shown in Fig. 17.4. The sync pulses of equal amplitude but of opposite polarity are obtained from the phase splitter circuit and coupled to the diodes D1 and D2, through capacitors C1 and C2 respectively. R1 and R2 are of equal value and act as load resistors to the two diodes. The diodes are so connected that the application of sync pulses of opposite polarity forward biases them simultaneously.

It is convenient to study the functioning of this circuit under three different conditions: (a) when sawtooth feedback voltage is only present, (b) when only sync pulses are present and (c) when both sawtooth and sync pulses are simultaneously present. (a) When only sawtooth voltage is applied (no sync pulses) D1 conducts during the negative and D2 during the positive half of the sawtooth cycle, charging C1 and C2 with polarities shown. When voltage peaks have passed C1 discharges via R3, B+, B–, (ground) R8, R7 and R1. Similarly C2 discharges through R2, R7, R8, B– and R4. This raises the lower end of R2 (marked V2) above ground potential and the upper end of R1 (marked V1) below ground potential. Equilibrium is reached when | V1 | = | V2 | =
1 2

E2 (peak-to-peak) of the applied sawtooth voltage. Thus VB

equals zero and the upper end of R7 stays at ground potential. When put in another way this means that the two equal and opposite currents flowing through R8 develop a net zero voltage across it. In steady state, small pulses of make-up current flow at the positive and negative peaks of the sawtooth through D2,C2 and C1,D1 respectively. (b) When only sync signal is present, the positive going sync pulses are coupled from the collector of the sync splitter through C1 to the anode of D1. At the same time negative going sync pulses are coupled via C2 to the cathode of D2. As a result of these pulses, current flows along the path C1, D1, D2, C2, R4, B–, B+, and R3 thereby charging the capacitors C1 and C2 to approximately peak value of the sync pulses with polarities shown across them. During the time between sync pulses, the capacitor C1 discharges through R3, B+, B–, R8, R7 and R1. At the same time the capacitor C2 discharges via R2, R7, R8 and R4. Two equal but opposite voltages, caused by equal capacitor discharge currents develop across R8. These voltages cancel each other leaving a net zero voltage across R7 and R8. Thus VB equals zero and this point continues

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to be at ground potential. Note that V1 and V2 are of such polarity that both the diodes are reverse biased during the time between sync pulses. (c) Now if sync and sawtooth voltages are applied simultaneously, as they would be under normal conditions, three cases are possible—(i) pulses in synchronism, i.e. oscillator frequency is correct, (ii) oscillator frequency is more than the sync frequency, and (iii) oscillator frequency is less than the sync frequency. (i) For this case as shown in Fig. 17.5 (a), sync pulses and sawtooth voltage arrive symmetrically, i.e. the sync pulses occur when the sawtooth is passing through its zero point during retrace. As a result the circuit behaves as though each signal were applied to the phase detector independent of each other. Thus the sync pulses deposit equal charges on C1 and C2, causing smaller but identical make-up currents to be drawn from the sawtooth source. The magnitudes of V1 and V2 remain equal (VB = 0) and no control voltage is developed across R8. Therefore, the frequency of the horizontal deflection oscillator remains unchanged. (ii) When the oscillator frequency is high (see Fig. 17.5 (b)) the sync pulses arrive late in a relative sense, i.e. when the sawtooth is already in its positive half cycle. Thus the sawtooth forward biases D2 and reverse biases D1 with the result that D2 conducts harder than D1. Thus more charge is added to C2 and less to C1. During the discharge periods of these capacitors unequal currents flow through R8 and a positive error voltage is developed across it. The discharge currents also cause V2 to increase and V1 to decrease from their quiescent values making VB positive. The low-pass network R9, C5 filters the control voltage and feeds a dc error voltage to the horizontal oscillator. This forces the oscillator frequency to return to its normal value.
Sync pulses Sync frequency = 15625 Hz E1 0 VAK(D1) = VAK(D2) Sync pulses

(iii) As shown in Fig. 17.5 (c) when the oscillator frequency is low, sync pulses occur during the time the sawtooth is in its negative half cycle forward biasing D1 more than D2. Analogous arguments establish that for this case a negative control voltage is developed across R8. This, after filtering is applied to the horizontal oscillator causing its frequency to increase to its normal value. Thus the sync discriminator continuously measures the frequency difference between the sawtooth and sync pulses to produce a dc correction voltage that locks the horizontal oscillator at the synchronizing frequency. In Fig. 17.4, R10 and C6 constitute an anti-hunt, circuit whose function is described in a later section of the chapter.

Circuit Operation Assume that only sync pulses are applied to the phase detector. During the time T1 (see sync pulse waveform) the sync separator is cut-off and its collector voltage is equal to B+ voltage. As a result the coupling capacitor C1 is fully charged. When the negative sync pulse (T2) arrives, i.e. Q1 saturates, the two diodes are simultaneously forward biased. Note that the diodes are effectively in parallel for the sync input due to the large (820 pF) capacitor C5 to ground from the anode of D1. This enables C1 to discharge through two different paths. One path for discharge is through R3, Q1, B– and D2 while the other is completed via R3, Q1, B–, C4, C3 and D1. In this

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process C3 is charged with positive polarity towards C4. Because of very short time constant during the discharge period, C1 discharges almost completely. During the following long period (T3) between sync pulses both diodes are turned off and C1 charges through two independent paths. One charging path is through B+, R4, R3, C1, and R2. The second path is completed via B+, R4, R3, C1, R1, C3 and C4. The capacitors C3 and C4 being practically short at the sync frequency, two equal and opposite voltage drops (V1 and V2) develop across R1 and R2 respectively producing a net zero voltage at point X (across the two diodes) with respect to ground. In addition C3 discharges and attains its earlier status. The voltage drops V1 and V2 provided reverse bias across D1 and D2 respectively and thus permit conduction only during peak values of the applied signals. It may be noted that a small (62 pF) capacitor C2 is connected across D2. This provides frequency compensation and ensures that the sawtooth voltage-drops, across D1 and D2 are identical in magnitude and waveshape. However, despite this precaution to correct mismatch between the components that constitute the phase detector, in practice a small voltage of the order of a few tenths of a volt does exist between point X and ground. When only sawtooth voltage is applied at X, practically the entire voltage appears across D1 and D2 in series opposition and forces them into conduction alternately. When the sawtooth is positive to ground, D1 is turned on and the current which flows through it and R2 charges C3. During the negative half cycle, D1 is turned off, but D2 conducts through R1, C3 and C4 thereby discharging C3. Once again voltage drops V1 and V2 are equal and opposite and zero net voltage is developed at point ‘X’. When sync pulses and sawtooth voltage are simultaneously applied, as is necessary for normal operation of any AFC circuit, three possible conditions can occur. These are illustrated in Fig. 17.6 (b) with sync waveform ‘A’ as the reference.

A

0

Reference horizontal sync

B

0

Trace

Retrace f osc = f sync Control voltage is zero

C

0

f osc > f sync Control voltage is positive

D

0

f osc < f sync Control voltage is negative

Fig. 17.6 (b). Three possible conditions of horizontal sweep relative to sync pulse in a single ended AFC circuit.

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First (see waveform B), the sync pulses may occur when sweep (sawtooth) voltage is passing through zero. It is as if the sync pulses were applied without the sawtooth and vice versa. Consequently, no dc error voltage is produced. In a practical circuit, however, as stated earlier, a small voltage appears across the two diodes. Second, the frequency of the oscillator may be higher than the sync pulse frequency. When this happens (see waveform C in Fig. 17.6 (b)) horizontal sync will occur during negative half cycle of the feedback sawtooth voltage. In this event D2 conducts harder than D1, leaving a net potential across C3 with positive on its plate that is tied to the anode of D1. Subsequently when C3 discharges via D1 and R2, V2 becomes greater than V1. Thus a net positive voltage is developed across R1 and R2. This control voltage at X when fed after filtering to the horizontal oscillator forces its frequency to decrease. In the third case the frequency of the oscillator may be lower than the sync pulse rate. Under this condition (waveform D) sync pulses will occur during positive half cycle of the feedback sweep voltage with the result that D1 will conduct harder than D2. Consequently C3 will now attain a net negative charge on its plate tied to D1. On discharging through D2,R1, the capacitor C3 now develops a higher voltage across R1 than R2. Thus the net voltage ( | V1 | – | V2 |) across point ‘X’ and ground will be negative. Such a control voltage forces the oscillator to increase its frequency returning it to its normal operation.

17.7 PHASE DISCRIMINATOR (AFC) WITH PUSH-PULL SAWTOOTH
A sync discriminator in which flyback pulses of opposite polarity are coupled to the two diodes is illustrated in Fig. 17.7 (a). The flyback pulses of large amplitude are obtained from the two ends of a centre tapped winding, wound on the horizontal output transformer. These pulses are coupled at the points, marked A and B on the circuit diagram, where they are integrated by two RC networks (RA, CA and RB, CB) to develop sawtooth outputs of opposite polarity. The amplitude of the sawtooth voltage around its centre-zero axis is of the order of 20 volts.

The anodes of diodes D1 and D2 are tied together and positive going sync pulses from the sync separator and invertor circuit are applied at this junction. The dc control voltage that develops across the diode load resistors R1 and R2 depends on deviation of the horizontal oscillator frequency from that of the incoming sync pulses. The waveforms in Fig. 17.7 (b) show how the net voltage across each diode varies in accordance with variations of the oscillator frequency. As explained in the previous (single ended) circuit it is the charging and discharging action of the sync coupling capacitor C1, through two different paths that results in the development of a net zero positive or negative control voltage. The control voltage is coupled to the grid of a reactance tube, normally the triode portion of a pentode-triode tube. Network C2,R3 is provided to compensate for any imbalance in the ac voltage applied to the two diodes. The pentode section together with its triode acts as a sinusoidal oscillator cum waveshaper. The reactance tube acts as a variable capacitor/inductor across the tank circuit of the oscillator to keep the frequency of oscillations at 15625 Hz. It may be noted that in any sync discriminator circuit it is possible by reversing the polarity of the sawtooth voltage, which is fed to the discriminator, to obtain control voltage of opposite polarity for the conditions of a fast or a slow oscillator. The dc control voltage, as obtained from the various discriminator circuits, varies between ± 2 and ± 6 volts.

17.8 DC CONTROL VOLTAGE
The time constant of the RC filter provided at the output of the discriminator, determines how fast the dc control voltage can change its amplitude to correct the oscillator frequency. A time constant much larger than 64 µs is needed for the shunt capacitor to bypass horizontal sync and sawtooth components in the control circuit while filtering out noise pulses. However, a large time constant may not permit the control voltage to change within a fraction of a second when sync is temporarily lost while changing channels. Also, if the time-constant is too large,

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the dc control voltage may be effected by the vertical sync pulses, causing bend at the top of the picture. A typical value of the AFC filter time-constant is about 0.005 second, i.e., a period nearly equal to 75 horizontal lines. Hunting in AFC Circuits The filtering circuit that follows the diode section of the discriminator controls the performance of the AFC circuit. Too large a time constant makes the control sluggish, while insufficient damping, on account of too small a time constant, causes the oscillator to ‘hunt’ returning to the correct frequency. Excessive hunting in the AFC circuit appears as ‘weaving’ or ‘geartooth’ on the picture. The manner in which the oscillator frequency deviates from the correct value on account of hunting is illustrated in Fig. 17.8 (a). In order to prevent this a double section filter is often used for the dc control voltage. In this network a shown in Fig. 17.8 (b), the R1C1 time constant of 0.005 sec is large enough to filter out noise, horizontal sync and flyback pulse effects. The
Oscillator hunting due to difference in timing between error voltage and oscillator frequency +
Error voltage Error voltage

second section, R2 and C2 in series, is known as the ‘anti-hunt network’. The relatively low resistance of R2 serves as a damping resistance across C1 making the output voltage more resistive and less capacitive, thereby reducing time delay (see Fig. 17.8 (a)) in the change of control voltage.

Review Questions
1. Draw basic low-pass (integrating) and high-pass (differentiating) filter configurations, which are employed to separate vertical and horizontal sync information. Comment on the choice of time constants of these circuits. Sketch accurately, output voltage waveforms of the filter circuits, when fed with a pulse train separated from the incoming composite video signal. Why is a cascaded network preferred for developing vertical sync pulses ? Why is it not necessary to employ an AFC circuit for developing control voltage for the vertical deflection oscillator ? Draw a basic (block schematic) AFC circuit and explain how the control voltage is developed. Explain fully how the effect of noise pulses is leiminated. Draw a typical push-pull sync discriminator (AFC) circuit and explain with the help of neatly draw waveforms, how a control voltage, proportionate to the deviation of horizontal oscillator frequency is developed. Why is a single-ended AFC discriminator preferred to the push-pull circuit ? Draw its circuit diagram and explain with the help of necessary waveforms, how the control voltage develops, when the oscillator frequency is (i) correct, (ii) fast and (iii) slow. Why is an anti-hunt circuit used while filtering the error voltage obtained from any AFC discriminator ? Draw its circuit configuration and explain how hunting is suppressed.

2. 3. 4.

5.

6.

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18
Deflection Oscillators

18
Deflection Oscillators
In order to produce a picture on the screen of a TV receiver that is in synchronism with the one scanned at the transmitting end, it is necessary to first produce a synchronized raster. The video signal that is fed to the picture tube then automatically generates a copy of the transmitted picture on the raster. While actual movement of the electron beam in a picture tube is controlled by magnetic fields produced by the vertical and horizontal deflection coils, proper vertical and horizontal driving voltages must first be produced by synchronized oscillators and associated waveshaping circuits. As illustrated in Fig. 18.1, for vertical deflection the frequency is 50 Hz, while for horizontal deflection it is 15625 Hz. The driving waveforms thus generated are applied to power amplifiers which provide sufficient current to the deflecting coils to produce a full raster on the screen of picture tube. Free running relaxation type of oscillators are preferred as deflection voltage sources because these are most suited for generating the desired output waveform and can be easily locked into synchronism with the incoming sync pulses.

It may be noted that complementary pair circuits are possible only with transistors while all other types may employ tubes or transistors. As explained earlier, both vertical and horizontal deflection oscillators must lock with corresponding incoming sync pulses directly or indirectly to produce a stable television picture.

18.1 DEFLECTION CURRENT WAVEFORMS
Figure 18.2 illustrates the required nature of current in deflection coils. As shown there it has a linear rise in amplitude which will deflect the beam at uniform speed without squeezing or spreading the picture information. At the end of ramp the current amplitude drops sharply for a fast retrace or flyback. Zero amplitude on the sawtooth waveform corresponds to the beam at centre of the screen. The peak-to-peak amplitude of the sawtooth wave determines the amount of deflection from the centre. The electron beam is at extreme left (or right) of the raster when the horizontal deflecting sawtooth wave has its positive (or negative) peak. Similarly the beam is at top and bottom for peak amplitudes of the vertical deflection sawtooth wave. The sawtooth waveforms can be positive or negative going, depending on the direction of windings on the yoke for deflecting the beam from left to right and top to bottom. In both cases (Fig. 18.2) the trace includes linear rise from start at point 1 to the end at point 2, which is the start of retrace finishing at point 3 for a complete sawtooth cycle.
+I Trace 0 3 (a) +I 1 Trace 3 I(P–P) t –I 2 (b) Retrace One cycle 2 Retrace t

Driving Voltage Waveform The current which flows into the horizontal and vertical deflecting coils must have a sawtooth waveform to obtain linear deflection of the beam during trace periods. However, because of inductive nature of the deflecting coils, a modified sawtooth voltage must be applied across the coils to achieve a sawtooth current through them. To understand this fully, consider the equivalent circuit of a deflecting coil (Fig. 18.3) consisting of a resistance R in series with a

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pure inductance L, where R includes the effect of driving source (internal) resistance. The voltage drops across R and L for a sawtooth current, when added together would give the voltage waveform that must be applied across the coil. The voltage drop across R (Fig. 18.3 (a)) has the same sawtooth waveform as that of the current that flows through it. However, the voltage across L depends on the rate of change of current vL = L

FG H

di dt

IJ K

and the magnitude of

inductance. A faster change in iL, produces more self induced voltage vL. Furthermore, for a constant rate of change in iL, the value of vL is constant. As a result, vL in Fig. 18.3 (b) is at a
i 1 vR v vL Deflection coil L R 0 Coil current vR t

0 (a) vL 0

t

t

(b) v

0

t Trapezoid (c)

Fig. 18.3. Current and voltage waveshapes in a deflection coil (a) voltage drop across the resistive component of coil impedance (b) voltage drop across the inductive component of coil impedance (c) resultant voltage ‘v’ (VR + vL) across input terminals of the coil for a sawtooth current in the winding.

relatively low level during trace time, but because of fast drop in iL during the retrace period, a sharp voltage peak or spike appears across the coil. The polarity of the flyback pulse is opposite to the trace voltage, because iL is then decreasing instead of increasing. Therefore, a sawtooth current in L produces a rectangular voltage. This means, that to produce a sawtooth current in an inductor, a rectangular voltage should be applied across it. When the voltage drops across R and L are added together, the result (see Fig. 18.3 (c)) is a trapezoidal waveform. Thus to produce a sawtooth current in a circuit having R and L in series, which in the case

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under consideration represents a deflection coil, a trapezoidal voltage must be applied across it. Note that for a negative going sawtooth current, the resulting trapezoid will naturally have an inverted polarity as illustrated in Fig. 18.4.
i 0 t

Trace v

Retrace

0

t

Trace

Retrace

Fig. 18.4. Inverted polarity of i and v.

As explained above, for linear deflection, a trapezoidal voltage wave is necessary across the vertical deflecting coils. However, the resulting voltage waveform for the horizontal yoke will look closer to a rectangular waveshape, because voltage across the inductor overrides significantly the voltage across the resistance on account of higher rate of rise and fall of coil current. Effect of Driving Source Impedance on Waveshapes In deflection circuits employing vacuum tubes, the magnitude of R is quite large because of high plate resistance of the tube. Therefore, voltage waveshape across the vertical deflection coils and that needed to drive the vertical output stage is essentially trapezoidal. However, in a horizontal output circuit employing a tube, the waveshape will be close to rectangular because of very high scanning frequency. When transistors are employed in vertical and horizontal deflection circuits, the driving impedance is very low and equivalent yoke circuits appear to be mainly inductive. This needs an almost rectangular voltage waveshape across the yoke. To produce such a voltage waveshape, the driving voltage necessary for horizontal and vertical scanning circuits would then be nearly rectangular. Thus the driving voltage waveforms to be generated by the deflection oscillator circuits would vary depending on deflection frequency, device employed and deflection coil impedance.

18.2 GENERATION OF DRIVING VOLTAGE WAVEFORMS
Sawtooth voltage is usually obtained as the voltage output across a capacitor that is charged slowly employing a large time constant to generate the trace period and then quickly discharged through a short time constant circuit to obtain the retrace period. The initial exponential rise

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of voltage across the capacitor is linear, and thus alternate charging and discharging of the capacitor at the rate of deflection frequency results in a sawtooth output voltage across it. This is illustrated in Fig. 18.5(a) where CS is allowed to charge through a large resistance R1 from a dc (B+) source. The charging time is controlled by switch ‘S’ which is kept open during the trace period at the deflection frequency. At the end of trace, switch ‘S’ is closed for a time equal to the retrace period, and the capacitor discharges quickly through a small resistance. R2. Actually, switch ‘S’ represents a vacuum tube or a transistor that can be switched ‘on’ or ‘off ’ at the desired rate. When the tube or transistor is in cut-off state, it corresponds to ‘off ’ position of the switch. In the ‘on’ state during which the active device is allowed to go into saturation, the tube or transistor conducts heavily allowing the capacitor to discharge through its very low internal resistance which corresponds to resistance R2 shown in series with the switch. In this application the tube or transistor is called a ‘discharge device’ and capacitor CS is often referred to as ‘sawtooth capacitor’ or ‘sweep capacitor’. As mentioned earlier the trace voltage should rise linearly. For this, only linear part of the exponential volt-time characteristic is used. To achieve this, the time constant (RC) of the circuit should at least be thrice the trace period. Same result can also be achieved by employing a higher B+ voltage. The waveshapes shown in Fig. 18.5(b) illustrate the effect of charging time constant (RC) and source voltage B+ on linearity and magnitude of the sawtooth voltage. Trapezoidal Voltage Generation As explained earlier it is often necessary to modify the sawtooth voltage to some form of a trapezoidal voltage before feeding it to the output stages for obtaining linear deflection. Figure 18.6 shows a basic circuit for generating such a voltage. It is the same circuit discussed earlier but employs a transistor as discharge switch and has a small resistance RP (peaking resistance) in series with the sawtooth capacitor CS. The transistor which is biased to cut-off by battery VBB is driven into saturation by the incoming, large but narrow positive going pulses. It thus acts as a discharge switch to produce a fast retrace. During long intervals in-between positive pulses, CS charges towards B+ through the large resistance R1 to provide trace voltage. Since the value of RP is small as compared to R1, voltage developed across it is quite small while CS charges. However, on arrival of a positive pulse, Q1 goes into full conduction thus providing a very low resistance path (small RC) for the capacitor to discharge. The high discharge current which also flows through RP develops a large negative voltage pulse across it. This is illustrated by the waveform drawn along RP in Fig. 18.6. As shown by another waveform, the spiked voltage across RP adds to the sawtooth voltage across CS to produce a trapezoidal voltage v0 between point ‘A’ and ground. Note that exact charge and discharge periods must be in accordance with the synchronized vertical and horizontal scanning rates. This function is assigned to the vertical and horizontal oscillators.
R1 S DC supply Charging path + CS Discharge path R2 v0

Fig. 18.5 (a). Basic circuit for generating a sawtooth voltage.

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329
High B+ Low B+ Same time constant

v0

Linear trace

Retrace Nonlinear trace

0 Long time constant circuit Short time constant circuit Time

Fig. 18.5 (b). Effect of charging time constant (RC) and source voltage (B+) on linearity and magnitude of v0.
B+ R1 v0 A

Vin RB – +

Q1

CS

VC

S

VR RP

P

VBB

V0

Fig. 18.6. Generation of trapezoidal sweep voltage.

18.3 BLOCKING OSCILLATOR AND SWEEP CIRCUITS
This oscillator may be thought of as a tuned-plate (or collector) configuration that is disigned to produce an extreme case of intermittent oscillations. The feedback transformer is polarized to produce such a large amount of feedback that the cumulative action is almost instantaneous. In circuits employing tubes, the grid current that flows on account of regeneration develops such a large negative self-bias that the tube is immediately driven to much beyond cut-off. This prevents the circuit from generating continuous sinusoidal oscillations, at the natural resonant frequency of the feedback transformer, depending on its inductance and stray capacitance. Thus only a single short pulse of large amplitude is generated. The cycle is repeated when the self-bias returns to the conduction region. The number of times per second the oscillator produces pulse and then blocks itself, is the pulse repetition rate or oscillator frequency.

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Similarly in a transistor blocking oscillator, feedback action switches the transistor between saturation and cut-off at a rate that is controlled by choice of time constant in the input circuit of the oscillator. This repetition rate or oscillator frequency is chosen to be 50 Hz and 15625 Hz for the vertical and horizontal deflections respectively. Vacuum Tube Blocking Oscillator and Sweep Generator The switch S in Fig. 18.5 (a) can be replaced by a blocking oscillator to control the charge and discharge periods of the capacitor. Such a circuit is illustrated in Fig. 18.7(a) where tube V1 acts both as a blocking oscillator and discharge tube. The grid voltage waveform is illustrated in Fig. 18.7 (b). When the oscillator is cut-off by blocking action, CS charges through the series resistance of R1 and R2 towards B+. During oscillator pulses the tube conducts heavily and its plate resistance falls to a very low value. This provides a very low time constant path, for the capacitor CS to discharge, with discharge current in the same direction as the normal plate current during oscillator conduction. Thus the blocking oscillator behaves like a switch where the ‘on’ and ‘off’ periods are automatically controlled by the frequency of blocking oscillator. The ‘on’ and ‘off’ periods are set equal to the retrace and trace periods respectively.
C2 V1 C1 LS C3 R1 0.01 mF R3 2M R2 4M Hold control (a) + 0 –10 V
Grid voltage

Frequency Control Frequency of the oscillator can be adjusted by varying resistance R4 which is part of resistance in the grid-leak bias circuit. A lower time constant and smaller value of R4 will allow a faster

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331

discharge for a higher frequency. Increasing the time constant C1(R3 + R4) results in a lower oscillator frequency. The oscillator frequency control is adjusted to the point where sync voltage can lock the oscillator at the sync frequency to make the picture hold still. For this reason the frequency adjustment resistor R4 is generally called the hold control. The range of frequency control for the vertical oscillator is usually from 40 to 60. For the horizontal oscillator, which employs an automatic frequency control, the horizontal hold adjustment is usually provided in the AFC circuit. Height Control The capacitor CS is allowed to charge through R1 and R2 towards B+ for a fixed interval equal to the trace period. During this period tube V1 (Fig. 18.7(a)) remains is cut-off. Decreasing the resistance R2 reduces time constant. Hence CS charges at a faster rate and attains a higher voltage at the end of trace period. Thus a reduction in the value of R2 increases amplitude of the sawtooth voltage which, after amplification, causes a larger current through the deflecting coils to increase size of the picture. Similarly, increasing resistance R2 results in reducing size of the picture. The fixed resistance R1 in series with potentiometer R2 limits the range of variation for easier adjustment. This method of size control is generally used in vertical deflection circuits and is known as height control. Synchronizing the Blocking Oscillator A blocking oscillator in its free running state is not very stable and its frequency changes with variations in electrode voltages. This, however, can be easily controlled and kept constant by an external sync signal. The frequency can be synchronized either by sync pulses that trigger the oscillator into conduction at the sync frequency or by changing the grid bias with a dc control voltage. The vertical sweep oscillator is usually locked with pulses obtained by integrating the vertical sync pulses, whereas the horizontal oscillator frequency is synchronized with the dc control voltage produced by the horizontal AFC circuit.
Synchronized operation Free running 0 t

The waveforms of Fig. 18.8 (a) illustrate how a vacuum tube blocking oscillator can be synchronized by small positive pulses injected in the grid circuit of the oscillator. The sync voltage is applied in series with the grid winding of the transformer through a capacitor as shown in Fig. 18.7 (a). The positive sync pulses arrive at the time marked ‘sync’ when the declining grid voltage is close to cut-off and cancels part of the grid bias voltage produced by the oscillator. A small sync voltage is sufficient to drive the grid voltage momentarily above the cut-off voltage. This initiates plate current flow and then the oscillator goes through a complete cycle. The next positive sync pulse arrives at a similar point, in the following cycle, forcing the oscillator to begin the next cycle. As a result the sync pulses force the oscillator to operate at the sync frequency. Free Running Frequency of the Oscillator Operating the oscillator at the same frequency as the synchronizing pulses does not provide good triggering, because the oscillator frequency can drift, above the sync frequency, resulting in no synchronization. This is because the sync pulse will have no controlling influence when the tube or transistor has already been switched into conduction by its own bias. For best synchronization, the free-running oscillator frequency, is adjusted slightly lower than the forced or sync frequency so that the time between sync pulses is shorter than the time between pulses of the free running oscillator. Then each synchronizing pulse occurs just before an oscillator pulse and forces the tube or transistor into conduction thereby triggering every cycle, to hold the oscillator locked at the sync frequency. If the free-running frequency is too low, the sync pulses will arrive early and fail to raise the bias to a level that can cause conduction because the grid bias will still be at a large negative value. This also explains why equalizing pulses or any noise pulses which occur at the middle of the cycle fail to trigger the oscillator. False triggering due to noise pulses that occur close to the sync pulses can be reduced by returning the grid to a positive voltage instead of the chassis ground.

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333

DC Control Voltage A dc control voltage (see Fig. 18.8 (b)) can also be added to the existing bias voltage on the grid of the blocking oscillator tube or base of the oscillator transistor, to control its frequency. As a result of this addition, less time is needed for the bias voltage to reach its conduction level to start the next cycle. This method is often used to synchronize the horizontal oscillator frequency. Transistor Blocking Oscillator Sweep Generator In a blocking oscillator which employs a transistor, the method employed to turn the transistor ‘on’ and ‘off’ for generating sawtooth output is different than that employed with vacuum tubes. This frequently involves the use of sawtooth wave generated by the transistor itself, rather than by a grid-leak bias action as used in tube circuits. Figure 18.9 is the circuit of a typical vertical blocking oscillator-sawtooth generator, where sweep capacitor CS is in the emitter lead of the transistor. The combination of voltage divider resistors R1 and R2 and potentiometer R3 that are connected across the VCC supply, provide necessary forward bias to the transistor. When VCC supply is switched on, the rising collector current through the primary of feedback transformer T1 generates a voltage across its secondary, which drives the base more positive relative to emitter. This has two effects. The first is an increase in collector current, which in turn increases positive feedback at the base. This action is regenerative until saturation is quickly reached. The second effect is, that as the increasing base current drives the transistor to saturation, the capacitor C2 discharges making the conduction period still shorter. During the brief period when Q1 conducts heavily, the capacitor CS charges quickly (from the emitter current) to produce retrace period of the sawtooth wave. As Q1 saturates, the magnetic field about the transformer T1 stops expanding. The positive voltage that was induced at the base of Q1 then disappears. As a result, the combined effect of near-zero voltage at the base, because of potential drop across R2 and a positive voltage on the
+ R7 LP T1 Vertical hold R2 C2 + D2 LS R8 18 V VCC

emitter due to charge on CS, reverse biases the base-emitter junction and the transistor immediately goes to cut-off. This sudden change in collector current and the consequent collapse of magnetic flux in the primary of T1 induces a large reverse voltage in the secondary winding which aids in keeping the transistor in cut-off state. This voltage at the base could damage the transistor but for the protective action of diode D1. The diode acts as a short across the secondary (base) winding of the transformer during the brief back-emf period, protecting the transistor. After Q1 is cut-off, base capacitor C2 starts charging towards positive voltage at the junction of R1 and R2. It reaches this positive voltage level very quickly and then levels off. This is indicated by the base voltage waveform drawn along the circuit diagram. As soon as Q1 cuts off and emitter current ceases, CS starts discharging through R5 producing trace portion of the sawtooth waveform. When the capacitor has discharged to the point where emitter voltage no longer keeps Q1 cut-off, it turns on and the entire cycle is repeated. The frequency of the oscillator is controlled primarily by the time constant R5, CS. Note that during discharge of capacitor CS, Q1 is cut-off and network R5, CS is isolated from rest to the circuitry. This provides excellent frequency stability. Any change in the setting of potentiometer R3 alters forward biasing and thus controls the instant at which Q1 breaks into conduction. This changes frequency of the oscillator and so potentiometer R3 acts as ‘hold control’. Similarly potentiometer R6 which controls the magnitude of sawtooth voltage that is fed to the driver is called ‘size’ or ‘height control’. In transistor circuits the driver, which is an emitter follower, is necessary to isolate the oscillator from low input impedance of the corresponding output stage. Synchronization Sync pulses received as part of the composite video signal from the transmitting station to which the receiver is tuned are applied after due processing through diode D2 to the tertiary winding of the feedback transformer. While D2 prevents vertical oscillator waveforms from being fed back to the sync circuit, the tertiary (third winding) on T1 provides isolation between the oscillator and sync circuit. The positive going sync pulses drive the base more positive and turn on Q1 a little earlier that it would have normally under free running condition. The sync pulses thus lock the vertical oscillator to the transmitter field frequency to prevent any rolling of the picture. Another transistor blocking oscillator circuit is shown in Fig. 18.10. In this circuit sawtooth network is in the collector circuit and the sawtooth capacitor CS charges during trace period and discharged during retrace interval. The setting of potentiometer R2 controls initiation of conduction of the transistor and thus acts as ‘hold control’ over a narrow range of the oscillator frequency. The output voltage across CS is of the order of 1 volt and is enough to operate the following driver stage.

DEFLECTION OSCILLATORS
+ 18 V

335

330 W

R1

To input of the vertical stage driver

R2

D1

CS

+

100 mF

Hold control R3 5 mF + C1

T1 Q1 VCS D2

Sync input

0

t

Fig. 18.10. Vertical deflection blocking oscillator.

18.4 MULTIVIBRATOR DEFLECTION OSCILLATORS
A multivibrator is another type of relaxation oscillator which employs two amplifier stages, where the output of one is coupled to the input of the other. This results in overall positive feedback and the circuit operates such that when one stage conducts, it forces the other to cutoff. Soon the stage that cuts off returns to conduction to force the first stage to cut-off. This sequence repeats to generate square or rectangular output with a frequency that is controlled by the coupling networks between the two amplifier stages. As in the case of a blocking oscillator the multivibrator is used as a controlled switch to charge a capacitor through a resistance to generate the required sawtooth wave output. The amplifiers may employ tubes or transistors as active devices. Multivibrators may be classified as bistable, monostable and astable. A bistable multivibrator has two stable stages and needs two external trigger signals to complete one cycle of oscillation. The monostable type has one stable stage and completes one cycle of output with only one external pluses. However, and astable multivibrator is a free running type and does not need any external trigger pulse for its normal operation. It is this type of multivibrator that is employed as a deflection oscillator and its frequency is synchronized with the horizontal AFC voltage or vertical sync pulses. Multivibrators can also be classified on the basis of coupling between stages. The two types that are used in TV receivers are plate (or collector) coupled and cathode (or emitter) coupled. Transistor Free Running Multivibrator The circuit configuration shown in Fig. 18.11 (a) is of a free running collector coupled multivibrator where two common emitter amplifiers are cross coupled to provide positive

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feedback. Note that the base resistances are returned to VCC supply to ensure precise operation of the multivibrator. The circuit operation can be easily explained if the sequence of operations is followed from the instant when one transistor just conducts and the other goes to cut-off. Figure 18.11 (b) illustrates the collector and base voltage waveforms for one complete cycle. At the instant marked t0, transistor Q2 just conducts to saturation and transistor Q1 returns to cut-off. As this happens, the rising voltage at the collector of Q1 charges the capacitor C2 to VCC, since, VBE(sat)) ≈ 0. The charging current of C2 flows through the base of Q2 to complete its circuit. RB2 is selected to provide enough current from VCC to the base of Q2, to keep it in saturation, even when the charging current of C2 becomes zero and C2 charges to VCC.
+ VCC VCE1 VCC RL1 RB2 C2 + RB1 R1 R2 C1 + Q2 vBE1 C3 vBE2 vCE2 RL2 v0 Q1 vCE1 (Q1) –VCC vCE2 VCC (Q2) 0 vBE2 (Q2) –VCC Off Q1 C1 On RB1 – + vC1 + VCC +VCC 0 –VCC (c) (d) t vC1 (vC2) (b) tA On t0 t1 tB Off t2 t (Q1)

In a similar manner C1 would have got charged to VCC in the previous cycle when Q1 was in saturation. Actually at t = t0 the capacitor C1 which was previously charged to VCC gets earthed with its positively charged plate towards ground, the moment Q2 goes into full conduction. As shown in Fig. 18.11(c), C1 is then in parallel with emitter-base junction of the ‘off’ transistor Q1. This puts a reverse bias on Q1 equal to – VCC at t = t0, which is well beyond cut-off bias of the transistor. The capacitor C1 now starts charging from – VCC towards + VCC as shown in Fig. 18.11 (d). At t = t1 the negative voltage across C1 reduces to zero and permits base current flow in the

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337

transistor Q1. This action is regenerative and Q1 instantly goes to saturation which in turn cuts-off Q2. The time constant RB1C1 controls the off period (t1 – t0) of Q1 and can be set equal to the retrace period of the required sawtooth wave. The moment Q1 goes into saturation the positively charged plate of C2 is effectively grounded through Q1. It then begins to charge towards + VCC at a rate determined by the time constant RB2C2. Once again when VC2 = 0, the second transition takes place to complete one cycle of operation. This cycle then repeats and permits the circuit to function as a free running multivibrator. The time constant RB2C2 can be made equal to the trace period. As shown in the circuit, RB1 consists of R1 and R2 where R2 is a potentiometer to adjust the retrace period and thereby controls the frequency of the multivibrator. The output voltage at either of the collectors is a rectangular wave with an amplitude = VCC – VCE(sat). Multivibrator Frequency As shown in Fig. 18.11 (b) the total period

1 = tA i.e. (t – t0) + tB i.e. (t2 – t1). f A reference to Fig. 18.11 (c) will show that VC1 takes a time equal to tA to return to zero from – VCC while charging toward + VCC . Then at the instant t1, νc1 = 0 and we can write.
T= 0 = VCC – (VCC + VCC) exp (– tA/RB1C1) This expression when solved yields tA = 0.69 RB1C1. Similarly it can be shown that tB = 0.69 RB2C2. ∴ Synchronization The synchronizing pulses may be positive or negative and may be applied to the emitter, base or collector of one or both the transistors. The frequency of switching action of the astable multivibrator is kept lower than that of the synchronizing pulses, to force the transistor to switch states slightly before the free-running transition time. The sync pulses are applied at the base of the controlling transistor. The ‘on-off’ periods of the multivibrator are used to generate sawtoothed output across a capacitor as explained in the previous sections. Cathode/Emitter Coupled Multivibrator In this type of multivibrator only one RC coupling is provided between the two amplifiers. Positive feedback for regenerative action is obtained through a common resistance in the cathode/emitter leads of the two amplifiers. The oscillator action is explained by considering a cathode coupled multivibrator. As shown in Fig. 18.12 (a) the coupling from tube V1 to V2 is through an RC network while from V2 to V1 it is through the common cathode resistance RK. When V1 conducts the reduced plate voltage is coupled to the grid of V2 via C2 thereby cutting it off. Thus the cut-off period of V2 depends on the time constant C2 (100 KΩ + R2) while C2 discharges. However, when V2 goes into conduction, the cut-off period of V1 depends on the time constant for charge of C2. The charge path is through the low resistance of grid-cathode of V2 (when grid current flows), RK and RL1 while V1 is off. As a result C2 charges fast to provide a small cut-off period for V1. The cathode coupled multivibrator therefore automatically produces T = tA + tB = 0.69 RB1C1 + 0.69 RB2C2.

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an unsymmetrical output as V2 must remain in cut-off for a much longer period than V1. Thus V2 provides the trace period and V1 the retrace period. Sawtooth Generation The circuit of Fig. 18.12 (a) can be modified as shown by dotted chain lines to generate a sawtooth output for feeding into the horizontal output deflection amplifier. The sawtooth capacitor is connected in the plate circuit of V2, which stays in cut-off for a long time and conducts for a short time. This stage then acts as a discharge tube to generate a sawtooth output as explained earlier. Figure 18.12 (b) illustrates how the sawtooth voltage output corresponds to cut-off and conduction periods of V2. The variable resistance R2 in the grid circuit of V2 serves as the frequency (hold) control. The grid resistance R1 for V1 does not control the oscillator frequency because its grid voltage is controlled by the voltage drop across. Rk. However, this grid is best suited for frequency control because of its isolation from the oscillator voltages. Frequency Control The horizontal sweep oscillators are normally synchronized by the negative dc control voltage obtained from the AFC circuit. In the cathode coupled multivibrator (Fig. 18.12 (a)), negative dc control voltage is applied at the grid of V1. The added negative grid voltage reduces plate current of V1 when it conducts. This results in a smaller drop in its plate voltage and less negative drive at the grid of V2. Then less time is needed for C2 to discharge down to cut-off. This reduction in the cut-off period results in an increased multivibrator frequency. Variations in the AFC negative voltage thus apply necessary correction in the cut-off period of V2 to keep the oscillator synchronized with the horizontal sync pulses.
B+ Ip1 RL1 C2 V1 Sync input 1M R1 330 PF 100 K 100 K Frequency control R2 Cut off Conducting

Fig. 18.13. Grid voltage waveform of a multivibrator synchronized by sync pulses. The oscillator is pulled in to the sync frequency at time t3.

Multivibrator Synchronization As stated earlier, while discussing sync processing circuits, either positive or negative sync polarity can be used with multivibrators. A Positive pulse, applied to the grid of a tube in cutoff, can cause switching action if the pulse is large enough to raise the grid voltage above cutoff. A negative sync pulse at the grid of V1 when it is in conduction, results in a more stable operation and is generally used. In fact, the negative pulse applied to the gird of V1 gets amplified and inverted to appear as a large positive pulse at the grid of V2. It is highly improbable that the first sync pulse which arrives will succeed in synchronizing the oscillator. The waveforms shown in Fig. 18.13 illustrate how the oscillator is gradually pulled into synchronism. It would be pertinent to mention here that for a blocking oscillator only a positive sync pulse can cause synchronization. Multivibrator Stabilization As the grid voltage approaches its cut-of value, it becomes increasingly sensitive to noise pulses which might have become part of the signal. A strong such pulse, arriving slightly before the synchronizing pulse can readily trigger the oscillator prematurely and cause rolling or tearing of the picture. To ensure stability of operation, resonant circuits are employed in some sweep circuits. A cathode coupled multivibrator of the type shown in Fig. 18.12 (a) is redrawn in Fig. 18.14 (a), with a resonant stabilizing circuit placed in the plate circuit of tube V1. the frequency of the resonant circuit is adjusted to 15625 Hz. The tuned circuit is shock excited by periodic switching of the tube from an ‘on’ to an ‘off ’ condition. As a result the sinusoidal output of the resonant circuit modifies the voltage waveforms at the plate of V1 and grid of V2. This is illustrated in Fig. 18.14 (b), where of particular importance is the grid waveform of triode V2. It may be noted that the grid voltage now approaches cut-off very sharply and only a very strong noise pulse will be able to trigger the second triode prematurely.

18.5 COMPLEMENTARY-SYMMETRY RELAXATION OSCILLATOR
A complementary-symmetry relaxation oscillator, designed to drive the vertical deflection output circuit, is illustrated in Fig. 18.15 (a). Transistors Q1(p-n-p) and Q2(n-p-n) which are directly coupled, constitute the oscillator pair, while Q3 is the waveshaping transistor. Resistors R1 and R2 form a potential divider across VCC supply through the decoupling network R3, C6, to provide positive voltage both at the base of Q1 and collector of Q2. The voltage at the emitter of Q1 is developed by capacitor C1, when it charges towards VCC (+ 20 V), through resistance R4 and potentiometer R5 connected in series. At the instant dc supply is switched on to the circuit, both the transistors are at cut-off, because the base of Q1 is biased positively and its emitter is at zero potential. However, capacitor C1 starts charging at once driving emitter of Q1 positive. When the rising voltage across C1 offsets the positive voltage at the base of Q1, the transistor turns on. This makes the base of Q2 positive which also goes into conduction. The collector current of Q2 flows through R1 and the resulting drop across it, lowers the potential at the base of Q1, thus making it more negative with respect to its emitter. This results in increased current through Q1 and the regenerative feedback action that follows soon saturates Q1. When Q1 is on, its emitter current starts discharging C1. As soon as the emitter voltage drops sufficiently to remove forward bias on Q1 it is driven out of conduction. This in turn cuts off Q2, thereby completing the regenerative cycle. The capacitor C1 starts charging again towards VCC to repeat the sequence of events explained above. In the absence of any sync input, Q1 and Q2 repeat the on-off cycle at a rate determined by the time constant C1(R4 + R5). This time constant determines frequency of the oscillator. Potentiometer R5 is the ‘hold control’ and can be varied to change the frequency. A negative going vertical sync pulse applied at the base of Q1, through the integrating networks R9, C3

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341

and R10,C4, brings the transistor out of cut-off before it normally would. This synchronizes the oscillator to the applied sync pulses. The voltage waveform (i) at the emitter of Q1 (see Fig. 18.15 (b)) is a sawtooth wave developed by the charging and discharging of C1. Waveform (ii) is the sharp positive pulse developed at the emitter of Q2 when both the transistors are turned on.
+ 20 V VCC R3 C6 R1 R4 R9 Sync input C3 R10 C4 R2 Q2 ii Q3 R6 + Q1 i + R7 C1 C5 To vertical output amplifier iii C2 R5 + VCC From horz output circuit R8 i

ii

iii

Fig. 18.15 (a). Complementary symmetry relaxation oscillator.

Fig. 18.15 (b). Wave shapes at the emitters of Q1, Q2 and collector of Q3.

Wave Shaping The wave shaping transistor Q3 is normally biased to cut-off. It is triggered ‘on’ by the positive pulse developed at the emitter of Q2 and directly coupled to its base. Capacitor C2 connected across the collector and emitter of Q3 charges during the ‘off ’ period of Q3 and discharges when the transistor turns on for a short interval of time. The resulting sawtooth voltage across C2 (waveshape(iii)) is coupled through C5 to the vertical output stage of the receiver. The rate at which C2 charges during the ‘off ’ interval of Q3 is determined by resistors R7 and R8. R7 is a potentiometer that can be varied to control the amplitude of the vertical sweep and through this the height of the picture. The voltage towards which C2 charges is derived from a rectifier in the horizontal output circuit. Any change, in the horizontal deflection output voltage, will also alter the dc voltage fed to this circuit and affect the amplitude of the sawtooth voltage. Thus, if horizontal size of the picture changes, vertical size also changes proportionately, preserving the ratio of height to width of the picture.

18.6 SINE-WAVE DEFLECTION OSCILLATORS
High frequency sine-wave oscillators are more stable in their operation as compared to corresponding relaxation oscillators. Since the horizontal sweep frequency is quite high such oscillators are often used in horizontal deflection circuits. For such an application the oscillator is overdriven so that the tube or transistor acts like a switch and allows a sawtooth forming

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capacitor to charge and discharge. The frequency of the oscillator is controlled by a reactance tube or transistor placed between the AFC circuit and oscillator. Reactance Tube Sweep Generator A typical circuit of a vacuum tube horizontal deflection oscillator cum sweep generator is shown in Fig. 18.16. It employs a tuned grid configuration where mutual coupling between coils L1 and L2 can be varied to change the oscillator frequency. The desired frequency is 15625 Hz. The screen grid of pentode V2 acts as plate for oscillator action. This isolates the oscillator circuit from large plate voltage variations which occur on account of switching action of the tube.

The oscillator uses self-bias and develops about – 15 V at the control grid. For a pentode, this voltage is enough for class C operation. The oscillator is overdriven so that plate current flows in short-duration pulses only during extreme positive peaks of the grid voltage, The capacitor C1 charges towards B+ to develop trace period when the tube is off. The retrace is formed when the tube conducts for a short duration. The high (39 K) plate load resistance R2 ensures a large (130 V P-P) sweep voltage. Frequency Control The reactance tube V1 shunts the oscillator tank circuit and appears as a capacitive reactance. As explained in Chapter 7, Such a behaviour can be simulated by a capacitive feedback between plate and control grid of the tube. The equivalent capacitance across the output terminals of the tube is proportional to mutual conductance (Ceq = gm × RC) of the triode. In the circuit under consideration, the bias of the reactance tube is the combined result of AFC derived grid voltage and voltage drop across R3, the cathode resistor. And drift in the oscillator frequency results in a change in the dc control voltage. This in turn shifts the operating point to change gm of the reactance tube. The consequent change in the simulated equivalent capacitance Ceq, forces the oscillator to correct its frequency. The continuous feedback action results in a very stable operation of the oscillator.

As shown in the circuit diagram a reactance transistor (Q1) is used between the AFC output and horizontal oscillator. It acts like an inductor instead of a capacitor, In order to simulate such a behaviour the phase shift network C2, R2 couples some of the oscillator’s voltage into the emitter of the reactance transistor and at the same time shifts its phase by 90°. Since the feedback is returned to the emitter, transistor Q1 can be considered to be a common base amplifier. As such, the emitter voltage and collector current are 180° out of phase. Thus the transistor current lags behind the collector voltage by 90°. Therefore, the oscillator ‘sees’ the reactance transistor as if an inductor has been connected across its tank circuit. Any change in the oscillator frequency is sensed by the AFC circuit which couples a proportionate dc error voltage into the base of the reactance transistor. This shifts the operating point of the transistor in the same way as AGC bias does to control gain of tuned amplifiers. Any decrease in oscillator frequency results in a positive dc control voltage thereby increasing forward bias of the reactance transistor. The resulting increase in collector current shifts the operating point where the transistor gain is lower. In turn, this acts to decrease the effective reactance at the output terminals of the transistor. This causes the total inductance shunting the oscillator tuned circuit to decrease. As a result, frequency of the oscillator is increased to return to its correct value. The reverse would occur if the bias were to decrease for any increase in the oscillator frequency.

Review Questions
1. 2. Sketch and label the current waveforms that must flow in the deflection yoke coils to produce a full rester. Explain the basic principle of generating such waveforms. Why is a trapezoidal voltage waveform necessary to drive the vertical deflection coils ? What is the effect of source impedance and frequency on the shape of the driving voltage waveform ? How is the basic sawtooth voltage modified to obtain the desired driving voltage waveform ?

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Draw the circuit diagram of a blocking oscillator-cum-waveshaper which employs a single triode and discuss its operation. Explain the operation of ‘hold control’ and ‘height control’ as provided in the circuit drawn by you. A blocking oscillator that employs a transistor is shown in Fig. 18.9. Explain how the circuit operates to develop a sawtooth voltage. In particular, explain the operation of hold and height controls. Why are diodes D1 and D2 provided in the feedback transformer circuit ? Explain with suitable waveforms how the frequency of the blocking oscillator is controlled with the help of sync information. Why is the free running frequency of the oscillator kept somewhat lower than the desired frequency ? What will happen if the uncontrolled frequency is higher than the correct frequency ? Draw the basic circuit of a multivibrator employing transistors. How are the feedback networks designed to obtain correct trace and retrace periods ? Explain how the sync pulses control the frequency of such an oscillator. The circuit of a complementary-symmetry relaxation oscillator is given in Fig. 18.15 (a). Describe briefly how the circuit operates to generate sawtooth output voltage. Discuss with suitable waveforms how the sync pulses keep the oscillator in synchronism with corresponding oscillator at the transmitter. Why is a sine-wave oscillator preferred in horizontal deflection circuits ? Explain with a circuit diagram how a reactance tube/transistor connected between the AFC circuit and oscillator operates to maintain a constant frequency.

4.

5.

6.

7.

8.

19
Vertical Deflection Circuits

19
Vertical Deflection Circuits
The output from the vertical and horizontal oscillators is too low to drive the deflection coils to generate a full raster. Therefore, amplifiers are used to produce enough power to fully drive the coils in the deflection yoke. The horizontal oscillator drives the horizontal output stage to produce horizontal scanning and the vertical oscillator drives the vertical output stage to produce vertical deflection. These two motions occur simultaneously to produce raster on the screen.

19.1 REQUIREMENTS OF THE VERTICAL DEFLECTION STAGE
The purpose of a field-scan output amplifier is to convert the input sawtooth or trapezoidal voltage waveform into a sawtooth current in the field deflection coils, of sufficient magnitude to scan the face of the picture tube. There are many different vertical-output circuit configurations in use that employ all tubes, combined tubes and transistors (hybrid) and only transistors. However, all circuits operate under class ‘A’ or ‘AB’ condition to ensure linear operation. The limitations of inductive coupling that is employed between the output stage and yoke coils, together with the imperfection of active devices, make it necessary to modify the output voltage waveform to achieve linear deflection. This necessitates the use of negative feedback and other waveshaping techniques. Furthermore, transistor circuits need somewhat different approach than the one employing vacuum tubes. All this tends to make the deflection circuits somewhat complex. However, this problem is solved by first discussing various sections of the circuit separately and then putting them together to study the complete set-up. (a) Vertical Yoke Drive A voltage stepdown transformer provides an efficient means of coupling the output stage to the deflection coils. Figure 19.1 (a) shows such an arrangement that employs a vacuum tube as the vertical sweep amplifier. The output transformer matches the relatively high impedance of the tube to the low impedance deflection coils. The impedance of the coils is mainly resistive because of the low frequency spectrum that constitutes the 50 Hz trapezoidal sweep voltage. This makes the design of the output circuit basically the same as that of an audio output stage to match a loudspeaker. In order to amplify the 50 Hz sawtooth with little distortion the frequency response of the amplifier must extend down to 1 Hz. The turns ratio of the output transformer is chosen to match the scanning coil impedance to the optimum output impedance of the tube. This is in the range of 30 : 1 to 6 : 1 depending upon the impedance of the coils. Typical value of primary inductance is around 45 mH with a dc resistance of the order of 40 ohms. The resistance of the secondary winding is quite low and varies between 5 and 10 ohms. 346

In many circuit designs an autotransformer coupling (Fig. 19.1 (b)) is employed because of its higher efficiency. The position of the tap determines the effective turns ratio of the transformer. If all other conditions are indentical, for the same input power an autotransformer gives a greater output power. The primary and secondary ac currents flow in opposite directions through a common portion of the autotransformer winding and thus tend to cancel each other. Since the net current flowing through these windings is smaller, power losses are less as compared to an isolated winding transformer. This means that for the same power ratings a thinner wire can be used in an autotransformer. In practice this results in a smaller and less expensive transformer. A shown in Fig. 19.1 (b) a coupling capacitor (C2) is used to prevent any dc current flow in the deflection coils which would otherwise shift centering of the electron beam. The capacitor C3 connected between secondary tap and ground is effectively in parallel with the secondary circuit. It improves frequency response of the transformer by speeding up collapse of the induced field during retrace. The self-induced voltage in the secondary winding during retrace is stepped up because of the large turns ratio between the primary and secondary. Therefore, the transformer winding must be insulated to withstand induced voltage peaks. As shown in Figs. 19.1 (a) and (b) the

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network R1,C1 provides decoupling, to prevent any undesired feedback from output stage to any other stage through the common dc source. In vertical output stages that employ transistors, an output transformer is not necessary, because power transistors provide large output current at a relatively low impedance level, and thus can be coupled through a capacitor directly to the yoke coils. The circuit arrangement that is often used is shown in Fig. 19.2. The choke shown in the collector circuit isolates the collector signal path from the power supply. There are some solid state vertical output circuits which use an isolation transformer with a unity turns ratio. This arrangement provides ac driving current to the yoke without the need for a large electrolytic coupling capacitor. A VDR (voltage dependent resistance) is often used in the output circuit across the choke (Fig. 19.2) to protect the output transistor against voltage transients in the collector circuit.
+ 12 V R1 vin CC Choke Q2 Output VDR L1 Deflection coils L2

Q1 Driver R2

1V

+

120 V

Fig. 19.2. Basic circuit of a transistor vertical amplifier.

(b) Vertical Deflection Coils The magnetic field for vertical deflection is developed by two coils mounted 180° apart on the neck of the picture tube. Physically these coils are a part of the yoke assembly that also contains horizontal windings. During the vertical deflection period the flux generated by the sawtooth current in the windings, moves the electron beam down and then returns it to the top for a quick retrace. The vertical yoke inductance and resistance are kept as low as possible to accomplish higher deflection efficiency. Most field scanning coils are designed with an inductance that varies between 3 mH and 50 mH with a resistance of 4 to 50 ohms. The impedance and number of turns of the coils are dependent on the required deflection angle. Wider deflection angle tubes require a stronger magnetic field to cover the entire raster. The peak-to-peak sawtooth current needed by different vertical deflecting coils varies between 300 mA to 2.5 Amp. Typically, a short neck 36 cm picture tube requires a 45 mH deflection coil and a peak-to-peak driving current of 400 mA. The two halves of the deflecting coils are normally connected in series across the secondary of the drive transformer in vacuum tube circuits. However, in transistor circuits these are connected in parallel to provide impedance match for maximum efficiency. (c) Vertical Linearity The patterns of Fig. 19.3 illustrate the effect of three different current waveforms on the reproduced raster. When the sawtooth wave is linear (Fig. 19.3 (a)), the vertical scan allows equal spacing between horizontal lines and there is no distortion. In Fig. 19.3 (b) the sawtooth wave tends to become flat towards its close, hence the magnetic field increases at a reduced rate with the result that the horizontal lines crowd together at the bottom of the raster. In the

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third pattern (Fig. 19.3 (c)) the horizontal lines are crowded at the top of the raster because of the slow rate of change of current at the beginning of the sawtooth wave. Non-linear vertical scanning is more noticeable than horizontal non-linearity. Crowding of horizontal lines in any region of the raster makes the objects look flatter in that part of the picture. However, when the scanning rates becomes faster than normal in any portion of the raster the corresponding picture area appears elongated. Non-linearity may be caused due to any of the following reasons.
Top Vertical deflection sawtooth Trace Retrace

Fig. 19.3 (b). Non-linear sawtooth causing crowding of lines at the bottom of the raster.

Fig. 19.3 (c). Non-linear sawtooth causing crowding of lines at the top of the raster.

(i) A short time-constant sawtooth forming RC circuit develops a non-linear sweep because too much of the exponential charging curve forms part of the trace period. Similar nonlinear compression at the bottom of the raster may be caused by positive peak clipping of the trapezoidal voltage driving the grid/base of the vertical output stage. This is generally the result of incorrect tube/transistor bias. However, such shortcomings can be easily avoided by a careful design of the sawtooth forming circuit and appropriate choice of the device’s operating bias. (ii) The magnetizing current of the coupling transformer will also cause nonlinear distortion. This can be made quite small if the transformer has a high primary inductance. This, however, leads to a heavier and costlier output transformer. The solution to this problems, which is often employed, is to use a transformer with nottoo-high a primary inductance but modify the input voltage to the amplifier for optimum results.

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(iii) Another aspect, which needs attention, is the attenuation caused at low frequencies by the finite primary inductance of the output transformer, that effectively appears in parallel with the deflecting coils. This is corrected by applying negative feedback from the output circuit to input of the amplifier through a frequency selective network. (iv) Another problem is the pincushion distortion. Since magnetic deflection of the electron beam is along the path of an arc, with point of deflection as its centre, the screen should ideally have corresponding curvature for linear deflection. However, in flat screen picture tubes it is not so and distances at the four corners are greater as compared to central portion of the face plate. Therefore electron beam travels farther at the corners causing more deflection at the edges. This results in a stretching effect where top, bottom, left and right edges of the raster tend to bow inwards towards the centre of the screen. The result, as illustrated in Fig. 19.4, is a pincushion like raster. Such a distortion is more severe with large screen picture tubes having deflection angles of 90° or more.
Top pincushioning

Side pincushioning

Raster

Fig. 19.4. Pincushion distortion of the raster.

In black and white receivers, pincushion distortion is eliminated by suitable design of the yoke and the use of small permanent magnets mounted close to the yoke. These magnets are so positioned that they stretch the raster along the sides of the picture and thus compensate for the pincushion distortion. In colour picture tubes the deflection current in the yoke is modified by special pincushion correction circuits.* In output stages, (see Fig. 19.5) which use power triodes, the linearity is often controlled by shifting the operating point on the transfer characteristic, by varying a part of the cathode bias resistance. The variable resistance then performs as Linearity Control. Transistor vertical output stages are generally similar to their vacuum tube counterparts. Since the input impedance of a power transistor is only the order of a few hundred ohms, it becomes necessary to drive the output stage from an emitter follower (driver) so as not to place too high a load on the sawtooth generator. In high power transistor circuits some form of thermal stabilization is also provided for stable operation. This ensures linear operation by
*This is explained along with dynamic convergence circuits in chapters devoted to colour television.

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maintaining the operating point at the set optimum value. Negative feedback is also used to maintain a frequency response that is flat down to 1 Hz.

Fig. 19.5 (b). Cancellation of non-linear curvature of the sawtooth wave by optimum choice of the operating point on the output characteristics of the tube.

Whenever possible direct coupling is employed between the driver and output stage to avoid use of a large coupling capacitor. If necessary, linearity correction is also employed by modifying the waveshaping network. This is fully explained in a later section of the chapter while describing a typical transistor output circuit. (d) Suppression of Undesired Oscillations During retrace time when the current through vertical deflecting coils suddenly drops to zero, energy stored in the collapsing magnetic field sets up high frequency oscillations. The frequency

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of ‘ringing’ thus produced, depends on the inductance and the distributed capacitance of the deflecting coils. The effect of such oscillations, if not suppressed, would be to pull the beam upwards and downwards alternately instead of the smooth motion downwards. This results in a narrow white strip at the top of the screen. The strip then appears more bright than the rest of the raster. To prevent ringing two shunt damping resistances are connected across the two halves of the vertical scanning coils. Figure 19.6 (a) illustrates the ringing effect on the deflection current and Fig. 19.5 (a) the use of damping resistors R1 and R2 across the vertical scanning coils. It may be noted that the output from the amplifier contains considerable power and the damping resistors do not attenuate the signal appreciably. (e) Height of the Raster To fill the entire raster from top to bottom of the screen a definite amplitude of sawtooth current must flow into the deflecting coils. The output amplifier is designed to meet this requirement. In addition a control is provided at the input of the amplifier to finally adjust the height of the picture. This control is known as ‘height control’. In some designs the linearity control resistor when provided in the cathode (or emitter) lead of the amplifying device is left unbypassed (by a capacitor) and also acts as the height control. A change in the linearity control resistance varies gain of the stage which in turn controls the height of the raster. The effects of either insufficient or excessive drive are illustrated in Fig. 19.6 (b) which clearly indicate the need for a ‘height control’.
Screen T1 T2 T3 i(v) Undesired oscillations (S1) Correct amplitude (T1) (S2) Excessive amplitude (T2) 0 t (S3) Insufficient amplitude (T3)

Fig. 19.6 (a). Effect of ringing on the vertical deflection current.

Fig. 19.6 (b). Effect of deflection current amplitude on the height of the raster.

Another factor which affects the height of the picture is the increase in resistance of the deflection coils when current flows through them. To counteract this, a thermistor (see Fig. 19.5) is added between the coils in the yoke to equalize the resistance under high heating conditions. When temperature of the windings rises, the thermistor also gets heated and its resistance decreases sufficiently to compensate for the increase in winding resistance. This maintains a constant deflection current to prevent any change in the height of the raster. (f) Vertical Rolling of the Picture The rolling of the picture upwards or downwards on the screen occurs on account of incorrect vertical scanning frequency. This is illustrated in Fig. 19.7 for the case when the vertical

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oscillator frequency is higher than 50 Hz. It may be noted that relative timings of vertical blanking in the composite video signal and vertical retrace in the sawtooth deflection current are not the same, with the result that vertical blanking occurs during trace time instead of during retrace time. For a vertical frequency that is higher than 50 Hz, each sawtooth advances into trace time for succeeding blanking pulses. As a result the black bar produced across the screen, by the vertical blanking pulse, appears lower and lower down the screen for successive cycles. The opposite effect takes place when the oscillator frequency is lower than 50 Hz. The picture rolls at a faster rate if the oscillator frequency deviates much from 50 Hz. However, when the sync frequency and sawtooth frequency are the same, every vertical retrace occurs within the blanking time. The blanking bars are then produced at the top and bottom edges of the raster and are not visible. Thus, when the vertical oscillator is locked by sync pulses each frame is reproduced over the previous one, and the picture holds still.
Vertical sync Vertical sync

Video signal Vertical blanking time Vertical sawtooth

50 Hz

60 Hz

Picture

Rolling bar

Fig. 19.7. Vertical rolling when the deflection frequency is more than 50 Hz. Note that horizontal blanking details have been omitted for clarity.

As explained in the previous chapter, the oscillator frequency is controlled by the ‘hold control’. Since no elaborate sync processing circuitry is provided for the vertical sync pulses, the vertical hold control is provided on the front panel of all TV receivers, for resetting the frequency, when picture rolls continuously up or down. The vertical jumping of the picture that sometimes occurs should not be confused with vertical rolling. Vertical jumping occurs on account of unequal sync pulse amplitudes for odd and even fields. Occasional jumping is caused by wrong triggering of the vertical oscillator by a stray noise pulse. (g) Internal Vertical Blanking The voltage pulses produced during retrace intervals, in the vertical output circuit are coupled to the picture tube to provide additional blanking during vertical retrace time. This is in addition to the blanking voltage at the cathode or grid circuit of the picture tube, which is part of the composite video signal. In receivers, that do not fully retain dc component of the video signal at the picture tube, the blanking pulses fail to reach the required level to cut-off the beam,

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when vertical retrace is taking place. Thus the horizontal lines that get traced during this period, appear on the screen. However, the use of blanking pulses ensures complete blanking during retrace for any setting of the brightness control, with or without the dc component. All receivers use a vertical retrace suppression circuit, to turn-off the electron beam, until it is back at the top of the screen to begin the next vertical trace. Figure 19.8 shows two different circuit arrangements, one for obtaining positive pulses, that are suitable for injecting in the cathode circuit and the other for obtaining negative pulses to be applied at the grid of picture tube. The amplitude of these pulses varies between 50 to 150 V in different circuits.

In transistor circuits, where such high amplitudes cannot be obtained, the low amplitude pulses as obtained from the output stage are fed into the video amplifier, which amplifies them before feeding to the picture tube circuit. In the circuit shown in Fig. 19.8 (a) the pulses are obtained at the primary of the output circuit to yield positive going pulses, whereas in the circuit of Fig. 19.8 (b) negative going pulses are obtained from the secondary of the same output transformer. Based on the above discussion about the problems of vertical output stage and the techniques used to overcome them, it is reasonable to assume that a large number of circuit arrangements are possible to achieve linear vertical deflection. However, a few typical circuits are discussed to broadly identify various configurations that have found wide acceptance in television receiver practice.

19.2 VACUUM TUBE VERTICAL DEFLECTION STAGE
Figure 19.9 is the circuit of a vertical output stage which is widely used in vacuum tube receiver. As is the common practice, a single tube (triodepentode), serves both as oscillator and output stage. While the pentode section performs as vertical output stage, both triode and pentode (V1 and V2) operate together as an unsymmetrical free-running plate coupled multivibrator. Positive

For improved linearity and constant aspect ratio the dc voltage for charging C2 is obtained from the horizontal output circuit. The VDR1 connected across R3 and ground stabilizes this dc source. Any change in width of the picture will indirectly affect B++ supply and hence vary amplitude of the sawtooth voltage. The resultant change in drive voltage to the vertical output stage will vary height of the picture, thereby maintaining a constant aspect ratio. The setting of potentiometer P3 makes it possible to alter the shape of sawtooth voltage applied to the grid of V2. This is done to obtain a certain curvature in the sawtooth current that flows through the deflection coils for improving overall linearity. In addition, setting of potentiometer P4 (100 K) in the negative feedback path from winding LS2 to the grid of V2 (pentode) affects the form of drive voltage just after flyback and hence serves as top linearity adjustment. The network R6 (2 M), C5 (0.2 µf) provides a small negative feedback from grid of V2 to the input circuit of V1 to ensure stable operation of the circuit. The vertical sync pulses are integrated by network R4, C4 which constitutes a low-pass filter. The output across C4 is passed through the parallel combination of diode D1 and resistance R5 to obtain sharp positive going pulses. These are fed to the grid of V1 to synchronize the frequency of the multivibrator. The vertical blanking pulses are taken from the secondary of the output transformer through capacitor C3. The voltage dependent resistance VDR2 provides protection to the output tube and transformer by suppressing high voltage transients that develop across the primary winding during retrace periods.

The capacitors C1 and C2 charge towards VCC2 to generate the trace portion of the sawtooth wave. Initially dc voltage at the collector of Q1 is low and it stays at cut-off. As the capacitor C1 charges and reaches a certain value, Q1 starts conducting. It is reinforced by positive feedback from Q3 to the base of Q1. This action being regenerative, Q1 instantly goes to saturation and Q3 is cut-off. The series capacitors C1 and C2 then quickly discharge to almost zero voltage through the collector-emitter path of Q1 thus generating retrace period. At this point Q1 again returns to cut-off because of disappearance of dc voltage, both at its collector and base. A portion of the sawtooth wave at the emitter of Q3 is fed through potentiometer P2 and resistance R9 to the junction of C1 and C2, where capacitor C2 integrates it to form a parabolic wave. Since this is in series with the voltage across C1, the shape of the sawtooth wave fed to the base of Q2 and through that to the base of Q3 is also modified. This is done to improve linearity of the sawtooth current to the yoke coils. Potentiometer P2 thus acts as linearity control. Similarly potentiometer P3 is adjusted to control negative feedback in the output stage and acts as vertical height control.

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The base current to transistor Q1 is adjusted by potentiometer P1, which in turn determines the instant when Q1 will start conducting in each cycle. In this way potentiometer P1 controls frequency of the multivibrator and operates as ‘hold control’. The voltage VCC2 for charging C1 and C2 is obtained from the dc voltage, that develops across C4 by horizontal flyback pulses coupled through diode D1 to charge it. This capacitor is sufficiently large to provide a steady dc source of about 12 volts. The reason for using horizontal flyback pulses to provide dc voltage is to make the vertical output dependent on the horizontal scanning amplitude. If excessive load current reduces horizontal scanning width, the vertical height will also decrease thus maintaining correct aspect ratio of the raster. Synchronization is achieved by feeding positive sync pulses at the base of Q1 through C3, R3. The circuit has very good frequency response because of direct coupling between Q1 and Q3. However, the value of coupling capacitor C5 limits the response at very low frequencies.

is set by P2 (10 K Pot), to adjust height of the picture. The sweep voltage is shaped by network C2, C3, R2 and P3. Potentiometer P3 is used as linearity control. Linearity is also improved by negative feedback (i) from the output circuit to emitter of Q1 through R3, and (ii) by passing the deflection yoke current through R4, the emitter resistance of driver transistor Q2. A thermistor (Th1) connected in the base circuit of output power transistor Q3 is placed on the heat sink as a check against any possibility of thermal runway. Any undue increase in collector current of Q3 will cause more heating. The consequent increase in temperature of the heat sink will lower resistance of the thermistor. In turn, the reduced emitter-base drive to Q3 will

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lower its collector current. Transistor Q3 is also stabilized by leaving its emitter resistance R5 unbypassed. Potentiometer P4 is varied for optimum bias, more so, when it becomes necessary to replace Q3. The VDR1 connected across T2 keeps the amplitude of output waveform within narrow limits. The sweep output is fed to the deflection coils (connected in series) through C4, a large electrolytic capacitor. The induced sharp pulses (typically 50 V) during retrace serve as vertical blanking pulses. The oscillator frequency is set by varying P1 (hold control) to be slightly less than 50 Hz. Sync pulses, which are injected at the base of Q1 through T1, synchronize the frequency with that at the transmitter. In practice, the vertical hold control is varied until the oscillator frequency locks with the incoming sync pulses to provide a steady picture.

19.5 TRANSFORMERLESS OUTPUT CIRCUIT
The high wattage transistor needed for the output stage can be replaced by two medium power transistors in a suitable push-pull configuration as is the common practice in audio amplifiers. One such method uses p-n-p and n-p-n transistors in a complementary symmetry as illustrated by the basic circuit of Fig. 19.12. The output impedance provided by the complementary pair is quite low and so there is no need for an output transformer. This results in higher efficiency and improved performance because the output transformer is often heavy, expensive and introduces frequency distortion.
Input voltage Driver Q2 Yoke current From vertical oscillator Q3 D1 + 24 V Q3 Vertical power amplifier Deflection coils 24 V 2.5 Amp (P–P)

Feedback

– 24 V

Fig. 19.12. Transformerless vertical output amplifier.

The output transistors Q2 and Q3 operate class B and are alternately driven into conduction by a common trapezoidal input signal. When Q2 is on and Q3 is off, current flows through the yoke from the positive 24 V supply. On alternate half of the input signal when Q3 is on and Q2 is off, current flows in the opposite direction from the negative 24 V supply. This amounts to an ac current flow through the yoke. Diode D1 is forward biased and voltage drop across it provides suitable bias to Q2 and Q3 thereby preventing any crossover distortion. The conduction of D1 also ties the bases of Q2 and Q3 allowing signal output from the driver (Q1) to feed both of them simultaneously.

Oscillator A explained in the previous chapter, transistors Q1 and Q2 operate as complementary symmetry oscillator and provide output at 50 Hz. Hold control forms part of this circuit. Its output is waveshaped by transistor Q3 and the associated capacitor, to provide a sawtooth voltage. The wave shaping circuit has provision for varying amplitude of the sawtooth voltage and this is the height control. Differential Amplifier Transistors Q4, Q5 constitute a differential amplifier with two ac and two dc inputs. One ac input is the sawtooth voltage developed by Q3 and the other is the feedback signal from vertical windings of the yoke. Such an arrangement acts to eliminate, by negative feedback, any distortion introduced by the output stages, ensuring a linear vertical sweep. It also makes the use of linearity control and consequent adjustments unnecessary. The single ended output of the differential amplifier (transistors Q4, Q5) is direct coupled to the output stage. The two dc inputs to the differential amplifier determine the output level which in turn controls vertical (dc) centering of the beam. One of the two dc inputs is a clamped dc voltage that holds the base of Q4 at near zero voltage. However, the second dc input (at the base of Q5) is made dependent by feedback on the dc output voltage at the point where deflection windings are connected. The reference dc voltage is obtained from the two dc supplies through a potentiometer which is initially varied to obtain correct centering of the beam. Driver, Phase Invertor and Output Amplifier The driver transistor Q6 couples output from the differential amplifier direct to the output transistor Q8 and via phase invertor Q7 to the other output transistor Q9. Transistors Q8 and

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Q9 are connected as quasi-complementary pair and conduct alternately to provide the total vertical scan. DC Supply The power supply block has two rectifiers D1 and D2 with associated filter capacitors C1 and C2. It provides both positive and negative dc supply sources for the various blocks. The input for dc supplies is derived from the receiver’s horizontal scanning circuit. Any change in horizontal scanning will be reflected as a change in dc voltage for the vertical module thus maintaining a fixed aspect ratio of the picture.

19.7 THE MILLER DEFLECTION CIRCUIT
Linearity requirements of colour receivers are very rigorous and need a careful design of the vertical stage. While circuits described in the previous two sections are used both in monochrome and colour receivers, a new circuit known as Miller sweep circuit has found wide acceptance in recent colour receiver designs. This circuit is capable of generating a sawtooth voltage of exceptionally good linearity. As a result, no linearity control is necessary. The basic sweep generation circuit makes use of the Miller effect to enormously increase effective size of the sawtooth forming capacitor. The simulation of such a large capacitance in the ramp forming circuit can be illustrated by the circuit arrangement shown in Fig. 19.14. The basic amplifier used in this circuit has a very high gain (|A| ≈ 106), inverted output and a very high input impedance. Because of large gain and heavy feedback provided by capacitor CM the net voltage vi at the input terminals of the amplifier is nearly equal to zero and hardly any current flows across the terminals marked X and Z. Thus X and Z are virtually a short from the circuit analysis point of view.
CM vCM i R S1 V vi Z Ceff i X Gain » 10
6

Y v0 Z

Fig. 19.14 Basic Miller sweep circuit.

V V – vi ≈ , (vi ≈ 0). Since R R practically no current flows between X and Z, the current i completes its path through CM and
The sawtooth forming current i that flows through R = charges it. Therefore i is also equal to

vi – v0 – v0 ≈ , where XC is the reactance of capacitor M X CM X CM

CM. The current i being equal to V/R is independent of v0, the effective voltage across CM and

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so remains constant to charge the capacitor thus developing a linear ramp across it. The admittance across the terminals X, Z =

i –v0 1 = × vi vi X CM

F since i = –v I GH J X K
0 CM 0 i

= 2πfACM because

Fv GH v

= – A and X CM =

1 2πfCM

I JK

Thus effective value of the capacitance in series with R is gain times the value of CM. With |A| ≈ 106, the effective capacitance (Ceff) will appear to be one million times the external sawtooth forming capacitor CM. Consequently the charging time constant of the sawtooth forming RC circuit becomes very large and only a small portion of the RC charging curve is used to form trace portion of the sawtooth. Switch S1 (see Fig. 19.14) can be opened and closed at appropriate moments to generate a sawtooth voltage at the output. In actual deflection circuits a transistor switch which is a part of the oscillator circuit controls the charge and discharge periods. In this manner a sawtooth of excellent linearity is obtained which is later amplified to the desired level. Note that the ramp voltage that appears at the output terminals of the amplifier increases in negative direction because of 180° phase reversal between input and output of the basic amplifier. Miller Vertical Sweep Circuit Figure 19.15 is a simplified schematic diagram of a Miller vertical defection circuit. The circuit employs a complimentary symmetry power amplifier to drive the yoke. The waveshaping circuit has a high gain amplifier; a driver circuit and an oscillator switching transistor. Because of overall positive feedback the circuit behaves like a relaxation type of free running oscillator. Transistor Q1 switches at the vertical frequency rate to provide trace and retrace periods. The circuit employs four feedback paths to accomplish all the functions. Feedback path A effectively increases the value of Miller capacitor CM thus ensuring excellent sweep linearity. Feedback path B provides positive feedback needed to convert the high gain amplifier into a sawtooth generating multivibrator. CM charges through Q1 but discharges through Q2. The adjustments of this part of the circuit include height control, and hold control (frequency control) but there is no linearity control. Feedback path C is used to improve frequency stability of the oscillator circuit. While the Miller sweep circuit provides a very linear sawtooth but the curvature of the tube face plate and its rectangular construction (pincushion effect) necessitate modification of the current that flows through the deflection windings. Therefore feedback path D is used to introduce ‘S’ shaping or correction to the yoke current to compensate for the stretch at the top and bottom of the raster. ‘S’ compensation is also achieved by generating a parabola of current and combining it with the sweep output. The modified yoke current then provides linear deflection despite curvature of the face plate. In addition, a pincushion correction circuit is connected in series with the yoke to cause necessary change in the current waveshape.

19.8 INTEGRATED CIRCUIT FOR THE VERTICAL SYSTEM
Several dedicated ICs are now available which perform all the functions necessary for obtaining linear vertical deflection of the beam. One such IC is BEL 1044, the schematic diagram of which is given in Fig. 19.16. It is a silicon monolithic integrated circuit containing all stages necessary for vertical deflection in black and white television receivers. It performs the functions of a linear sawtooth generator, a geometric ‘S’ correction circuit, a flyback booster and an output amplifier. This integrated circuit requires only a few external components to complete the total vertical deflection system. Such components and controls are suitably labelled in the figure. The BEL 1044 is supplied in a 16-lead dual-in-line plastic package, with an integral bent-down wing-tab heat sink, intended for direct printed circuit board insertion. The device is capable of supplying deflection current up to 1.5 A p-p and positive blanking pulses of 20 V amplitude. The block which needs special mention is the flyback booster circuit. This ensures development of spiked trapezoidal voltage of amplitude almost equal to the supply voltage. The charge stored in the 100 µF capacitor supplies necessary power to the circuit during sharp flyback periods. Note that the diode (BY 125) stays reverse biased during flyback intervals.

Review Questions
1. A well designed vertical scanning section of the television receiver must include the following provisions: (i) adjustment for vertical linearity, (ii) means for suppressing undesired oscillations in the yoke current, (iii) constant height of the raster, (iv) correct aspect ratio. Explain either with the help of separate circuits or by drawing complete circuit of a typical vertical deflection section, the techniques employed to meet the above requirements. 2. Draw simplified circuit diagram of a vertical deflection amplifier employing transistors and explain its operation. Indicate how positive retrace pulse can be obtained from the output transformer of the circuit drawn by you. The circuit of a commonly used vertical deflection section, that employs tubes is shown in Fig. 19.9. Explain in brief, the operation of this circuit and the need for various controls provided in different parts of the circuit. The circuit of a typical multivibrator controlled vertical output stage that employs a somewhat different method for generating sweep voltage and linearity control is shown in Fig. 19.10. Describe briefly all the essential features of this deflection circuit and also explain how vertical rolling of the picture is prevented.

3.

4.

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Specially designed modules for the vertical stage are often used in TV receivers. Draw schematic block diagram of one such module and explain the operation of each section. Why is the dc supply source for the module often made dependent on output of the horizontal scanning circuit ? What do you understand by Miller effect ? Explain with a suitable circuit diagram how a large capacitance is simulated at the input of a high gain inverting amplifier for developing a linear sweep. Draw schematic diagram of a typical Miller vertical deflection circuit and explain how a highly linear sweep output is developed by employing suitable positive and negative feedbacks. Draw functional diagram of the vertical system IC-BEL 1044 and explain how various sections of the IC function to develop deflection sweep output.

6.

7. 8.

20
Horizontal Deflection Circuits

20
Horizontal Deflection Circuits
The horizontal output stage is a power amplifier which feeds horizontal deflection windings in the yoke to deflect the beam across the width of the picture tube screen. Though the function of this stage is similar to the vertical output stage, its design and operation are very much different on account of the higher deflection frequency, very short retrace period and higher efficiency. Since the horizontal output load is primarily inductive*, the deflection current induces very high voltage pulses in the output transformer while falling sharply during the retrace period. This energy which is in the form of high voltage pulses is rectified to produce a high voltage dc source (EHT) for the final anodes of the picture tube. Furthermore, the energy associated with the high voltage pulses tends to setup high frequency oscillations in the output circuit. To suppress this high frequency ringing, a diode damper is used instead of resistors employed in the vertical system. The diode while conducting, charges a capacitor, the voltage across which adds to the existing B+ supply to produce a higher dc source voltage commonly known as boosted B+ supply. In this manner, part of the energy from the deflection coils is recycled to develop a voltage source for the horizontal stage and other circuits. Another interesting feature of the horizontal output circuit is that the unidirectional damper diode current which also flows through the deflection coils is used to complete a part of the horizontal trace. This results in improved efficiency because the amplifier need not supply deflection current for the entire cycle.

20.1 HORIZONTAL OUTPUT STAGE
A schematic block diagram of the horizontal output stage is shown in Fig. 20.1. The main sections of the stage are (a) output amplifier V1, (b) output transformer T1 (commonly known as flyback transformer), (c) damper diode D1, (d) high voltage rectifier D2 and (e) deflection windings in the yoke. The amplifier operates with self-bias and is designed to perform as a switch which closer for a fixed period during each horizontal deflection cycle to supply power to the output transformer. The transformer’s main functions are to provide impedance match between the output stage and deflection windings and to act as a step-up high voltage transformer for D2, the EHT
* L/R for the vertical deflection yoke is about three times the L/R for the horizontal deflection yoke, but the horizontal scanning frequency is about 300 times the vertical scanning frequency. That is why the horizontal scanning load looks essentially inductive whereas the vertical scanning load looks effectively resistive.

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rectifier. The damper diode D1 conducts soon after the flyback stroke to suppress shock excited oscillations. While doing so the damper also causes horizontal scan of the beam on the left-half of the raster.
High voltage pulses Horizontal output (Flyback) transformer Horz drive 15625 Hz D2 High voltage rectifier

10 to 16 kV (EHT) To anode of picture tube

V1 Horz output amplifier B++

T1 D1 Damper diode Horizontal deflection coils in the yoke

+ CB B+

Fig. 20.1. Block diagram of the horizontal output stage.

20.2 EQUIVALENT CIRCUIT
The various functions which the horizontal stage performs make its design and operation quite complex. Therefore, it is convenient to first consider a basic equivalent circuit of the stage to help understand its functions and sequence of operations.
S1 S2 V R C r Electron beam Picture tube screen Amplifier circuit Damper circuit Deflection winding L Raster

Fig. 20.2. Simplified equivalent circuit of the horizontal output stage. With no excitation, the electron beam is in the centre of the picture tube screen.

The circuit shown in Fig. 20.2 is a simplified equivalent circuit of the deflection part of the horizontal output stage where the amplifier* has been simulated by a constant voltage
* For a typical horizontal deflection coil L = 2 mH, r = 4 Ω, I (p-p) = 3 A, trace period (ts) = 54 µS and retrace or flyback time (tf) = 10 µS. ∴ VL = L

source V together with switch S1 to control the flow of current to the deflection coils. The inductor L and the small series resistance r represent lumped inductance and resistance of the flyback transformer and deflection windings. The capacitor C in parallel with this circuit accounts for distributed and stray capacitance of the output stage. The damper circuit is represented by switch S2 and a resistance R in series with it. The switch S2 represents damper diode D1 which when conducts charges CB through R. The sequence of operations which cause deflection of the beam across the screen are illustrated in Fig. 20.3 by various circuits and associated waveforms. The location of the beam on the screen is also shown in each case. The direction of beam deflection has been arbitrarily chosen to be from left to right for the direction of current flow from the voltage source. When switches S1, S2 are open, i.e., when both the amplifier and damper diode are not conducting, no current flows through L and r (see Fig. 20.2), and the beam is at the centre of the screen. At t0 (Fig. 20.3(a)) when S1 closes (amplifier is turned on) current iL flows and rises linearly to deflect the beam towards right side of the raster. The beam reaches the edge of the screen when iL attains a value equal to I at instant t1. Since the coil resistance is very small, the voltage vL′ across the coil iL r + L

FG H

di dt

IJ is nearly constant during this period. Its polarity is K

positive and magnitude small because of the relatively slow rate of rise of current. At instant t1, the sharp negative spike of the input signal turns off the amplifier, i.e., switch S1, opens. With S1, S2 both open (Fig. 20.3 (b)) the C-L-r circuit gets into free oscillations with a period T, very nearly equal to 2π LC seconds (since r is small). The initial conditions are such that iL continues to grow further until vc = 0 (t = t2). The entire circuit energy,
1 2

L (Imax)2 is now in the

inductance. T/4 seconds later (t = t3) this energy transfers to C (less a very small loss in r) so that
1 2

L (Imax)2 =

1 2

C(vc max)2 or vc max = Imax

L . The capacitance C, being only stray capacitances C

of the circuit, is very small (a few hundred pico-farads atmost). Therefore V*C max is usually very high, of the order of a few thousands of volts. After another T/4 seconds (t = t4), once more VC = 0 and iL = – Imax. In fact the current varies in a co**-sinusoidal manner from its positive peak to its negative peak in a time equal to T/2. If the switches (S1, S2) remained open this process of energy exchange would go on for many cycles, the successive values of VC max and Imax decaying slowly until the whole energy
1 * 2 L (Imax)2 = 1 2

Fig. 20.3. Sequence of operations during scanning of one horizontal line. (a) amplifier turns on but damper is open (b) amplifier turns off and damper continues to be open (c) amplifier continues to be off but damper diode turns on.

However, continued oscillations beyond the first half-cycle are not desirable because the continuing oscillatory current in the coil will shift the beam back and forth at the left side of the raster instead of allowing it to trace the next horizontal line at its proper place. Therefore, in order to suppress free oscillations beyond its first half cycle, switch S2 closes at instant t4 to dissipate the energy stored in the L, C, r circuit. In fact polarity of the oscillatory voltage changes at t4 to become positive and thus forward biases the damper diode D1 (closes switch S2). With R shunted across L, C, the combination becomes non-oscillatory and the current iL and voltage vL then decrease to zero with a time constant approximately equal to L/R where R represents the resistance of the boosted B+ circuit in series with S2. The resulting current and voltage waveforms are shown in Fig. 20.3 (c). L and C are so chosen that the periods of self-oscillations T is twice the horizontal retrace time. Thus as the coil current decreases from positive maximum to zero and reverses to attain its maximum negative value in time T/2, the beam gets deflected to the left edge of the raster

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to complete a fast retrace. The resonant circuit is set at about 60 KHz to provide the desired flyback time of about 8 µS

F 10 GH 2 × 6 × 10
6

4

= 8 µS .

I JK

Further, the impedance of the B++ circuit is so set, or in the circuit under consideration R is so chosen that the current decays to zero in time (t5 – t4) which is approximately equal to half the trace period. The decaying current from its maximum negative value to zero deflects the beam to the centre of the screen without any supply of energy from the voltage source. This amounts to recovery of energy, because if not utilized this would not only be wasted but also cause undesired oscillations spoiling the raster. The switch S1 (amplifier) closes at t5 to again connect the voltage source to the LC circuit and just then S2 (damper) opens to cut-off the damping circuit. The current then smoothly rises to take over deflection of the beam to the right of the raster. This is illustrated in Fig. 20.3 (c). The cycle described above repeats line after line to trace the entire raster.

20.3 HORIZONTAL AMPLIFIER CONFIGURATIONS
Horizontal output amplifiers may employ (i) vacuum tubes, (ii) transistors and (iii) silicon controlled rectifiers. While the end result is same, the mode of operation is somewhat different in each case. Therefore, horizontal output stages employing different devices are described separately. It may also be noted that deflection circuits in colour receivers differ somewhat from corresponding circuits in black and white receivers. However, initially, the discussion is confined to monochrome receivers only.

20.4 VACUUM TUBE HORIZONTAL DEFLECTION CIRCUIT
Fig. 20.4 is the circuit diagram of an earlier version of horizontal output stage used in monochrome receivers. It employs an autotransformer between the amplifier and deflection coils. Each section of the stage is described separately. (a) Output Amplifier The amplifier employs a beam power tube having high voltage and power ratings. It is driven by a trapezoidal voltage of approximately 80 V (P-P) which is obtained from the horizontal deflection oscillator. In some circuits a sawtooth voltage that has little or no negative going spike is used to drive the amplifier. In any case, on application of the input signal, grid current flows to develop self-bias. The time constant of the coupling network C1, R1 is so chosen that for approximately half of the cycle the input signal drives the grid voltage less negative than cut-off. This results in a linear rise of plate current (see Fig. 20.4) to cause part deflection of the electron beam. With no input drive present, the bias would be zero because there is no provision for self-bias in the cathode circuit. Therefore, the output tube (V1) should not be operated without grid drive because in the absence of any protective bias the tube will draw excessive current. A slow-blow fractional ampere fuse ‘F ’ is provided in the cathode circuit to protect the tube and transformer against excessive current. The screen grid has the usual decoupling network and a provision to control screen grid voltage. The variation of R2 changes

(b) Output Transformer The circuit employs an autotransformer because of its higher efficiency as compared to one having isolated primary and secondary windings. It is wound on a ferrite core to minimize losses. As shown in Fig. 20.5(a) a small air-gap is provided between the ‘U’ and ‘I ’ sections of the core to prevent magnetic saturation due to heavy plate current that flows through a section of the winding. The EHT winding and other sections of the autotransformer winding are wound on opposite limbs of the core to obtain higher leakage inductance necessary for third harmonic tuning. In Fig. 20.4 the winding between 1 and 4 is the primary (Lp) for plate current of the amplifier. This includes L1, L2, L3 and has an inductance of about 100 mH with a winding resistance of about 20 ohms. Winding L1 is the secondary, to step-down voltage for the deflection coils. Since there is no isolated secondary winding for polarity inversion, all the taps on the auto-transformer have voltages of the same polarity as the plate voltage on the amplifier. The drop in plate current induces voltage of positive polarity across the plate coil for retrace. The

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winding L4 at the top steps up the primary voltage to supply ac input to the high voltage rectifier. For a vacuum tube rectifier, its filament power is taken through a small winding L5 on the same transformer L4 is a separate winding with a large number of turns of fine wire. It has an inductance of the order of 650 mH. The amount of step-up is limited by the fact that any increase in inductance reduces resonant frequency of the output circuit.

Beam deflection. As shown in the waveforms drawn along Fig. 20.4, plate current starts flowing just when the drive signal crosses cut-off bias of the tube. The sawtooth rise of current in the primary circuit induces a negative going rectangular voltage across the secondary winding. This causes a flow of sawtooth current in the deflection windings from terminal B to A to deflect the beam from centre to right side of the raster. Note that the direction of current flow in the yoke windings is opposite to that assumed in the equivalent circuit. However, this does not interfere with the deflection of beam towards the left side of screen because the yoke is so wound that the resulting magnetic field has the correct polarity for causing deflection towards right edge of the raster. The deflection current is alternating in nature and is coupled to the yoke through capacitor C2 which blocks the flow of any dc in the deflection windings. This (dc) if allowed will shift the beam from its central location. In some circuits where the blocking capacitor is not used, other techniques are used to bring the beam back to the centre. Horizontal retrace. Figure 20.6 illustrates input voltage, deflection current and yoke voltage waveforms. As shown, the grid voltage drops sharply to a value much below cut-off soon after the peak of the sawtooth part of the drive voltage is reached. This cuts off the output tube instantly and V1 ceases to sustain current in the output circuit. The tube remains in cutoff until the grid voltage again rises sharply to permit conduction during the next cycle. The inductance of the output transformer and deflection coils together with their self capacitance act as an L, C circuit that can oscillate at its resonance frequency when excited from a suitable source of energy. When plate current drops to zero, the sudden collapse of magnetic field in the deflection circuit generates a high value of induced voltage (typically 6 KV). The energy which thus becomes available in the deflection circuit sets the L, C circuit into ringing at its natural

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resonant frequency. As explained earlier with the help of an equivalent circuit, the first half cycle of free oscillations results in a quick flyback of the electron beam. It is often necessary to add an additional capacitor (C3 in Fig. 20.4) to make the resonant frequency nearly equal to 60 KHz for obtaining the desired retrace period.
Raster Horz amplifier on Retrace time (D2 on, D1 and V1 off) Path of beam deflection

Damper on

V1 on Sawtooth current I through yoke 0 I Transition point Damper on Trace vg 0 Input grid voltage Retrace t2×I Trace

(a)

Next cycle

t Cut-off

Bias (–20 V) HV rectifier ‘off’ HV rectifier ‘on’ v 6 kV (b)

One half-cycle of undamped oscillations Plate voltage (V1)

0

Damper on (c)

Oscillations damped by damper current t

Fig. 20.6. Voltage and current relationships in the horizontal output stage.

During retrace the polarity of induced voltage across D1 is such that it remains cut-off along with the output tube. This is necessary because any flow of current either in V1 or D1 will induce a counter voltage that will oppose the current to fall at a fast rate and thus lengthen the retrace period. Thus the first half-cycle of oscillations continues undamped for a fast flyback. If

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the oscillations are not suppressed after this, they would cause ripples on the scanning current waveform. This results in white vertical bars on the left side of the raster. The bars are on the left side of the raster because the oscillations occur immediately after flyback and these are white because the electron beam scans these areas several times during each horizontal line traced. (c) Damper Diode As soon as the polarity of the oscillatory voltage (Fig. 20.6 (c)) reverses, the damper diode (D1) is forward biased and it conducts to dissipate energy stored in the deflection circuit. The diode, while suppressing undesired self-oscillations after the first half cycle, charges CB and causes deflection of the beam (see Fig. 20.6) from the left edge of the screen to its centre. This is best explained by considering the two effects separately. Boosted B+ supply. The part of circuit in Fig. 20.4 which is associated with the production of B+ boost is redrawn in Fig. 20.7 (a). Initially when the flyback transformer is not energized the damper diode conducts to supply B+ voltage to the plate of V1. However, when the circuit is

functioning, the diode stays reverse biased most of the time because of high voltage VCB

d i

across capacitor CB. Immediately after retrace when induced voltage reverses polarity its magnitude is higher than VCB and so the damper diode is turned on. A part of the damper current completes its path through the boost capacitor CB and charges it to a still higher potential. In practice the voltage that develops across the capacitor varies between 200 to 600 V. In the circuit under consideration the damper has been tapped up on the autotransformer winding and the voltage that builds up across CB is about 500 volts. As illustrated in Fig. 20.7 (a) this voltage is in series with the B+ supply. Therefore the potential difference between the positive side of CB and chassis ground is around 800 V and is knowns as boosted B+ supply. This source is used as a dc supply for the horizontal output amplifier and several other circuits.

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While supplying current the booster capacitor partially discharges during the time the damper is off. However, this is replenished when the damper conducts during the next cycle. The partial charge and discharge of capacitor CB at the rate of horizontal scan results in a ripple voltage at this frequency. To remove this ripple a filter circuit, (R6, C6 in Fig. 20.4) is often provided. Figure 20.7 (b) is a boosted B+ circuit of a horizontal deflection stage employing an output transformer with isolated primary and secondary windings. While the direction of induced voltage can be independently decided but in the circuit under consideration the polarity of the induced voltage in the secondary is opposite to that in the primary. As such, the anode of the damper diode is connected to the upper end of the secondary winding and B+ supply is connected at its lower end. The diode, initially conducts to charge CB to the B+ voltage. However, when the transformer is energized the ac voltage across LS, soon after flyback, forward biases D1 which conducts to charge the boost capacitor to a still higher voltage. Thus the entire boosted B+ voltage appears across the boost capacitor. The generation of B++ voltage explains the principle of energy recovery, where a major part of the energy stored in the coils at maximum deflection is recycled. This improves the efficiency of the horizontal output circuit. Reaction Scanning. The damper diode D1 conducts and charges capacitor CB as explained above. As shown in Fig. 20.8, a part of the diode current also flows through the deflection windings. As CB charges, the damper diode current decreases. Finally it becomes zero when the voltage across CB biases the damper diode out of conduction. The diode current that flows through the yoke coils is made to decrease linearly by careful circuit design and this deflects the beam towards the centre. Note that direction of current flow is opposite to that due to plate current of V1. This is illustrated in Fig. 20.6 (a).
To plate of V1

4

3 T1 2 K 1d B++ 1 CB P B+ C2 1d D1 Deflection coils

Fig. 20.8. Reaction scanning by the damper current.

The circuit is so designed that before the damper current declines to zero, output tube V1 starts conducting and the resulting current then completes the trace to the right. As shown

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in the waveform (Fig. 20.6 (a)) the amplifier and damper currents combine to cause horizontal deflection of the picture tube beam. In fact the declining diode current and the rising plate current add to provide a sawtooth current for linear horizontal deflection. This is known as ‘Reaction Scanning’. (d) Generation of High Voltage During flyback periods the magnitude of induced voltage across the primary (windings L1 to L3 in Fig. 20.4) coil is very high. In typical vacuum tube circuits it varies between 3 to 6 KV. By placing a suitably wound coil (L4) in the magnetic circuit and connecting it to the primary winding in a series aiding fashion, an additional voltage of about 6 to 8 KV is developed by autotransformer action. The corresponding circuit diagram and associated waveshapes are shown in Fig. 20.9. This circuit arrangement results in a total voltage of 9 to 14 KV across two ends of the combined winding. Further, by suitably controlling the distributed parameters of the primary and secondary circuits and utilizing the technique of third harmonic tuning, (discussed in a subsequent section), the total voltage can be enhanced by about 10 to 15 percent. Thus a pulse of the order of 11 to 16 KV is generated. This is fed to the high voltage rectifier D2 which conducts to charge the filter capacitor to provide EHT supply. It may be noted that the high voltage rectifier conducts for a short time during the flyback period because D2 is reverse biased at other times on account of 11 to 16 KV dc at its cathode (filament). The conduction of D2 partially dicharges CB but it is soon made up when D1 conducts. Note that while D2 conducts both V1 and D1 remain cut-off to avoid any loading.
5 T1 L4 4 L3 3 L2 2 L1 To other load circuits 1 + VCB + B+ supply (a) (b) 0 t To deflection windings L5 500 pF To damper circuit kv 22.5 15 (EHT) 7.5 15625 Hz ripple + C4 R4 HV rectifier D2 + EHT 12 to 16 kV To picture tube anode 0 – Pulse frequency = 15625 Hz t D2 plate voltage (ac)

To amplifier circuit

Fig. 20.9. High voltage (EHT) generation (a) circuit (b) waveshapes.

(e) Monochrome Yoke Since the current in the scanning coils deflects the electron beam, the yoke is rated in terms of deflection angle in addition to other essential parameters necessary for generating the required

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magnetic field intensity. Small screen tubes employ yokes having a deflection angle of 90°. However, the broad screen tubes are provided with 114° deflection angle yokes. Deflection coils suitable for 31 to 61 cm picture tubes are available. The deflection yoke consists of two sets of coils, one for horizontal deflection and the other for vertical deflection. The coils are independently wound in two halves. These are placed on the yoke at right angles to each other in alternate quadrature segments. The yoke assembly is slipped on to the neck of the picture tube and is oriented to produce a raster parallel to the natural X and Y axes. In earlier deflection coils additional capacitors were used to balance the distributed capacitance of the two halves and their capacitance with respect to chassis earth. This is necessary if each half coil is to produce equal retrace period for eliminating the possibility of multiple harmonics. However, in the latest designs of scanning coils lumped components are no longer necessary. The two halves of each coils are so wound that they provide the exact desired balance necessary for undistorted scanning. The ratio of inductance to resistance (L/R = time constant) is kept low to achieve minimum permissible limits of distortion. The flared portion of the coils may not exactly match the slope on the neck of the picture tube and, therefore, a pull-back of about 3 mm is provided to permit adjustment of the scanning coils. Circuit efficiency. The peak energy stored in the inductance L of the deflecting coils is
1 2

L(Imax)2 where Imax is the peak current. If the damper diode technique is not used, this energy
1 8 2 L I y fh where Iy is the

is dissipated in a resistor during each cycle. The power lost is equal to

peak-to-peak amplitude of the deflection current and fh is the horizontal scanning frequency. For a 61 cm picture tube yoke L = 2 mH, and Iy = 3 amp. The power consumed is thus about 35 watts. This value is about 25 percent of the total power taken by the television receiver. By the simple technique of replacing the damper resistor R by an efficiency diode the power loss is cut to about one quarter of this value for the same deflection. As illustrated in the waveforms drawn in Fig. 20.6 (a) the total sweep corresponds to a current of 2I and yet the peak energy stored in the inductor is
1 2

LI2 and not

1 2

L(2I)2. This observation verifies the statement made

above that the power loss in the magnetic coils can be cut to one quarter of its previous value by the principle of energy recovery. The efficiency of the stage is further improved by storing a part of the excess energy in a capacitor (CB) and feeding it to auxiliary circuits in the receiver. The ingenious way the flyback pulses are utilized, to produce EHT supply, results in overall economy, because, if the same voltage were obtained from the supply mains, it would require a very bulky and expensive step-up transformer and filters. The use of first half-cycle of the sinusoidal oscillation is yet another important feature of electronic circuit design which results in such an inexpensive method of obtaining flyback period of a few micro-seconds. All these innovations have resulted in very efficient horizontal output circuits that form part of all present day television receivers.

20.5 SEQUENCE OF OPERATIONS
It is now obvious that efficient design of the horizontal output circuit and the various auxiliary functions that it performs make its operation quite complex. It is thus desirable to summarize

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the sequence of operations for a clear understanding of the functioning of horizontal output stage. These are: (a) Self-bias is used to control conduction of the output tube and is so adjusted that the tube conducts only during a part of the input voltage swing to supply sawtooth current. The rising plate current provides a little more than half of the total rater scanning on the right side of the screen. The damper diode remains reverse biased during this interval. The high voltage rectifier also does not conduct because of the large dc voltage at its cathode. (b) The output tube suddenly goes to cut-off because of sharp fall of the grid signal voltage. This shock-excites the secondary circuit and it sets into oscillations. The onset of oscillations induces high voltage pulses which make the anode of the high voltage rectifier highly positive. It then conducts during the short pulse periods to provide EHT. Note that the damper diode and output tube continue to be in cut-off during this interval. (c) The first half-cycle of oscillation is allowed to continue undamped for a fast flyback that brings the picture tube beam to the left side of the raster. During this half-cycle the oscillatory voltage has a polarity which makes the damper diode plate negative. Therefore, both the output tube and damper remain in cut-off while the retrace takes place. Note that the high voltage rectifier goes off soon after the sharp high voltage pulse occurs. (d) After the first half-cycle of oscillation is over the voltage wave polarity reverses to make the damper diode plate positive. It then conducts (i) to damp oscillations, (ii) to charge the boost capacitor to provide boosted B+ supply and (iii) to deflect the beam towards the centre of the screen by ‘reaction scanning’. The damper capacitor (CB) which is in series with the damper diode continues to build up charge and finally biases the damper out of conduction. (e) Before the damper current declines to zero the output tube once again starts conducting to continue the trace to right side of the raster.

20.6 HORIZONTAL AMPLIFIER CONTROLS
A horizontal output amplifier of a monochrome receiver may have a drive control, a width control and a linearity or efficiency control. (a) Drive Control This control can be used to adjust the width of the raster by varying peak-to-peak amplitude of the drive voltage to the amplifier. A capacitive or resistive potentiometer is provided in the grid circuit to control the magnitude of input voltage. In modern receivers such a control is not provided and instead width of the picture is adjusted by controlling current in the deflection coils. However, optimum drive voltage is obtained by adjusting output voltage of the horizontal oscillator. (b) Width Control Figure 20.10 (a) illustrates five possible methods of controlling width of the picture. Each control has been suitably numbered for convenience of explanation.

(i) This control operates by varying screen grid voltage of the amplifier tube. Any increase in screen voltage causes an increase in plate current. This means an increase in deflection energy which is converted into an increase in horizontal deflection. Reducing the screen voltage will have the opposite effect. (ii) In this method a small variable coil (10 to 40 mH) is placed across part of the flyback transformer. It acts as an additional load on the horizontal output stage and absorbs some

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energy. A reduction in its inductance causes higher current flow through the coil and thus less current flows through the yoke. This results in reduction of picture width. (iii) A variation of the above method is to place a coil in series with the deflection windings. In this case, the width control limits the current that can flow through the yoke. Shorting it will result in maximum deflection. Note that using a coil rather than a resistor as a width control has the advantage of controlling the yoke current without any appreciable loss of power. (iv) In this method of width control, a small capacitor C1 (50 to 150 pF) is connected across the deflection coils. This lowers the resonance frequency of self oscillations. A lower frequency increases retrace time which in turn causes the high voltage to decrease. Any reduction in the picture tube anode voltage lowers the beam velocity. Thus the beam stays for a longer period under the influence of the deflection field and more deflection is caused. The value of C1 is chosen for optimum deflection. (v) Another method of width control, part circuit of which is shown in Fig. 20.10 (a) by dotted chain lines, accrues from the EHT and raster width stabilization circuit which is often provided in modern receivers. A fraction of the high voltage pulse (see Fig. 20.11) which develops in the flyback transformer is fed to a series circuit consisting of a VDR and a potentiometer (R2). The VDR rectifies the unsymmetrical pulse voltage. The resulting current develops voltage drops across it and resistor R2. The amplifier input circuit is modified to include these voltage drops in the self-bias circuit. The negative voltage drop across the VDR adds to the self-bias voltage and counteracts any increase in the pulse voltage. However, drop across the potentiometer which is in opposition to the self-bias voltage can be varied for optimum output circuit current which is turn controls the width of the raster. Thus R2 can be used as a width control. The stabilizing action of the circuit is explained in Section 20.9. (c) Linearity Control Some non-linearity can be caused in the centre of the screen where the deflecting damper current drops and tube current takes over. As shown in Fig. 20.10 (a), one method of correcting this consists of a variable inductor connected in series with the damper diode. The coil L1 forms a parallel tuned circuit with C2 to be resonant at the horizontal scanning frequency. The switch like operation of the damper (D1) shock excites the tuned circuit into ringing at the same frequency. The self-oscillatory voltage at the plate of D1 is modified when damper conducts. It attains the shape of a parabolic ripple voltage of magnitude 30 to 50 volts. The phase and timing of this ac (15625 Hz) voltage controls the conduction time of the damper. Adjusting the coil inductance changes the phase of the anode voltage which in turn changes damper conduction. Thus by properly adjusting the coil inductance a smooth transition from damper current to amplifier current can be obtained so as to have linear deflection of the beam from left edge of the screen to its right side end. In fact when the coil is properly adjusted for linear deflection, dc current of the horizontal output stage will be minimum. This improves efficiency of the stage. Therefore, this coil is also known as efficiency coil and the control as efficiency control. Yoke coils of present day monochrome receivers are so well designed that they do not need any linearity adjustment. However, in colour receivers, on account of other complexities adjustment of the efficiency coil is very important and such a control is always provided.

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Nonlinearity is also caused due to resistance of deflection coils and on-resistance of the switching device which is not zero. During scans, the voltage available to the yoke inductance decreases as the deflection current rises because of increasing voltage drops across these resistances. This causes appreciable nonlinearity and a correction becomes necessary. This is provided by saturable core reactor (LC) connected in series with the deflection coils (see Fig. 20.11). The saturable reactor (also called linearity coil) consists of a small coil wound on a ferrite core. When the scan current is low, core does not saturate and hence voltage drop across the coil is maximum. However, as the deflection current increases, the ferrite core saturates and drop across it is reduced. Thus the decreasing voltage drop across the saturable reactor compensates for the increasing drop in the deflection windings and the driving device’s forward resistance. This ensures an almost constant voltage source across the coil inductance and a sawtooth current flows to cause linear deflection.
8 R8

20.7 ‘S’ CORRECTION
A linear variation of current in the deflection coils will cause linear displacement of the spot on the picture tube screen if the face plate has a suitable curvature. However, in modern wide angle picture tubes which have nearly a flat screen, a uniform angular motion of the beam will result in nonlinear deflection. As the deflection angle increases the electron beam has to travel farther to reach the face of the picture tube. Thus more displacement occurs for the same angle of deflection. As shown in Fig. 20.10 (b) a deflection angle of 10° is necessary to scan a distance

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d at the centre of the screen, whereas an angle of 8.5° is necessary for the same deflection at the edges. Therefore, to produce linear deflection the rate of change of deflection should decrease as the beam approaches edges of the screen. Hence, in order to produce an undistorted picture display, the rate of change of scanning current must be reduced as the deflection angle increases. This means that nearly an ‘S’ shaped current should flow into the deflection windings. This is illustrated in Fig. 24.10 (c). A capacitor in series with the yoke coils (C3 in Fig. 20.10(a)) provides the required S-correction. A parabolic voltage develops across the capacitor when the sawtooth deflection current tends to flow through it. This modifies the sawtooth current in such a way that a faster rate of deflection occurs while the beam is at central portion of the screen than when it approaches the edges. Thus the effect of a flat screen is counteracted and linear deflection occurs across the entire width of the faster.

20.8 IMPROVED LINE OUTPUT CIRCUIT
Modern horizontal output circuits employ feedback techniques to achieve linear deflection and make the raster size almost independent of mains voltage variations. Figure 20.11 is the schematic diagram of such a circuit where V1 is the output tube, D1 the damper or efficiency diode and D2 the EHT rectifier. It may be noted that the damper diode is in the primary circuit of the autotransformer. Circuit Operation The circuit employes self-bias technique to control conduction and cut-off periods of the output tube. The resistor R3 prevents excessive grid current flow during positive peaks of the trapezoidal drive voltage. In each cycle, when V1 conducts, the linearly rising current in winding 1, 2 develops a rectangular voltage across the secondary (5, 7) winding. The resulting sawtooth current through the yoke coils deflects the beam from centre to right side of the raster. As soon as the grid drive voltage goes negative, tube is cut-off and current flow ceases both in V1 and D1. The standing current in the primary winding shock excites the associated L, C circuit. The resulting variation of the self-oscillatory current from its positive to negative peak during the first half cycle of oscillation deflects the beam to the left side of the raster thus providing a fast flyback. The associated induced voltage keeps the cathode of D1 highly negative during this period and no current flows through V1 and D1 during flyback. As soon as the second half cycle of self oscillation begins, cathode of D1 becomes negative and damper diode conducts via winding 2, 4, capacitor CB, winding 5, 6, ground points E1, E2 and B+ supply. The damper current induces a voltage in winding 5, 6, which circulates a sawtooth current in the yoke to deflect the beam from left to middle of the screen. The current that flows through CB charges it with the polarity marked across it. As soon as potential at the cathode of D1 becomes equal to B+ supply, V1 takes over and conducts again to deflect the beam from middle to right side of the raster. This sequence continues till the boost capacitor CB is charged to the maximum possible value. Once CB is fully charged it becomes the supply source for V1. However, B+ voltage source continues to supply make-up current during each cycle to replenish part energy lost by CB while feeding V1 and other auxiliary circuits.

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The capacitor C5 in the secondary circuit is for ‘S’ correction while the linearity coil LC compensates for any nonlinearity introduced by resistive voltage drops in the output circuit and supply source. Resistors R6 and R7 connected across C5 and LC respectively suppress any self-oscillations when the circuit is shock excited by deflection current flow. The voltage induced in winding 1, 8, during flyback together with the voltages across other sections of the autotransformer becomes the high voltage ac source which is rectified by diode D2 to provide EHT supply. Many line output circuits employ a stack of selenium rectifiers (for example TUS 15) instead of a vacuum tube rectifier. Third Harmonic Tuning In the equivalent circuit shown in Fig. 20.12 (a) L1 is the effective primary inductance and C1 the primary capacitance. The inductance LK represents leakage inductance of the EHT winding when referred to the primary side. Similarly CK is the effective capacitance across LK while C2 accounts for the capacitance between EHT winding and ground. The leakage inductance is deliberately made large so that it resonates with the stray capacitance at about third harmonic (2.8 times) of the flyback frequency. The correct tuning is obtained by varying location of the third harmonic ring (see Fig. 20.5) between the two winding and observing voltage waveshape on a cathode ray oscilloscope. When tuned properly, the phase of the third harmonic voltage cancels part of peak voltage at the plate of V1. At the same time, phase of the third harmonic ringing tends to increase the amplitude of high voltage pulse delivered to the EHT rectifier. The two effects are illustrated in Figs. 20.12(b) and (c). While the reduction in peak voltage by about 10% at the plate of output tube permits the use of a tube having lower voltage breakdown ratings, the enhanced high voltage pulse yield 15 to 20% higher EHT voltage. All this results in increased economy and more efficient operation.
Ck HV rectifier S Lk V L1 C1 C2

Fig. 20.12. EHT winding equivalent circuit and third harmonic tuning. (a) Equivalent circuit with HV (EHT) circuit referred to the primary side (b) Effect of third harmonic tuning on the plate voltage (c) Increased HV pulse amplitude due to in-phase third harmonic voltage.

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20.9 OUTPUT CIRCUIT STABILIZATION
Output tube’s operating conditions influence EHT supply and raster width. This is stabilized by regulating the plate current through a feedback loop from the output circuit. A component of the grid bias voltage is obtained by rectifying a part of the pulse voltage which develops across the output transformer. The feedback circuit in Fig. 20.11 includes a VDR and a potentiometer. A VDR is a semiconductor the resistance of which decreases as the voltage across its terminals increases. It has symmetrical nonlinear V-I characteristics (see Fig. 20.13) and therefore allows equal current flow in either direction if a symmetrical voltage is impressed across it. However, if an unsymmetrical voltage of sufficient amplitude is applied, rectification takes place. This implies that unequal current flows in forward and reverse directions. Thus a net voltage develops across the VDR and its value depends on the magnitude of the impressed voltage. In the circuit under consideration (Fig. 20.11) a direct current flows through VDR and R2 because of dc voltage across the boost capacitor. On this direct component is superimposed a pulsating current which is derived via C3 from the flyback pulses. This is illustrated in Fig. 20.13 where the voltage and current waveshapes have been assumed to be rectangular for the sake of simplicity. As indicated in Fig. 20.13 the working point ‘w’ is determined by the mean value of this current. The voltage across the VDR fluctuates and its mean value is negative. This actually means that voltage drop across the VDR decreases because the magnitude of reverse current is more than the forward current.
I(mA) –1200 –800 –400 6 5 4 3 2 1 V 0 W 1 2 3 4 5 6 –I +V VDR (mean) 0 400 800 1200 R2

VDR

Vg mean

Negative control voltage

I

Fig. 20.13. VDR characteristics and development of control voltage.

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The pulsating negative voltage is filtered by the RC network formed by grid-leak resistor R1 and capacitance of the oscillator drive circuit. Thus an additional dc voltage proportional to the pulse amplitude becomes part of the amplifier grid bias. The potentiometer R2 is set for optimum bias so that both the EHT voltage and raster width are of correct values. It due to any reason the EHT voltage rises, it is obvious that the magnitude of the negative voltage across the VDR will increase. However, this cannot happen in practice because an increase in the negative control voltage reduces EHT pulse voltage on account of reduction in tube plate current. Similarly any decrease in EHT pulse will be counteracted by a decrease of the negative control voltage. Thus any tendency for increase in raster width or EHT voltage which may occur due to mains voltage fluctuations or any other reason is prevented by adding or subtracting a control voltage to the self-bias circuit. Width Control The steady current that flows through the VDR and R2 fixes the threshhold value for proper functioning of the VDR. Since the voltage drops across the VDR and R2 are opposite in polarity, the potentiometer R2 can be varied to control the operating point of V1 for optimum deflection current. Thus R2 acts as a width control. The capacitor C4 is chosen to obtain the correct flyback period (frequency to selfoscillations) and thus the required EHT voltage. The boosted B++ supply which is developed in the line output stage feeds the vertical (frame) output circuit in addition to the first and second anodes of the picture tube. The feedback circuit tries to maintain the boosted B++ supply at a constant value. However, if any small changes occur, they affect the horizontal and vertical output stages equally thus maintaining correct aspect ratio of the reproduced picture. Auxiliary Windings Instead of providing separate windings, AGC and AFC circuits are fed from tappings on the autotransformer. The AFC circuit receives voltage that develops across tap 7 and ground. Similarly the AGC circuit is connected across tap 5 and ground. The connections can be interchanged between taps 5 and 7 depending on the polarity of the pulses needed at the AFC and AGC circuits.

22.10 TRANSISTOR HORIZONTAL OUTPUT CIRCUITS
The two major disadvantages of a vacuum tube horizontal output stage are (i) lot of power is wasted in the tube’s heater circuit and (ii) a vacuum tube has a relatively high plate resistance that necessitates the use of a matching transformer between the tube and yoke coils. In a transistor circuit such problems are not there. However, the output transistor while having a high current rating must be able to switch at fast speeds. This is difficult to obtain in the same transistor and proved to be the major obstacle in transistorizing the horizontal output stage. While transistors are now available which can switch a peak power of 200 watts and have an average power rating of 20 to 50 watts but their breakdown voltage capability VCE(max) and VCB(max) is limited to about 1500 volts.

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The output transformer of a transistor line output stage has a somewhat different role than that used in a corresponding vacuum tube circuit. In tube circuits the output transformer is primarily responsible for obtaining rapid retrace because of its self-oscillations. In addition, it is used for providing impedance match to the yoke windings and high voltage pulses to the EHT rectifier. In transistor circuits, retrace time is determined by the resonant circuit formed by effective inductance of the output circuit and an external capacitor connected in the collector circuit of the transistor. This capacitor is known as the flyback capacitor. The output transformer is instead tuned with its distributed capacitance to about third harmonic of the yoke oscillations. This as explained earlier reduces collector to emitter voltage and increases amplitude of high voltage pulses. Boosted B+ Voltage In transistor circuits the high dc voltage is developed in a different way. While the circuit has a damper diode but there is no boost capacitor. Instead, the B+ needs of transistor circuits are met from taps on the output transformer. The dc pulses from the taps are rectified and filtered to obtain different dc voltage. Driver Stage In vacuum tube circuits the output trapezoid of the horizontal deflection oscillator is a sufficient amplitude to drive the output stage directly. In transistor circuits a driver stage is provided between the oscillator and output stage. This stage is a buffer amplifier that isolates the oscillator from the output stage and has sufficient gain to meet the input power requirements of the horizontal output stage. The drive power is usually large. For example, during the interval when output transistor is on, the base current may exceed 400 mA. Note that because of low impedance values in transistor circuits, the drive voltage is usually an asymmetrical rectangular wave instead of a trapezoid.

20.11 TRANSISTOR LINE OUTPUT STAGE
Figure 20.14 is the circuit of a basic line (horizontal) output stage. The corresponding input and output signal waveforms are illustrated in Fig. 20.15. The damper diode D1 and flyback capacitor C1 are connected directly across the collector and emitter of the transistor. The value of the flyback capacitor C1 is so chosen that it forms a resonant circuit with the equivalent inductance L of the deflection circuit at a frequency whose period is twice the retrace interval. During the period marked T1 in Fig. 20.15(c) the input signal forward biases the output transistor into full conduction and the resulting current through the yoke coil deflects the beam to the right of the raster. The damper diode remains reverse biased by the VCC supply during this interval. At instant t1 the sharp negative pulse at the base of Q1 turns it off and the current sharply declines to initiate retrace. The beam returns to the left of the raster during the first half cycle of self-oscillations. The self-induced voltage across the coil keeps the damper diode back-biased during this (T2) interval. The polarity of the induced voltage reverses at instant t3 and forward biases D1 which then conducts to deflect the beam to the centre of the screen. The energy stored in the coil is

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thus expended and any subsequent oscillations are suppressed. The transistor Q1 continues to be off for most of the time interval T3, i.e., (t4 – t3). A little before the instant t4 (see Fig. 20.15(a) the sharp rise of base voltage forward biases Q1 and it conducts heavily to continue deflection of the beam to the right of the raster. The diode D1 immediately ceases to conduct because of the disappearance of self-induced voltage in the yoke. The VCC supply, however, keeps it off during this interval and thus the cycle repeats to scan the entire raster.
EHT Line output transformer

D2
C3

+ VCC D4 C4 + R1

400 V

D4 C5

+ R2

80 V

L1 T1 From driver R3 L2 Q1 D1 C1 C2

Deflection windings

Fig. 20.14. Transistor line output circuit.

The capacitor C2 is for ‘S’ correction. The correction occurs because of the voltage which develops across it when sawtooth current flows through the coils. This capacitor also blocks dc to the yoke coils which would otherwise cause heating of the coils due to dc resistance of the windings. The high voltage pulses are rectified to provide EHT supply. A stack of silicon diodes is usually used as a high voltage rectifier. As explained earlier, unlike tube circuits, the efficiency (damper) diode does not provide any boosted B++ supply. However, high voltage requirements for the picture tube anodes and other auxiliary circuits are obtained by transforming flyback pulses and rectifying them. As shown in Fig. 20.14 suitable ac pulse sources are rectified by diodes D3 and D4 to produce dc voltages of 400 V and 80 V respectively. The capacitors C4 and C5 are the filter capacitors to remove 15625 Hz ripple from the rectified outputs. Resistors R1 and R2 act as bleeder resistors to discharge the capacitors when the receiver is switched off.

20.12 HORIZONTAL COMBINATION IC CA 920
The ‘BEL’ CA 920 is a 16-pin dual-in-line plastic package, line (horizontal) oscillator combination IC which contains all the small signal stages for horizontal deflection in television receivers. The various functions performed by the device are (i) sync pulse separation with optional noise gating, (ii) current controlled line oscillator, (iii) phase comparison between sync pulses and oscillator output, (iv) high synchronizing noise immunity and (v) phase comparison between the oscillator waveform and middle of the line flyback pulse for automatic correction of switching delays in the horizontal driving and output stages.

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Phase comparison can take place either between the oscillator signal and sync pulse or between the sync pulse and line flyback signal. With phase comparison between sync pulse and line flyback signal, the various inherent phase shifts that are associated with the oscillartor, line driver and line output stages are automatically compensated for. The disadvantage, however, of this method is that the amplitude, shape and duration of the line flyback pulse are dependent on picture tube beam current and other dynamic variations. Hence design on the filtering circuits cannot be optimum both for good synchronizing and noise immunity. If, however, phase comparison between sync pulse and oscillator signal is used, the controlled phase position is independent of the amplitude, shape and duration of the line flyback pulse. The difficulty in this case is that the static and dynamic changes in the switching times of the output stage may now appear and suitable measures have to be taken for their correction. Simple discrete circuit for flywheel synchronization will not be able to provide a fully satisfactory solution because of the contradictory requirements involved. By incorporating two control loops, one comparing the sync pulse and the oscillator signal and the other the oscillator signal and the line flyback pulse, the problem is comprehensively solved in this IC (CA 920). Typical Application Circuit A typical drive circuit using CA 920 for horizontal deflection in TV receivers is shown in Fig. 20.16. Composite video signal having peck to peak amplitude of 1 to 7 volts with positive going sync pulses is applied at the input. The low-pass filter R1, C1 with cut-off frequency of about 500 KHz filters out noise present without introducing excessive delay to the signal. Before this video signal is applied to terminal (8) of the IC, proper clamping should be done so that only the sync tips drive the sync separator stage. This is achieved by the circuit consisting of R2, R3, C2 and C3 . R2, C2 along with base-emitter junction of the input transistor within the IC, clamps the video signal while R3, C3 combination ensures that only since tips drive current into terminal (8) irrespective of video signal amplitude. A good set of values for R2, R3, C2 and C3 are 1.5 M, 10 K, 100 KPR and 1 KPF respectively. Impulsive noise present in the composite video is separated from the signal by the circuit D1, R7, to R11, C6 and C7 and applied to terminal (9) of the IC. This voltage inhibits the sync separator stage under impulsive noise conditions thus making pin (9), the noise gating terminal of the IC. The composite sync at terminal (7) is processed into vertical sync by means of the integrator R4, C4. The time constant of this integrator is chosen to be around 100 µs in order to obtain as large an amplitude as possible for the vertical sync pulses and for maximum rejection of horizontal sync pulses. The horizontal sync pulses are separated by the differentiator R5, R6, C5 combination. This circuit provides voltage division, and has a time constant of approximately 8 µs. A value of 10 KPF for Cosc and 2.7 K for Rosc are chosen to be connected at terminals (14) and (15) respectively to determine free running frequency of the oscillator and to obtain optimum transconductance for the oscillator. The free running frequency can be adjusted by the resistor network R12, R13, R14 and RH. The capacitors C8 and C9 act as low-pass filter in this control loop. The control current I12 is converted into a dc voltage by this filter. This dc voltage drives a controlling current determined by R16 connected between terminals (12) and (15). A value of 33 K for R16 is optimum in order to achieve equal pull-in and hold ranges. The flyback pulse derived from the line output transformer is fed at terminal (5) of the IC through a suitable resistor to provide approximately 1 mA of current. The control current I4 at terminal (4) of phase comparator is converted into a dc voltage by R18, C11 combination and

The output, available at terminal (2) of the IC can drive both transistor and thyristor (SCR) output stages. The 12 volts power supply is fed at pin (1). This voltage is derived from a zener regulator. A 10 ohm resistance is connected in series with pin (1) to limit the supply

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current in case of increased supply voltage. The total supply current drawn by the entire circuit is about 25 mA at 12 volts. This circuit when used in conjunction with a typical line output stage using transistors BD 115, BU 205 and line output transformer AT 2048/12 provides equally large pull-in and hold ranges of about ± 4.5 percent. The output circuit shown in Fig. 20.16 in conjunction with this IC employs a BU 205 power transistor and the associated output transformer. This set-up is suitable for driving the yoke of a 51 cm picture tube. The stage operates from a 200 V dc source obtained from a regulated power supply. Besides providing beam deflection and other outputs, the circuit produces an auxiliary low voltage of 40 V required for audio, and RF-IF stages. The stage also provides heating power to the picture tube filament circuit.

20.13 HORIZONTAL DEFLECTION CIRCUITS IN COLOUR RECEIVERS
While the basic mode of operation is the same, horizontal deflection circuits in colour receivers are somewhat different than those provided in black and white receivers. The following are the main distinguishing features of colour receiver line circuits. (a) A colour receiver picture tube may require as high as 25-KV at its final anode but such a requirement seldom exceeds 18 KV in a monochrome receiver. In addition, many colour receiver picture tubes need another high dc source of about 6 KV for electrostatic focusing of the electron beam. Since these dc supplies are developed in the horizontal output circuit, colour receivers employ special high voltage rectifier circuits to generate them. High Voltage Tripler Circuit To generate a very high voltage dc source a voltage-tripler circuit is often used. This arrangement minimizes insulation requirements of high-voltage transformers. Figure 20.17 shows the schematic of such a tripler circuit. The entire circuit is potted and can only be replaced as a unit in the event of failure. Though there are five rectifiers in the module, two are used to couple the rectified dc from one rectifier to another. The other three rectifiers and associated capacitors make up the voltage tripler. Rectifiers instead of resistors are used to couple the dc voltages because there would be loss of both voltage and power if resistors were used.
HV tripler H.O.T.

25 kV 220 M 12 M R2 R5 + R1

10 M

R3 R4

6 kV To picture tube focus anode 200 pF 1.5 M

0.1 mF

27.5 M

Fig. 20.17. High voltage tripler circuit.

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R1 and R2 are special high voltage resistors. R1 (220 M) is a bleeder that discharges the high voltage picture tube capacitor after the receiver is switched off. The other high-voltage resistor R2 is connected to a tap on the tripler circuit. This together with potentiometer R3 and resistor R4 forms a voltage divider network to supply high voltage dc to the focus electrode of the picture tube. All the high-valued resistors can be adversely affected by handling. Leakage paths caused by fingerprints may result in a dramatic drop in resistance. Therefore, these parts should be handled as little as possible, and when necessary should be held by their leads. If dust or dirt accumulates on them, it should be removed by washing them with alcohol or any other ‘Freon’ type cleaning solvent. (b) The electron beam current of a black and white picture tube generally does not exceed 500 µA. The beam current of a colour picture tube may be two to three times this value. At maximum brightness a colour picture tube may dissipate as much as 25 W. The internal resistance of the high voltage power supply in the line output circuit is usually large and variation of brightness results in blooming and poor focus unless suitable regulator circuits are employed. Since the beam current of colour picture tubes is very large such effects are more pronounced in colour receivers. High Voltage Regulation In an attempt to keep the picture tube anode voltage almost constant under varying brightness conditions many monochrome and all colour receivers employ special high voltage regulator circuits. One such arrangement which is used in vacuum tube circuits was discussed in Section 20.9 while explaining an improved horizontal output circuit. Solid state colour receivers do not usually employ direct high voltage regulators, because, compared to vacuum tubes, transistor line output circuits have very low impedance levels. As such, changes in high voltage load current do not cause large changes in dc output voltages. Nevertheless, most transistor colour receivers do incorporate some form of high voltage regulation. It may be noted that it is not practicable to regulate the bias of an output transistor as is often done in tube circuits because the transistor acts more like a switch than an amplifier. Therefore, regulation of sweep and high voltage in transistor colour receivers is often done by regulating B+ supply voltage that feeds the horizontal output stage. Such regulator circuits are described in Chapter 24 which is devoted to receiver power supplies. (c) In order to reduce X-ray radiation from a television receiver under breakdown conditions, special hold-down circuits are used to automatically disable the rectifier when a fault occurs which would tend to increase excessively the generation of EHT supply. While most monochrome receivers do not include such circuits but a hold-down circuit is a must in colour receivers where the EHT voltage is very high. Hold-down Circuit An example of a high voltage hold-down circuit used in transistor colour receiver circuits is shown in Fig. 20.18. The circuit is so designed that an increase in high voltage above 30 KV causes an SCR to conduct and ground the horizontal oscillator supply voltage. This causes the horizontal oscillator to cease functioning resulting in a loss of both, the high voltage and the raster.

The gate of the SCR is fed from a voltage divider which is placed directly across the EHT supply. The zener diode (D1) placed in series with the gate circuit ensures that the SCR will turn on precisely at the desired voltage. As soon as the voltage exceeds 30 KV, zener breakdown occurs and a positive gate voltage turns on the SCR. The potentiometer R2 is set for optimum operation of the circuit. The SCR will remain on, until either the receiver is turned off or the cause of excessive high voltage is removed. (d) As shown in Fig. 20.19 colour receivers require dynamic convergence and antipincushion distortion circuits. These are in part activated by pulses obtained from the horizontal output amplifier. Such circuits are not required in monochrome receivers.
Phase Sync input Vertical oscillator and output circuit V Pincushion circuit Top and bottom pincushion control Height Hold Linearity Dynamic convergence circuit Deflection yoke 15625 Hz To HV rectifier (25 kV) To focus anode (6 kV) To HV regulator Damper Horz efficiency coil

50 Hz

To dynamic convergence assembly

Sync input

Horz AFC and oscillator H Hold

Horz output amplifier

To HV hold-down circuit

Fig. 20.19. Block diagram of the deflection system of a colour receiver.

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(e) Since the sawtooth deflection current requirements in colour receivers are much higher than monochrome receivers, horizontal output amplifiers of colour receivers need a drive voltage which is almost double than that needed to drive corresponding amplifiers in monochrome receivers. As a consequence line output amplifiers of colour receivers consume nearly double power than similar monochrome receiver circuits.

20.14 SCR HORIZONTAL OUTPUT CIRCUIT
In some colour receivers SCRs (silicon controlled rectifiers) are used to produce a sawtooth yoke current in horizontal output circuits. Such a circuit employs a pair of SCRs in conjunction with two special diodes and a number of resonant circuits to form a system of switches which generate the sawtooth deflection current. An SCR deflection circuit is operated directly from the rectified ac line voltage. It does not need a matching transformer and the purpose of fly-back transformer is to develop high voltages pulses for EHT and other circuits. It also provides, like other line circuits, pincushion correction, convergence, high voltage regulation and high voltage hold-down. However, SCR deflection circuits need exact component values and the tuning of various resonant circuits is quite critical.
CB B+ LG1 T1 LG2 LR SCR2 From horizontal osc Gate CR SCR1 G R1 Trace elements (a) Trace LR S2 CR S1 T2 LY CY T0 (b) (c) T1 T3 T4 D1 Flyback transformer CG LG3

The resonant circuits formed by various combinations of coils and capacitors in the circuit (Fig. 20.20(a)) are allowed to oscillate alternately and later turned off at appropriate moments by the switching action of silicon controlled rectifiers and associated diodes. Since a

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large number of resonant circuits, each tuned to a different frequency, are used to obtain different segments of the sawtooth current, the circuit operation is quite complicated. Therefore the explanation that follows has been greatly simplified for a better understanding of the basic principle of its operation. Circuit Components In Fig. 20.20(a) two SCR units are used in the horizontal output circuit. SCR1 conducts for trace periods while SCR2 turns on during retrace periods. Switching of SCR2 for retrace is called the commutating operation. Each SCR has a special fast recovery diode with inverse parallel connections. The purpose is to allow current in both directions from cathode to anode in either the SCR or its diode, depending on which anode is positive. When the diode conducts its low resistance drops the SCR anode voltage to the cur-off level. During trace time SCR 1 is on and SCR2 is off. In effect, SCR1 and D1 together comprise a controlled single pole-single throw switch (S1) while SCR2 and D2 form another similar switch (S2) as shown in Fig. 20.20(b). The components LR, CR, CH, and CY supply the necessary energy storage and timing functions. The inductance LG1 gives a charge path for CR and CH from B+, thereby providing a means to ‘recharge’ the system from the power supply. The secondary winding (LG2) on transformer T1 supplies necessary gating current to SCR1. The capacitor CH controls the retrace time because it is charged to B+ voltage through LR. Pulses from the horizontal oscillator at 15625 Hz turn on the retrace SCR at appropriate moments. The conduction in SCR2 cuts-off the retrace silicon controlled rectifier SCR1. Thus the trace and retrace periods are controlled and synchronized with the horizontal oscillator. Circuits Operation Figure 20.20(b) is the simplified equivalent circuit of the SCR deflection circuit. The waveforms shown in Fig. 20.20(c) illustrate approximate time intervals during which switches S1 and S2 (trace and retrace SCR-diode combinations) close or open to circulate sawtooth current in the deflection windings. If the circuit has been functioning for many cycles and retrace has just been completed, the yoke (LY) magnetic field is at its maximum value and the electron beam is on the left side of the raster. This corresponds to time T0 in Fig. 20.20(c) where the induced current through LY is at its peak negative value. Trace Time Referring to Fig. 20.20(b), during the first half of trace time (T0 to T1) switch S1 (SCR1 and D1 combination) is closed causing collapse of the magnetic field around the yoke inductance (LY). The resulting current deflects the beam to approximately middle of the screen and charges the capacitor CY. During the second half of the trace time interval (T1 to T2) the current in the yoke circuit reverses because the capacitor CY discharges back into the yoke inductance Ly. This current causes the picture tube beam to complete the trace. Note that SCR1 and D1 combination (S1) continues to provide current conduction path during the entire trace interval when current changes from negative to its positive maximum value. Retrace Initiation The forward scan stops at T2 because a pulse applied to SCR2 from the horizontal oscillator triggers it into conduction (S2 closes). Both S1 and S2 remain closed for a very short interval.

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The charge previously stored on CR is released into the commutating circuit comprising of LR, CR and CH. Thus the current through SCR1 rapidly falls to zero causing the turning off of switch S1 and the initiation of flyback. As stated earlier the frequency of various resonant circuits is so chosen that trace and retrace intervals are correctly obtained. Retrace With S1 open and S2 closed current flows through a series resonant circuit comprising of LR, CR, LY and CY. The natural resonant frequency of this circuit is much higher than that of the yoke circuit LY and CY because the value of capacitor CR is much smaller than that of CY. As a result a fast flyback becomes possible. The retrace current comes to zero (T2 to T3) at the midway point of the flyback interval, and just then SCR2 stops conduction. However, diode D2 now becomes forward biased due to reversal of polarity of the oscillatory voltage. Thus S2 continues to be closed and a retrace current flows in the opposite direction to complete the flyback at instant T4. This action (current flow) effectively transfer the energy stored in CR back to the inductance LY of the horizontal deflection coils. At time T4 the diode D1 becomes forward biased due to reversal of voltage polarity (S1 closes) and shortly afterwards S2 opens. The field about the yoke inductance (LY) starts to collapse and the resulting current again commences the next trace period. The cycle repeats to provide horizontal scanning at the rate of deflection frequency. Note that during flyback the primary of T1 is connected between B+ source and ground via SCR2 and D2. When D2 stops conduction, LGI is disconnected from ground and CR charges through this winding from the B+ supply to replenish energy in the deflection coil circuit. It may also be observed (see Fig. 20.20(a)) that the voltage developed across LGI (primary of T1) during the charging of CR is coupled to the gate of SCR1 through the secondary of T1 and waveshaping network comprising of CG, LG3 and R1. In turn, the gate waveform that is generated possesses adequate amplitude to enable SCR2 to conduct while its anode is sufficiently positive.

Review Questions
1. Why is the design of the horizontal output stage very much different than that of the vertical stage, while their functions are very similar to each other ? What are the special features which make the operation of the line output stage very efficient ? Explain fully with a suitable functional diagram and waveshapes, how the first half-cycle of selfoscillations in the output circuit is used to obtain a fast retrace of the beam. How is the continuation of ringing suppressed ? Show how the energy stored into the deflection windings is recycled to generate boosted B++ supply. Explain with a circuit diagram how the high voltage pulses, induced in the output transformer windings, are used to generate EHT supply. What do you understand by reaction scanning ? Explain step by step the sequence of operations in the line output stage for reaction scanning. Draw suitable wave-shapes and diagrams to illustrate your answer. Explain the functions of linearity and width controls normally provided in the horizontal output stage. Describe their operation by drawing relevant portions of the output circuit.

2.

3. 4. 5.

6.

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7.

With reference to Fig. 20.11 explain how the feedback circuit maintains a constant EHT supply and width of the picture through the voltage drop across the VDR. What is the function of potentiometer R2 ? What do you understand by third harmonic tuning in the line output transformer ? Justify that is results in lesser peak power from the output tube and more ac voltage from the EHT rectifier. In what respects a transistor line output stage is different from a corresponding vacuum tube circuit ? Why is a driver stage necessary in transistor output circuits ? Why is a separate flyback capacitor provided in such circuits ?

8. 9.

10. Draw the circuit diagram of a typical horizontal output stage which employs a transistor and explain how it functions to provide linear horizontal scanning. What is ‘S’ correction and how is this applied ? 11. Discuss briefly the main features of the horizontal combination IC, CA 920. With reference to Fig. 20.16 explain the circuit operation of this IC. 12. Describe fully the distinguishing features of horizontal deflection circuits of colour receivers. Why is the deflection current very high in colour receivers ? 13. Draw the circuit of a high voltage tripler and explain how EHT voltage of the order of 25 KV is developed. Why is it necessary to handle high valued resistors very carefully ? 14. Why is a hold-down circuit necessary in colour TV receivers ? Draw the circuit diagram of such a circuit and explain how it operates to disable the horizontal oscillator when EHT exceeds the prescribed upper limit. 15. Draw basic circuit of a horizontal output stage employing SCRs. Explain how the circuit functions to provide linear trace and retrace periods. What are the merits and demerits of such deflections circuits ?

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21
Sound System

21
Sound System
A television signal transmitted from a broadcast station consists of an amplitude modulated RF picture carrier and a separate frequency modulated RF sound carrier both within the 7 MHz channel bandwidth alloted to the station. This signal is intercepted by the receiver antenna and coupled through a transmission line to the RF section in the tuner. The block diagram of Fig. 21.1 shows various stages through which the sound signal passes in a monochrome and a colour receiver, before it is finally delivered to the loudspeaker. In the tuner FM sound signal is amplified along with the picture signal and then transferred to the sound IF frequency from the channel carrier frequency. The sound IF signal together with its sidebands is passed through the IF amplifiers without much amplification. At the video detector it is made to beat with the strong picture IF carrier frequency to transfer the modulating audio signal to yet another sound IF frequency which is equal to the difference of the two intermediate frequencies (38.9 HHz—33.4 MHz = 5.5 MHz). This amounts to heterodyning the sound signal twice. Since the 5.5 MHz sound carrier is obtained by mixing the two IF carrier frequencies, this method of processing the sound signal is known as ‘InterCarrier Sound System’. The 5.5 MHz FM sound signal thus obtained is coupled from either video detector or video amplifier to the sound IF amplifier. The sound IF signal, together with its sidebands is amplified to a level of several volts for the FM detector. This stage converts FM modulation to equivalent amplitude modulation and then detects it to recover the audio output. This is amplified in the audio section to have enough output to drive the loudspeaker.

21.1 SOUND SIGNAL SEPARATION
The output from the last IF stage consists of picture IF (38.9 MHz) amplitude modulated with the composite video signal and sound IF (33.4 MHz) frequency modulated with the audio signal. The two intermediate frequencies with their sidebands occupy a frequency spectrum of about 6.75 MHz. This signal is coupled to the video detector (see Fig. 11.6) which employs a diode and rectifies the centre zero modulated signal to produce a dc component, sum and difference frequencies of the various constituents of the input signal and their harmonics. The resultant band of frequencies is passed through a low-pass filter that forms part of the video detector. The cut-off frequency of the filter configuration is set at 5.75 MHz. Thus all the sum components and higher order harmonics which lie above the cut-off frequency (5.75 MHz) are filtered out. The output then consists of two distinct components. The band the lies between 0 and 5 MHz is the demodulated video signal and is fed to the video amplifier. The second component that 400

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lies around 5.5 MHz is formed as the difference of the picture IF and sound IF with its sidebands i.e., 38.9 — (33.4 ± sidebands) = 5.5 MHz ± sidebands.
(Colour receiver) Antenna Sound IF detector 5.5 MHz Tuner and IF section Video detector 5.5 MHz

5.5 MHz

Audio signal Audio amplifier Speaker

Sound IF amplifier

FM sound detector

These stages are identical for both colour and monochrome receivers Video amplifier Alternative path

(Monochrome receiver)

Fig. 21.1. Sound signal path in monochrome and colour receivers.

In fact the strong picture signal intermediate frequency acts as the carrier frequency to a relatively low amplitude sound signal intermediate frequency and its sidebands to provide another frequency translation to the sound signal. This shifts the sound mdulation around a much lower carrier frequency of 5.5 MHz which is the difference of the two intermediate frequencies. This technique of separating the sound signal is known as the intercarrier sound system. The output has some amplitude variations. If these variations are large, it becomes difficult to remove them at the limiter stage, with the result that the audio output is severely distorted. This can be minimized by keeping the sound carrier signal amplitude very low. This explains why the sound IF which lies on the left skirt of the overall IF response is attenuated by about 20 db in comparison with the picture IF frequency. In fact this is the pivotal point for making the intercarrier sound system a success. Accordingly, the sound carrier together with its sidebands is given only 5 percent gain, in comparison with the video signal, to obtain a beat note at the detector, which contains only the frequency modulation of the original sound IF carrier and practically nothing of the video modulation. Another factor that needs attention is the buzzing sound that results from the 50 Hz vertical sync signal interference with the 5.5 MHz sound signal. This is known as intercarrier buzz and is the result of cross-modulation between the two signals. Similarly, the 15625 Hz horizontal sync can result in a hissing sound in the loudspeaker. Here again the remedy lies in keeping the magnitude of the sound signal much lower than the picture signal to keep the beat note amplitude low. This, together with the limiting that is provided after the sound IF stage, eliminates practically all amplitude variations in the FM sound signal and normally no intercarrier buzz or hiss is audible in the loudspeaker.

21.2 SOUND TAKE-OFF CIRCUITS
The usual sound take-off point in a monochrome solid state receiver is at the output of video driver stage. In Fig. 21.2 the video driver is connected as a phase-splitter where the emitter

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circuit feeds the video output amplifier and the tuned transformer T1 in the collector circuit separates sound IF from the composite video signal. Since the emitter circuit contains full composite signal, including the 5.5 MHz sound IF and its sidebands, a shunt trap (L1, C1) tuned to 5.5 MHz is used to eliminate this source of intereference. The transformer T1 is tuned to 5.5 MHz and delivers sound signal to the base of Q2, the sound IF amplifier. This transformer is tapped to provide proper impedance match and to maintain a hight Q of the tuned circuit.
5.5 MHz sound take-off tansformer (T1) 1000 pF 680 Q2

Sound IF amplifier

From video detector Q1

5.6 K Bias 100 K 330 L1 – VCC C1

0.01 µF 30 µF 100 pF To video output amplifier

5.5 MHz

Fig. 21.2. Sound take-off circuit in a monochrome receiver.

A typical sound take-off and detector circuit used in colour receivers is shown in Fig. 21.3. It can be seen that the sound take-off point is located before the video detector to minimize the 1.07 MHz interference that would be produced if the sound IF (5.5 MHz) is permitted to beat
Sound take-off point Sound detector D1 C1 R3 L7 R2 C5 L4 + VCC 33.4 MHz trap L1 R1 C2 L2 C3 C4 To sound IF amplifier (5.5 MHz and its side-bands)

3rd video IF amplifier 10 pF From 2nd video IF

L3

5.5 MHz

C6

D2 Video L detector 5

C7 L6

C8

To video amp (0–5 MHz)

5.5 MHz trap

Fig. 21.3. Sound take-off and detector circuit in a colour receiver.

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with the 4.43 MHz colour subcarrier. Conversion of sound IF to intercarrier sound (5.5 MHz) is accomplished by the diode D1 which acts like a series detector. The input to the detector is via a series resonant circuit C1, L1. This circuit is tuned to about 36 MHz and is broad enough to include both picture and sound IFs. R1 and C2 are the load resistor and IF filter. C3 and L3 constitute a filter that separates the sound IF from the composite video that also appears at the output of the sound IF detector. The inductor L2 blocks the video IF and its harmonics. Since the sound take-off is before the video detector, greater attenuation of the 5.5 MHz sound IF is possible in the video detector circuit. This is done (see Fig. 21.3) by bridge type trap circuits located before and after the detector. The trap circuit located before the video detector (D2) is tuned to 33.4 MHz and the one after the detector is tuned to 5.5 MHz. Thus signals at these frequencies are highly attenuated and do not reach the cathode of the picture tube.

21.3 INTER-CARRIER SOUND IF AMPLIFIER
The amplitude of the inter-carrier sound signal at the output of the video detector is very low and so at least two stages of sound IF amplification are provided before feeding it to the FM detector. This second IF stage is also used as a limiter, if necessary. Each IF stage is a tuned amplifier with a centre frequency of 5.5 MHz and a bandwidth of over 150 KHz to provide full gain to the FM sidebands. The desired bandwidth, though large, is easily attained because it is a relatively small percentage (≈ 2%) of the intermediate frequency. This is similar to a radio receiver IF amplifier where the necessary bandwidth of 10 KMz is also nearly 2% of the 455 KHz IF frequency. However, the design and tuning of a TV sound IF stage is critical since in FM, narrow IF bandwidth causes amplitude distortion on loud signals because of insufficient response for maximum frequency deviation. This effect corresponds to clipping and limiting in an audio amplifier which can make the sound unintelligible. In AM, however, the loss of high frequencies that results on account of insufficient IF bandwidth is not too obvious in the reproduced sound.
R1, R2-Biasing network C2, R4-Decoupling circuit CN-Neutralizing capacitor CN 3 pF 3.7 K C1 Q1 IF input (5.5 MHz ± 75 KHz) .01 mF R1 R2 R4 C2 R 3 C3 50 pF

Sound IF output

+ VCC

+ VCC

Fig. 21.4. Circuit diagram of a typical sound IF amplifier.

Figure 21.4 shows the circuit diagram of a typical transistor IF amplifier. The signal from the video detector is coupled through an impedance matching transformer, the secondary of which is broadly tuned to the intercarrier sound IF frequency. The output transformer has

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several extra turns on the primary winding to provide neutralization through a very small capcitor. Without negative feedback the stage will have a tendency to oscillate which can cause undesired high pitched sound from the speaker.

21.4 AM LIMITING
The frequency modulated sound signal has some amplitude variations, both because of heat formation at the video detector and due to somewhat unequal amplification at the RF and IF stages of the receiver. AM interference can be eliminated by limiting amplitude variations. Amplitude modulation rejection can be accomplished either by using the last IF stage as an FM limiter or by using an FM detector that does not respond to amplitude variations. FM discriminator circuits require a limiter for removing AM if present from a FM signal before detection. However, the ratio-detector and the quadrature-grid detector donot need a limiting stage because they are insensitive to amplitude variations in the FM signal. The Limiter The limiter is similar to the preceding IF stage but its dynamic range is kept narrow to achieve the desired clipping action. The dynamic range is reduced by using self-bias at the input and causing early saturation by reducing dc supply to the output circuit. Figure 21.5 (a) shows a typical FET amplitude limiter. Note the use of Rg,Cg for self-bias and R1 for reducing the drain voltage. The stage operates as a class C amplifier. With varying signal amplitudes, the bias automatically adjusts itself to a value that allows just the positive tips of the signal to drive the gate positive and cause gate current flow. This maintains necessary self-bias to cause limiting.

CN C2 R1 v0 (limited)

FET Sound IF input (vin) Rs CS R2 C1

B+

Rg

Cg

Fig. 21.5 (a). Amplitude limiter circuti employing an FET.

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v0 5V Limiting threshold 2 3

1
Limiting range

4 vin

0

.5 V

4.5 V

Fig. 21.5 (b). Typical limiter response characteristics.

Figure 21.5 (b) shows typical response characteristics of the amplitude limiter. It indicates clearly that limiting takes place only for a certain range of input voltages, outside which output varies with input. When the input voltage is too low (range 1 to 2) the stage behaves like a class A amplifier and no limiting takes place. Once the signal amplitude (peak-to-peak) exceeds the dynamic range, limiting action commences to deliver a constant output voltage. In the limiting region (range 2 to 3), as the input voltage increases the output current flows for a somewhat shorter portion of the input cycle to maintain a constant output voltage. Note that, though the current flows in short pulses (class C operation), the output voltage is sinusoidal. This is due to the flywheel action of the output tuned (tank) circuit. However, when the input voltage increases sufficiently (range 3 to 4), the angle of output current flow is reduced so much that less power is fed to the output circuit and the output voltage is reduced. Thus to ensure proper limiting action the input signal amplitude should remain within the limiting region. In TV receivers this is ensured by AGC action. Most present day receivers employ transistors in the sound section and use a ratiodetector which does not need a limiter. However, the last IF stage is designed to limit large amplitude variations, if any, and also to provide reasonable gain to the FM sound signal.

21.5 FM DETECTION
FM detection is carried out in two steps. The frequency-modulated IF signal of constant amplitude is first converted into a proportionate voltage that is both frequency and amplitude modulated. This latter voltage is then applied to a detector arrangement which detects amplitude changes but ignores frequency variations. Slope Detection Amplitude variations in an FM signal can be provided by using one of the sloping sides of a tuned circuit. Consider a frequency-mudulated signal fed to a tuned amplifier (Fig. 21.6 (a)) whose resonant frequency is on one side of the centre frequency of the FM signal. As shown in Fig. 21.6 (b), the circuit is detuned to bring the centre frequency (5.5 MHz) to point A on the selectivity curve. The input to the amplifier is shown to have a frequency variation for a given audio modulation. As the input signal frequency shifts, the frequency deviations of the carrier signal are converted into proportionate amplitude variations (see Fig. 21.6 (b)). These AM variations result from the unequal IF gain above and below the carrier frequency. Thus the IF output varies in amplitude at the audio rate, in addition to its cotinuously changing frequency.

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The resulting AM signal can be coupled to a diode detector to recover the audio voltage.
Audio signal vin v0 Tuned circuit

Transformer Action While the slope detector described above illustrates the principle of FM detection, it is both inefficient and linear over a very limited frequency range. In practical FM detectors, tuned circuits and other coupling techniques are used to transform frequency deviations of the incoming carrier into corresponding amplitude variations. In order to fully understand this transformation, it would be necessary to examine the phase relationships in a tuned transformer as the frequency of the signal applied to it changes. As shown in Fig. 21.7 (a), the primary is a parallel reasonant circuit but the secondary behaves like a series resonant circuit because the voltage is induced in series with the secondary as a result of the primary current. The magnitude of the induced voltage (e induced) is equal to ± jωMIp where M is the mutual inductance between the two windings. The sign of the induced voltage depends on the direction of winding. It is simpler to assume connections giving negative

Three cases will be examine: (i) at resonance, (ii) above resonance and (iii) below resonance. (i) At resonance. The induced voltage (e induced) ‘sees’ a resistive circuit at resonance. Therefore, the secondary current (ic) is in phase with the voltage. This current while flowing into the secondary tuned circuit produces a voltage drop (ec) across C2 which lags the current by 90°. The voltage across the capacitor is also the voltage across the secondary winding and is not the same as the induced voltage. The relations between the primary and secondary voltages are illustrated in Fig. 21.7 (b). Notice that the output secondary voltage (ec) for case (i) is 90° out of phase with the voltage across the primary. (ii) Above resonance. If the frequency of operation is increased above resonance (above 5.5 MHz in FM sound signal) the secondary series circuit impedance will appear to be inductive. As shown in the corresponding phasor diagram the secondary current (ic) now lags the induced voltage by some angle that depends on Q of the circuit and the amount of input signal frequency deviation. Since the voltage across the capacitor C2 lags the current by 90° the phase angle θ1 between the primary voltage and the secondary terminal voltage is now less than 90°. (iii) Below resonance. When the input signal deviates below resonance, the secondary circuit becomes capacitive. The secondary current now leads the induced voltage (see phasor diagram for input frequency below resonace) by an angle which depends on the extent of

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frequency deviation and Q of the circuit. Therefore, the secondary terminal voltage (e2) and the primary voltage (e1) are more than 90° apart. Thus it is evident from the phasor diagrams that a change in frequency will cause a change in phase between the primary and secondary voltages. If these voltages are suitably added together vectorially, as is done in practical discriminators, their vector sum will be seen to change in amplitude. This explains the technique of generating amplitude variations in the FM signal before feeding it to the diode detectors for demodulation.

21.6 FM SOUND DETECTORS
Earlier television receivers used specially designed multifunction vacuum tubes in FM sound detectors. However, tubes were soon replaced by solid state circuitry. In fact FM detector was one of the first receiver circuit block to be made into an integrated circuit. Since that time a number of different sound ICs have been developed. One of the aims of this development has been to cut down or eliminate the need for tuned circuit alignment. The other factors which brought about quick IC development have been system performance, cost and reduced circuit complexity. In general FM sound detectors may be classified as under: 1. Discriminator 2. Ratio detector 3. Quadrature detector 4. Differential peak detector 5. Phase-locked loop detector One of the earlier sucessful detectors was the Foster-Seely discriminator. Since a discriminator must be preceded by a good limiter, it was soon replaced by a ratio detector which has an inbuilt limiter. The ratio detector was commonly used in hybrid television receivers till the advent of ICs for the sound section. Earlier ICs employed ratio or quadrature detectors for demodulating the FM sound signal. However, recent sound ICs employ either a differential peak or a phase-locked loop detector because they need least tuned circuit alignment. When solid state devices were not available specially designed tubes (gated-beam) were developed which perform as limiter, detector and first audio amplifier, all in one glass envelope. However, their use is now limited to the old ‘all tube’ receivers which are still in operation. 1. The Foster Seeley Discriminator As already explained detection of the frequency modulated signal is carried out by first modifying the frequency spectrum of the wave in such a manner that its envelope fluctuates in accordance with the frequency deviations. In the Foster-Seeley discriminator (see Fig. 21.8 (a)) this is carried out by the vector addition of the instantaneous voltages that develop across the input transformer. Since the magnitude of the resultant voltage depends on the instantaneous phase angle between the primary and secondary voltages this detector is also known as ‘phaseshift’ discriminator. The resulting amplitude-modulated wave is then applied to the diodes D1 and D2, that form part of the balanced AM detector, to recover the audio modulating voltage.

The action of the discriminator may be explained as follows: The input transformer is double-tuned with both primary and secondary resonant at the centre frequency of 5.5 MHz. In addition to the inductive coupling between the two tuned circuits, the primary voltage ep, is also coupled through capacitor C3 (250 pF) at the centre of the secondary winding. The centre tap of the secondary winding is also connected to the common points of R1, R2 and C1, C2 through L1, a radio frequency choke. Thus the circuit composed of C3 and L1, with the other end of L1 grounded, comes effectively across the primary winding. At and around 5.5 MHz the reactance of L1 greatly exceeds that of C3. Therefore, voltage across L1 is almost equal to ep, the applied primary voltage. Accordingly the voltage fed to each diode is the vector sum of the primary voltage and the corresponding half-secondary voltage. As is obvious from the circuit diagram (Fig. 21.8 (a)), the induced voltage across the secondary divides into two parts at the centre-tap and is thus applied in push-pull to the diode plates. However, the primary voltage which effectively appears between the centre-tap and ground is applied in parallel to the two diodes. Therefore, the IF signal voltage for the two diodes is the resultant (vector sum) of the induced secondary voltage applied in push-pull and the primary voltage applied in parallel to the two diodes. Diode Voltage Phase-Relations. As explained earlier, in a tuned circuit the secondary voltage is 90° out of phase with the primary voltage at resonance. As the applied IF frequency swings above and below the resonant frequency, the phase angle varies around 90°. In the circuit of Fig. 21.8 (a) the secondary voltage is devided into two equal halves es1 and es2 by the

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centre tap connection. The voltages are of opposite polarity with respect to each other. Thus at the resonant frequency of the tuned-secondary circuit, the secondary voltages es1 and es2 are in quadrature with voltage ep existing across the primary inductance. When the applied frequency is either higher or lower than the resonant frequency of the secondary, the phase angles of es1 and es2 relative to ep will differ from 90°. The vector relations and magnitudes of the resultant phasors e1 and e2 are illustrated in Fig. 21.8 (b). At the centre (IF) frequency both es1 and es2 are 90° out of phase with ep with the result that e1 is equal to e2. At some IF signal frequency swing above the centre frequency, ep is shown 75° out of phase with es1 instead of 90°. This makes ep closer to es1 and as a reuslt e1 increases while e2 decreases. Similarly for some carrier frequency swing below the centre frequency, angle of es with ep increases to 105°, thus making e2 greater then e1. The voltages e1 and e2 are applied across the diodes D1 and D2 respectively. With no modulation, i.e., when IF frequency is equal to the centre frequency, e1 is equal to e2 and both the diodes conduct to pass equal currents. Since R1 is equal to R2, the net output voltage is zero. However, when the centre frequency swings above resonance to make e1 greater than e2, id1 exceeds id2 and the net output voltage is positive. Similarly when the carrier frequency swings below the centre frequency, e2 exceeds e1, and id2 becomes greater than id1 to develop a net negative output voltage. Note that the RF choke does not permit any current flow at the IF frequency and all high frequency components of the rectified current complete their path through ground. The rectified dc component, however, complete its path through the RF choke. The time-constant R1C1 = R2C2 is kept much larger than the period of IF frequency so that voltages across C1 and C2 cannot follow high frequency variations. However, it is kept much smaller than the highest audio signal period so that the voltage variations across the two capacitors can follow the audio variations. Note that the voltages across C1 and C2 subtract from each other to develop audio voltage across the output terminals A and B. Discriminator Response. The output voltage is the arithmetic difference of e1 and e2 which vary with the instantaneous frequency as shown in Fig. 21.9 (a). Therefore, deviations in the instantaneous frequency away from the carrier frequency cuase the rectified output voltage VAB to vary in accordance with the curve of Fig. 21.9 (b). The slope of this curve shows
e1, e2 e2 Peak separation e1

Relative response

100 KHz 0

100 KHz f(MHz)

fC (5.5 MHz)

Fig. 21.9 (a). Variation of e1 and e2 with frequency.

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+ VAB

411
Peak separation = 200 KHz

0

fC

f(MHz)

5.5 MHz – Below fC Above fC

Fig. 21.9 (b). Discriminator (‘S’ shaped) response curve.

that the rectified output VAB will reproduce with reasonable accuracy, the variations of the instantaneous frequency as long as the operation is confined to the region between the peaks of e1 and e2. In the illustration this is shown to be ± 100 KHz with a centre frequency of 5.5 MHz. 2. Ratio Detector In the Foster-Seeley discriminator any amplitude variations of the input signal give rise to unwanted changes in the resulting output voltage. This makes limiting necessary. It is possible to modify the discriminator circuit to provide limiting, so that the amplitude limiter can be dispensed with. A circuit so modified is called a ratio detector.
Above resonance

Balanced Ratio-Detector. The circuit of a balanced ratio detector is shown in Fig. 21.10. Neglecting capacitor C3 for the moment, this arrangement is seen to differ from the FosterSeeley discriminator in (i) diode D1 has been reversed and (ii) the output voltage is obtained between the grounded junction of R1 and R2 and A, the common point of capacitors C1 and C2. The input coupling transformer T1 has the same function as in a phase-shift discriminator. Both the primary and secondary tuned circuits are resonant at the IF (centre) frequency. The secondary is centre-tapped to provide equal voltages of opposite polarity for the diode rectifiers. The primary voltage is applied in parallel to both the diodes by a tertiary winding Lt which is wound on the same core and is connected at the centre-tap of the secondary winding. Lt is wound directly over the primary winding for very close coupling so that the phase of the primary voltage across Lp and Lt is practically the same. This arrangement (with the tertiary winding) of providing primary voltage to the secondary circuit is preferred in order to match the high impedance primary to the relatively low impedance secondary circuit. The resistor R3 in series with Lt, limits the peak diode current for improved operation.
eS1 ep eS2

D1 iD1

e1 R3 + – v1 – v3 vR1 R1 R2 C3 + v2

.

e2 iD2

.

D2 A R4 CC To audio amplifier G vR2

C1

– + C2 C4 C5

G

Fig. 21.11. Equivalent circuit of the balanced ratio detector.

Circuit Operation. Consider the equivalent circuit of the ratio detector shown in Fig. 21.11. The primary voltage ep combines with the secondary voltages es1 and es2 to produce resultant voltages e1 and e2 as illustrated by the phasor diagrams of Fig. 21.8 (b). The two series circuits formed by equal resistors R1, R2 and equal capacitors C1, C2 are connected in parallel across the plate of D1 and cathode of D2. The voltages e1 and e2 are applied across the diodes D1 and D2 respectively. The rectified currents marked iD1 and iD2 flow through R3 to charge capacitors C1 and C2 with the polarity marked across each capacitor. The voltages v1 and v2 which develop across C1 and C2 are approximately equal to the peak value of the IF signal applied to each rectifier. The net voltage (| v1 | + | v2 |) which develops across the two capacitors also appears across the series combination of R1 and R2. Since R1 = R2 and their common point is grounded, the total voltage gets equally divided across the two resistors with the result that vR1 = vR2

|v1 |+|v2 | |v1 |−|v2 | = . 2 2 At the centre frequency where e1 = e2, equal voltages are developed across the capacitors making v1 = v2. This results in zero audio output voltage. When the FM input signal is above centre frequency, e1 is greater than e2, with the result that v1 exceeds v2. This makes point A more positive producing audio output voltage of positive polarity. Below the centre frequency e2 is greater than e1, and thus v2 exceeds v1. The result is audio output voltage of negative polarity at point A. The response curve of the ratio detector is also S-shaped and is essentially the same as shown in Fig. 21.9 (b) for the discriminator circuit.
| v1 | – An important characteristic of the ratio detector circuit can be illustrated by considering numerical values of v1 and v2. Assume that at the centre frequency, v1 = v2 = 4 V. As the frequency deviates above its central value, v1 increases in magnitude and v2 decreases by the same value. If v1 becomes 6 V, v2 reduces to 2 V. This makes point A more poisitive by 2 volts with respect to ground

frequency by the same amount, v1 reduces to 2 V and v2 rises to 6 V. This makes point A, 2 volts negative with respect to ground

value of 2 volts is generated. This is exactly half the magnitude to the output voltage obtained from a discriminator (Fig. 21.8 (a)) under similar input voltage conditions. This is so, because, in a discriminator the voltages across the capacitors combine to produce the audio output voltage whereas in the detector under discussion, the output arises as a result of variations in the ratio | v1 | / | v2 |, while the sum | v1 | + | v2 | remains substantially constant. In fact it is this behaviour that gives it the name—ratio detector. Stabilizing Voltage. In order to make the ratio detector insensitive to AM interference, the total voltage v3 = v1 + v2, must be stabilized so that it cannot vary at the audio frequency rate. This is achieved by connecting a large capacitor C3 across the series combination of C1 and C2. Since C3 is large it acts as a low impedance load to any change in amplitude that might otherwise occur. For example a momentary increase in the amplitude of v3 causes a large charging current to flow through the diodes into C3. This represents power absorbed from the primary and secondary resonance circuits, and so reduces the voltage applied to the diodes. Conversely, if the amplitude of the incoming signal tends to drop below the average amplitude, then C3 prevents the voltage v3 from dropping by discharging into R1 and R2. It is thus seen that the presence of C3 prevents amplitude variations which would otherwise occur in the voltage v3 and likewise in voltages v1 and v2. It may be noted that when a channel is changed and the magnitude of sound IF signal shifts to a new value, the voltage across C3 changes to another value and maintains itself. A 5 µF capacitor (C3) is considered adequate for this purpose. The capacitor C4 is connected across the audio output terminals to bypass higher frequency components. However, the audio

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voltage (VAG) effectively develops across C4 and varies in accordance with frequency modulation of the sound carrier. The audio output is followed by a de-emphasis circuit (R4, C5) before feeding into the audio section. Single Ended Ratio Detector. A simple and economical ratio detector is shown in Fig. 21.12. This is a single ended or an unbalanced version because the two diodes are not equally balanced with respect to ground. A component of the diode currents id1 and id2 flows through the series circuit formed by the two diodes, R1 and C1 in parallel and the secondary winding circuit. However, some current flows via C2 and ground to complete the circuit. Note that a part of the D1 current (id1) flows through C2 towards the ground while the corresponding part of D2 current (id2) flows through C2 in the opposite direction. Thus the two diode currents subtract across C2 to develop a voltage which varies at the audio rate. This function in the other ratio detectors is carried out by the capacitors connected across the series circuit formed by the diodes. The resistors R2 and R3 (1 K) in series with D1 and D2 respectively limit the peak diode currents to improve dynamic balance of the diodes at higher signal conditions. One of the series resistances (R2) is variable and may be adjusted for better balance. The time-constant of R1 in parallel with C1 is so large (200 ms) that the circuit neither responds to fast noise amplitude changes nor to the relatively slow changes in amplitude due to spurious amplitude modulation. This as explained ealier provides the desired limiting action. R4,C3 is the deemphasis circuit which also filters out unwanted IF components from the diode currents. Since the output voltage has only one ground terminal no dc level shift is necessary.
Sound IF amp M 0A 79 1K id1 R2 22 K eS2 L1 ep + VCC 27 K + VCC .002 mF C2 C3 0.01 mF .002 mF R5 AF out 100 K D2 R4 0A 79 id2 R3 C4 1K R1 C1

D1 Sound IF vin signal 5.5 MHz ± 75 KHz Q1 eS1 ep

10 mF

Fig. 21.12. Single-ended ratio detector.

3. Quadrature Detector Quadrature detection is another approach to FM demodulation. This method combines the functions of a limiter, a discriminator and an audio voltage amplifier. A specially designed tube known as gated-beam tube performs all these functions. The transistor version of this method employs several transistors to provide the same functions. Normally this forms a section of an IC specially developed for sound section of the receiver.

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Gated-Beam Tube. A dual-control quadrature detector or gated-beam tube is a specially constructed sharp cut-off pentode. The important characteristic of this tube is that the plate current can be sharply cut-off either by control grid or suppressor grid. When the control grid voltage changes from negative to positive the plate current rises sharply from zero to attain its maximum value. The cut-off voltage is usually close to – 2 V. Thus the control (limiter) grid provides limiting action to any amplitude variations which may be present in the incoming FM sound signal. The electrons passing through the limiter grid are accelerated by the positive potential on the screen grid. The quadrature grid (suppressor grid) also exercises similar control over the plate current to produce sharp cut-off and saturation characteristics. If the quadrature grid is made strongly negative the plate current of the tube is cut-off, on matter how positive the limiter grid is. Thus the two grids act as current control gates and unless both permit conduction simultaneously, no plate current can flow. It is this action which gives the tube its name of gated beam tube. Later circuits used a pentode having similar cut-off characteristics. Such circuits came to be known as quadrature grid detectors. Circuit Operation. A typical quadrature grid detector employing a pentode is shown in Fig. 21.13. The plate and screen grid are operated at positive potentials in a manner similar to corresponding circuits of conventional tube amplifiers. The FM signal vL is coupled to the limiter grid (G1) through a circuit broadly tuned to the intercarrier sound IF frequency. For an input signal over one volt peak-to-peak, the tube will be driven from cut-off to saturation or vice-versa to conform with the positive and negative excursions of the input signal. This results in excellent limiting and almost a square wave beam current is produced in the region beyond the input grid.
+ 680 K IP 300 V RL P 90 V G3 G1 From sound IF amplifier 10 to 500 mV K G2 2.5 V C3 –4V 120 V 100 K Vq C4 10 pF L1 Quadrature coil tuned to 5.5 MHz R1 C1 Iq 1.5 M CC

To audio power amplifier Volume control

vL

RK Tuned to 5.5 MHz ± 75 KHz 300 W CK 0.001 mF B+ 470 K R2 C2

.047 mF Bias network

Fig. 21.13. Quadrature grid FM detector circuit.

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The quadrature circuit consists of a slug tuned coil shunted by a fixed capacitor forming a parallel resonant circuit. The circuit is tuned to the sound IF centre frequency. It is connected to the quadrature grid G3, which is biased to about – 4 volts. Since grid 3 is at – 4 V and the screen grid at 120 V, the electrons in the space between the two grids are slowed down to form an outer space-charge. This is in addition to normal space charge near the cathode. The outer space-charge serves as a source of electrons for the suppressor or quadrature grid G3 and plate. As the input voltage swings from positive to negative, the amount of space charge at the virtual cathode near grid 3 also varies. This varying space charge causes current flow by electrostatic induction, in the circuit connected to the quadrature grid. The induced current lags the input voltage by 90°. The voltage across the tuned circuit also lags by 90° when the input signal frequency is 5.5 MHz to which the quadrature circuit is tuned. This is why G3 is called the quadrature grid. When the input signal frequency deviates below or above the centre frequency, the phase of quadrature voltage vq also varies below and above 90°. Figure 21.14 shows effect of limiter and quadrature grids on the plate current. When the signal swings above centre frequency, vq lage vL by more than 90°, because the tuned circuit behaves like a capacitive reactance at a frequency which is higher than its resonant frequency. The additional lag keeps either of the two grids in cut-off most of the time during the intervals of positive swing in the signal. This results in narrow pulses of plate current. Despite the fact that maximum value of plate current is the same, narrower pulses mean a smaller value of average plate current.
Space charge density f < 5.5 MHz q1 ip iq vq vL Space charge density f = 5.5 MHz vL iq ip Input signal Effect of limiter grid (G1) Effect of quadrature grid (G3) q2 vq Space charge density vq f > 5.5 MHz vL ip q3 iq

(ip) Plate current (T3 > T2 > T1)

I average T3 T2

I average T1

I average

Fig. 21.14. Effect of control and quadrature grids on the plate current of a quadrature grid detector.

Similarly when the signal frequency becomes less than the centre frequency the tuned circuit presents an inductive impedance and vq lags vL by less than 90°. The two signals permit longer conduction periods resulting in wider pulses with a higher average value of plate current.

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These pulses are integrated by the network RL and C3. Note that changes in the phase of vq result from frequency deviations in the FM signal. Similarly variations in the average value of plate current follow amplitude and frequency of the modulating audio signal. Thus the integrated output voltage at the plate of the tube gives the required audio output. C3 also acts as a plate bypass capacitor for 5.5 MHz signals. R1,C1 constitute the de-emphasis network. Tansistor Quadrature Detector. The transistor version of the quadrature detector employs three identical transistors. The circuit arrangement is shown in Fig. 21.15. Transistors Q1 and Q2 perform the same functions as the limiter and quadrature grids in a gated-beam tube. The sound IF signal is fed directly to the base of Q1, but is given a 90° phase-shift before applying it to the base of Q2. The phase shift is accomplished by driving the signal through a very small capacitor C2. The circuit formed by L and C3 does not produce any further phase-shift at 5.5 MHz because it is tuned to be resonant at this frequency.
+ VCC Section of a sound IC C1 From sound IF and limiter 20 pF 5 pF C2 R1 2K R2 1.7 K 300 W v1 v2 R5 4.7 K Q1 i1 Q2 i2 iR3 R3 R6 500 W Q3

Audio output C4 0.01 mF

L Bias voltage

C3

R4

C5

0.01 mF

Fig. 21.15. Circuit diagram of a solid state quadrature FM detector.

The sound IF signal attains sufficient amplitude before it is coupled to the detector circuit and the biasing is so set that the two transistors are driven into saturation during the positive swing of the input signal applied to each transistor. The emitters of the two transistors are grounded through a common resistor R3. Thus the output voltage that develops across R3 depends on the conduction of both Q1 and Q2. Since the voltage drop across R3 reverse biases the two transistors, the net bias automatically adjusts itself to keep the magnitude of peak current through R3 at nearly a constant value. However, the duration for which the two transistors conduct together varies and depends on the relative phase angle between the two input signals. As shown in the waveforms of Fig. 21.16, at the centre frequency where the phase difference between the two voltages is 90°, the current through R3 flows for a period T2 to develop an average current I2. However, when the signal frequency becomes more than the centre frequency, the phase angle between the two input voltages exceeds 90° and current

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through R3 flows for a longer period to develop an average current I1 which is greater than I2. When the input signal frequency become less than the centre frequency, the relative phase angle reduces to less than 90° and the magnitude of average current through R3 changes to I3 which is less than I2.
v1, v2 f < 5.5 MHz q < 90° Assumed saturation level v1, v2 f = 5.5 MHz q = 90° v1, v2 f > 5.5 MHz q > 90°

0

t

0

t

0

t

iR3 T3 I3 (T3 < T2 < T1) I2 T2 T1

I1

Fig. 21.16. Effect of frequency deviations on the average output current in a transistor quadrature detector. Note that cut-off bias for both Q1 and Q2 has been assumed as zero volt.

Since the relative phase change between the input voltages to Q1 and Q2 results from frequency deviations in the FM signal, the output voltage which develops across R3 due to variations in the average value of current through it follows the audio modulation. The audio voltage thus developed is in the input circuit of transistor Q3 where it is amplified before feeding it to the audio section. Note that capacitor C4 (0.01 µF) by passes all IF components to develop only audio voltage at the collector of Q3. It is interesting to note that in a gated-beam tube the limiter grid and quadrature grid act like two gates in series and when the phase angle between vL and vq exceeds 90°, the average output current decreases. However, in the transistor circuit, under similar conditions the average output current increases. This is because the two transistors Q1 and Q2 act as two gates in parallel to permit current flow when either of the two input signals is positive. Thus there is a 180° phase difference between the output currents of the two quadrature detectors for the same frequency swing. Since the audio signal is symmetrical about its centre zero axis there is no difference in the sound reproduced at the loudspeaker. 4. Differential Peak FM Detector In this system a new technique known as differential peak detection is used to develop audio voltage from the frequency modulated signal. Figure 21.17 (a) is the block diagram of this detector. It forms part of the IC, CA 3065 described in a later section of this chapter. The detector employs only a single tuning coil. The two peak detectors employ differential amplifier configurations with emitter followers at their inputs. The output circuits of the two detectors have identical RC networks of suitable

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time constant to provide peak detection of the signals fed to them. The high input impedance of the emitter followers isolates the detectors from the preceding sound IF section. The external frequency sensitive network comprises of a parallel LC circuit and a series capacitor. The voltage e1 obtained from the sound IF amplifier and limiter output is fed (actually applied at the input terminal of the IC) both to this frequency selective network and one of the peak detectors. At the centre frequency (fc = 5.5 MHz) the parallel circuit behaves like an inductor in series with capacitor C2. This combination provides a slight boost to the signal across C2 so as to make e2 equal to e1. The signal voltage e2 is applied at the input of the other peak detector. The two peak detectors develop proportionate dc voltages at their output terminals which feed into corresponding input terminals of the differential amplifier. Since e1 is equal to e2, the differential output voltage is zero in the absence of any modulation. As the intercarrier frequency swings to become low, the parallel network L1,C1 along with C2 in series, approaches its series resonant frequency. Thus e2 (across C2) attains a higher magnitude than e1.This in turn results in a negative output voltage from the differential amplifier. As the carrier frequency
Electronic attenuator

increases above fc, the parallel network (L1,C1) approaches its parallel resonant frequency and its impedance rises to make e2 less than e1. The differential amplifier output then swings to become positive. An analysis of the frequency response of e1 and e2 indicates that | e1 | – | e2| follows the normal ‘S’ curve (see Fig. 21.17 (b)) resulting in FM detection. The linearity of this detector is good and total harmonic distortion in the output is less than one percent. 5. Phase-Locked Loop FM Detector The latest in FM detection is the phase-locked loop detector shown in Fig. 21.18. It may be classed as an inductorless detector requiring no alignment. In this circuit a dc error voltage is developed by a phase detector that compares the phase of the input FM signal with that of a locally generated oscillator of the same frequency. The dc control voltage obtained at the output of low-pass filter is directly related to the degree of phase difference between the two signals. This control voltage which is returned to the voltage controlled oscillator (VCO) after amplification forces the oscillator to change its frequency thereby reducing the phase error.When the free-running frequency of the VCO becomes close to the input signal frequency, the system locks at this frequency and the VCO tracks the input FM signal. Since the dc control voltage will vary in step with the frequency deviations of the FM signal, it represents audio modulation. Thus the output from the dc amplifier is the demodulated audio signal. The IC contains a voltage amplifier (audio amplifier) the output of which can drive an audio power amplifier. The capacitor C1 is the de-emphasis capacitor connected to the corresponding pin of the monolithic IC package. For a noise-free reception, this FM detector requires a good limiter preceding the phase detector.
DC amplifier Phase comparator FM sound signal from the limiter output (VCO) Voltage controlled oscillator Low-pass filter C1 Limiter Audio voltage amplifier Audio output De-emphasis capacitor

21.7 SOUND SECTION INTEGRATED CIRCUITS
As already stated early hybrid receivers employed some form of a ratio detector for FM detection and vacuum tubes like ECL 82 (triode-pentode) for the audio section. The tubes were later replaced by transistors and a large number of circuits were developed for high quality sound reproduction. However, as soon as the integrated circuit technology was perfected, a large number of monolythic ICs were developed to perform the functions of sound IF limiter-amplifier, FM detector and audio pre-amplifier. In addition many ICs for the audio section are also now available. Out of the many sound section ICs that have been developed, TAA570, TBA750A and CA3065 are commonly used in modern circuits. All such ICs employ complex circuitry in order to minimize the use of transformer couplings and inductors.

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IC-TAA570 This IC employs a four-stage differential amplifier in the sound IF amplifier and limiter section. All the four stages receive their collector voltages through a multiemitter transistor which provides a self-limiting action on the voltage-swings at the collectors. All this results in very good noise suppression and excellent amplitude limiting. The FM detector section is a symmetrical phase detector which is a modified version of the transistor quadrature detector described earlier. The audio driver section is designed to give a maximum gain of about 80 db and can deliver an AF output equal to 1.8 volts for a frequency deviation of 50 KHz. Figure 21.19 shows necessary details of its external connections and the circuit of the associated audio output stage. This is commonly used in hybrid Television receivers.
Quadrature coil R2 C2 2
0.022 mF

+ P2 0.005 mF 500 K R L C10
330 K 2M

C6

0.022 mF

200 V T2 C9 Speaker

C4 0.1 mF T1 From sound take-off point

L1

10 9

1

Tone control 0.003 mF

R9

1.5 K

0.01 mF

V1

R6
0.01 mF

V2 1/2 PCL 86

200 pF 8

TAA 570 4 5 6 7 L2 20 H R1 + 15 V Zener diode

3 C8
5K

C11 P1 2 M Volume control 1/2 PCL 86 + 10 V

C1
0.22 mF

R3
22 K

R7

1K +4V 500 K R5 150 W

C5
0.1 mF

C3

0.022 mF

470 W C7

B+

Fig. 21.19. Sound section of a hybrid receiver employing IC TAA570.

The sound IF signal from the video detector is coupled through the tuned matching transformer T1 to the input terminals (pins 8 and 9) of the IC. Stabilized dc supply is fed at pins 3 and 5 through resistor R1 and decoupling capacitor C7. L2 is the choke and C1 the decoupling capacitor to the dc supply at pin 5. The components shown within the dotted-chain box consitute the quadrature circuit where L1,C2 form a tuned circuit resonant at 5.5 MHz and R2 is the damping resistance across the quadrature coil. De-emphasis is provided by capacitor C3 connected at pin 6 of the IC. The capacitors C4, C5 and C6 are bypass capacitors at the various pins of the integrated circuit. R3 is the load resistor of the audio amplifier transistor and its output is available at pin 3. The output from the IC is coupled to the voltage amplifier tube V1 (1/2 PCL86) through C8 and potentiometer P1 which is the volume control. Potentiometer P2, that is connected at the plate of V1 through a capacitor (C10) functions as tone control. The output tube V2(1/2 PCL86) provides power amplification and the sound output is delivered to the loudspeaker through T2 the

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matching output transformer. The series circuit formed by C9, R9 and connected across the primary circuit of the output transformer acts as tone correction network. Negative feedback is applied from the plate of V2 to that of V1 through R6 for improved tone control operation. R5, the cathode resistor of V2, has been left unbypassed to provide overall gain stability and improved frequency response of the audio amplifier. Resistance R7 in series with the control grid circuit is for suppression of any parasitic oscillations. IC-BEL CA 3065 This is a linear IC chip specially designed for the sound section of monochrome television receivers. Its circuit basically consists of (i) a regulated power supply, (ii) an IF amplifierlimiter, (iii) an FM detector, (iv) an electronic attenuator and buffer amplifier and (v) an audio driver. The schematic diagram of this IC is shown in Fig. 21.20.
RS B+ 0.047 µF 5 11 6 0.047 µF 100 K DC volume control

Fig. 21.20. Block diagram of the IC(BEL) CA 3065 in a typical circuit application. Note terminal 5 may be connected to any positive voltage through a suitable resistor (Rs) provided power rating of the IC is not exceeded.

(i) Regulated power supply. The input power supply to the IC is regulated through a combination of zener diodes. Separately regulated and decoupled dc supply is fed to different sections of the IC for stable circuit operation. (ii) IF amplifier-limiter. The single ended differential amplifier is the basic configuration for each of the three stages of the amplifier-limiter section of CA 3065. The quiescent currents of different transistors employed in the amplifiers are so adjusted that best possible limiting and AM rejection characteristics are obtained. The limiting amplifier is followed by

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an active low-pass filter which suppresses harmonics from the signal. The output of this filter feeds the FM detector. (iii) FM detector. A differential peak detector, the working principle of which was explained in an earlier section (see Fig. 21.17), is used for obtaining audio voltage from the frequency modulated signal. As shown in Fig. 21.20 voltage e1 obtained from the amplifierlimiter is applied at terminal 9 of the IC and thus fed simultaneously to the frequency selective network and one peak detector. The signal voltage e2 obtained at the output of the frequency selective network (L1, C1, and C2) is applied at the input (pin 10) of the other peak detector. The outputs from the two peak detectors feed into the differential amplifier, the output of which is the demodulated audio signal. (iv) Electronic attenuator and buffer amplifier. The output of the differential peak detector is fed to the electronic attenuator which performs the conventional function of the audio volume control. The stage employs a single ended differential cascade amplifier. The gain of this differential amplifier is controlled by varying the quiescent current of the transistors. This is achieved by changing the B+ voltage to one of its dc inputs, while the other dc input is kept at constant. An external variable resistance connected between terminal 6 of the IC and ground is used to vary the dc potential. Because there is no audio signal present across the variable resistance from terminal 6 to ground, there is no possibility of any noise or hum pickup in the audio signal. This type of volume control is called a dc volume control. The output of the attenuator is fed to a buffer amplifier which is an emitter follower and isolates the attenuator from the external load. The detected output is available at terminal 8 of the IC and the de-emphasis capacitor is connected at pin 7. (v) Audio driver. The output of the buffer stage feeds into an audio driver which is a simple common emitter amplifier sandwitched between two emitter followers for input isolation and low output impedance. Typical gain of this stage is 20 db. Terminal 13 is available for incorporating suitable tone control network.

21.8 AUDIO OUTPUT STAGE
The output from the IC (CA 3065) can drive a PCL84/PCL86 audio amplifier, a solid state complementary output stage, or a sound IC like the BEL CA810. A complete sound IF circuit employing CA 3065 along with a complementary transistor output amplifier is shown in Fig. 21.21. The output from the IC at pin 12 is fed to the driver transistor Q1 through coupling capacitor C3. Transistor Q1 (BC 148B) forms a boot-strapped driver stage which feeds the matched output pair Q2 and Q3 (AC 187/01, AC 188/01). The output transistors drive the loudspeaker (8 ohms) through the electrolytic capacitor C4, R1 and R2 and thermistor (Th) provide necessary temperature compensation in the biasing circuit of the output pair. The stage operates class AB to avoid cross-over distortion. Typical data of the audio amplifier is as under: Supply voltage No signal current Current drain at 2.5 W 15 V 55 mA 300 mA

Audio IC CA 810 The IC BEL CA 810 has been specially developed for the audio section of television receivers. This audio amplifier as illustrated in Fig. 21.22 is capable of delivering 4 W output power into an 8 ohm load. Good frequency response and high input impedance are some of the salient features of this circuit. Its typical performance characteristics are as under: (1) Supply voltage (2) Maximum output power at 10% THD (3) Input sensitivity at 50 mW output power (4) Input sensi