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Abstract:

An RF transmitter comprises a capacitive energy storage, an output stage
and a switching circuit with an open state and a closed state. The
capacitive energy storage forms with the antenna when connected thereto a
resonance circuit with a resonance frequency and a quality factor. The
output stage provides an electric transmission signal to the resonance
circuit. The switching circuit comprises a first transistor for switching
between the open state and the closed state and is connected to an
antenna output terminal through a capacitance formed by the capacitive
energy storage such that the maximum signal voltage occurring across the
switching circuit in its open state is lower than the maximum signal
voltage occurring across the antenna output terminals. The transmitter is
adapted to alter the quality factor by changing the series resistance of
the switching circuit in its closed state.

Claims:

1. A transmitter, comprising: an antenna output terminal for connecting
to a corresponding antenna terminal of an electrically short antenna for
transmission of an electromagnetic signal; a capacitive energy storage
having a total effective capacitance and configured to form with the
electrically short antenna when connected to said electrically short
antenna a resonance circuit with a resonance frequency and a quality
factor; an output stage configured to provide an electric transmission
signal to the resonance circuit; a switching circuit having an open state
and a closed state, the switching circuit switching between the open
state and the closed state and being connected to the antenna output
terminal through a capacitance formed by the capacitive energy storage
such that a maximum signal voltage occurring across the switching circuit
in its open state is lower than a maximum signal voltage occurring at the
antenna output terminal; and a battery supplying power to the transmitter
via a wired connection, wherein the transmitter is adapted to alter the
quality factor by changing a series resistance of the switching circuit
by changing the switching circuit to its alternate state.

2. The transmitter according to claim 1, wherein the capacitive energy
storage includes a plurality of capacitive energy storage circuits, each
of the capacitive energy storage circuits includes at least two tuning
switches controlled by a controller and has an effective capacitance
Ci based on an open/close state of its tuning switches, the total
effective capacitance of the capacitive energy storage is a sum of the
effective capacitances Ci of the capacitive energy storage circuits,
and the controller is configured to change the effective capacitance
Ci of a minimum number of capacitive energy storage circuits to
achieve a target total effective capacitance of the capacitive energy
storage.

3. The transmitter according to claim 2, wherein said plurality of
capacitive energy storage circuits includes a first set of capacitive
energy storage circuits and a second set of capacitive energy storage
circuits, each capacitive energy storage circuit of the first set
includes a capacitor with capacitance C, and each capacitive energy
storage circuit of the second set includes a capacitor with capacitance
1/2 C.

4. The transmitter according to claim 3, wherein said plurality of
capacitive energy storage circuits further includes a third set of
capacitive energy storage circuits, and each capacitive energy storage
circuit of the third set includes a capacitor with capacitance 1/4 C.

5. The transmitter according to claim 1, wherein the switching circuit
includes a first transistor for switching between the open state and the
closed state.

6. The transmitter according to claim 5 and further adapted to change the
series resistance of the switching circuit in its closed state by
controlling a gate voltage applied to the first transistor thereby
changing the intrinsic on-resistance of the first transistor.

7. The transmitter according to claim 6 wherein the switching circuit
further comprises a second transistor arranged in series with the first
transistor and wherein the transmitter is further adapted to change the
series resistance of the switching circuit in its closed state by
controlling a gate voltage applied to the second transistor.

8. The transmitter according to claim 5 wherein the switching circuit
further comprises a second transistor arranged in series with the first
transistor and wherein the transmitter is further adapted to change the
series resistance of the switching circuit in its closed state by
controlling a gate voltage applied to the second transistor.

9. The transmitter according to claim 8, wherein the switching circuit
further comprises a resistor connected in parallel with the second
transistor.

10. The transmitter according to claim 5, wherein the first transistor is
an output transistor of the output stage.

11. The transmitter according to claim 1, further comprising: a bandwidth
controller adapted to control the series resistance of the switching
circuit in its closed state.

12. The transmitter according to claim 11, further comprising: a
calibration unit adapted to calibrate the bandwidth controller.

13. The transmitter according to claim 12, wherein the calibration unit
is configured to have the output stage provide an electric transmission
signal to the resonance circuit and to determine the quality factor in
dependence on the electric transmission signal.

14. The transmitter according to claim 13, wherein the calibration unit
comprises a current measurement device for determining a current through
an output transistor of the output stage and is adapted to determine the
quality factor in dependence on the determined current.

15. The transmitter according to claim 1, further comprising: a receiver
circuit connected to receive signals from the resonance circuit.

16. The transmitter according to claim 1, wherein the transmitter further
is adapted to alter the resonance frequency by switching between the open
and closed states of the switching circuit.

17. The transmitter according to claim 1, wherein the switching circuit
includes an electronic switch for switching between the open state and
the closed state.

18. The transmitter according to claim 1, wherein the switching circuit
is comprised in the output stage.

19. A portable apparatus, comprising: an electrically short antenna; and
a transmitter connected to the electrically short antenna, the
transmitter including an antenna output terminal for connecting to a
corresponding antenna terminal of the electrically short antenna for
transmission of an electromagnetic signal; a capacitive energy storage
having a total effective capacitance and configured to form with the
electrically short antenna when connected to said electrically short
antenna a resonance circuit with a resonance frequency and a quality
factor; an output stage configured to provide an electric transmission
signal to the resonance circuit; and a switching circuit having an open
state and a closed state, the switching circuit switching between the
open state and the closed state and being connected to the antenna output
terminal through a capacitance formed by the capacitive energy storage
such that a maximum signal voltage occurring across the switching circuit
in its open state is lower than a maximum signal voltage occurring at the
antenna output terminal; and a battery within the portable apparatus
supplying power to the transmitter, wherein the transmitter is adapted to
alter the quality factor by changing a series resistance of the switching
circuit by changing the switching circuit to its alternate state.

20. The portable apparatus according to claim 19, wherein the capacitive
energy storage includes a plurality of capacitive energy storage
circuits, each of the capacitive energy storage circuits includes at
least two tuning switches controlled by a controller and has an effective
capacitance Ci based on an open/close state of its tuning switches,
the total effective capacitance of the capacitive energy storage is a sum
of the effective capacitances Ci of the capacitive energy storage
circuits, and the controller is configured to change the effective
capacitance Ci of a minimum number of capacitive energy storage
circuits to achieve a target total effective capacitance of the
capacitive energy storage.

21. A hearing device, comprising: an electrically short antenna; and a
transmitter connected to the electrically short antenna, the transmitter
including an antenna output terminal for connecting to a corresponding
antenna terminal of the electrically short antenna for transmission of an
electromagnetic signal; a capacitive energy storage having a total
effective capacitance configured to form with the electrically short
antenna when connected to said electrically short antenna a resonance
circuit with a resonance frequency and a quality factor; an output stage
configured to provide an electric transmission signal to the resonance
circuit; and a switching circuit having an open state and a closed state,
the switching circuit switching between the open state and the closed
state and being connected to the antenna output terminal through a
capacitance formed by the capacitive energy storage such that a maximum
signal voltage occurring across the switching circuit in its open state
is lower than a maximum signal voltage occurring at the antenna output
terminal; and a battery within the hearing device supplying power to the
transmitter, wherein the transmitter is adapted to alter the quality
factor by changing a series resistance of the switching circuit by
changing the switching circuit to its alternate state.

22. The hearing device according to claim 21, wherein the capacitive
energy storage includes a plurality of capacitive energy storage
circuits, each of the capacitive energy storage circuits includes at
least two tuning switches controlled by a controller and has an effective
capacitance Ci based on an open/close state of its tuning switches,
the total effective capacitance of the capacitive energy storage is a sum
of the effective capacitances Ci of the capacitive energy storage
circuits, and the controller is configured to change the effective
capacitance Ci of a minimum number of capacitive energy storage
circuits to achieve a target total effective capacitance of the
capacitive energy storage.

Description:

[0001] This application is a Continuation of co-pending application Ser.
No. 13/683,132, filed on Nov. 21, 2012, which claims the benefit under 35
U.S.C. §119(e) of U.S. Provisional Application No. 61/563,618 filed
on Nov. 25, 2011 and under 35 U.S.C. §119(a) of patent application
Ser. No. 11/190,731.7 filed in Europe on Nov. 25, 2011. The entire
content of all of the above applications is hereby incorporated by
reference.

TECHNICAL FIELD

[0002] The present invention relates to a radio-frequency (RF) transmitter
for transmitting electromagnetic signals via an electrically short
antenna.

[0003] The invention may e.g. be useful in wireless communication
involving portable and/or battery-operated apparatuses, such as wireless
communication between hearing devices and auxiliary devices. Hearing
devices may e.g. be hearing aids for compensating for a hearing-impaired
person's loss of hearing capability or listening devices for augmenting a
normal-hearing person's hearing capability.

BACKGROUND ART

[0004] RF antennas for transmission of electromagnetic signals are
preferably designed to have a size of at least one quarter of the
wavelength of the transmitted signal, since this generally allows high
antenna efficiency, wide bandwidth and substantially real input
impedance. However, many apparatuses do not have room for an antenna
large enough to satisfy this condition. For an RF signal with a frequency
of e.g. 100 MHz, one quarter of the wavelength equals 0.75 m. It is thus
common to utilise antennas that are considerably smaller than one quarter
of the wavelength. Such antennas are generally referred to as
"electrically short" or "electrically small" antennas. Electrically short
antennas inherently exhibit low radiation resistance and low efficiency.
Their efficiency may be increased by reducing resistive losses in the
antenna and the associated circuits, which, however, increases the
quality factor (Q) of the antenna so that the bandwidth decreases. At a
typical quality factor of 50, the 3-dB bandwidth of the antenna is 2% of
the centre frequency.

[0005] The co-pending patent application EP 11 184 079.9 and the
corresponding provisional patent application U.S. 61/543,821 disclose a
transmitter for transmitting an electromagnetic signal via an
electrically short antenna. The transmitter comprises a reactive energy
storage, which forms a resonance circuit with the antenna. The
transmitter further comprises a number of output stages, which provide an
electric transmission signal to the resonance circuit. The electric
transmission signal is frequency- and/or phase-modulated with an
information signal and thus has an instantaneous frequency that varies in
dependence on the information signal. In order to allow efficient
transmission of electromagnetic signals with a bandwidth exceeding that
of the resonance circuit, the transmitter dynamically changes the
resonance frequency of the resonance circuit by changing the effective
reactance of the energy storage in dependence on the information signal.
The electronic switch elements used for changing the effective reactance
of the energy storage must be protected against high voltages occurring
at the antenna terminals. Therefore, the electronic switch elements are
not connected directly to the antenna terminals but instead indirectly
via voltage dividers formed by serially connected capacitors. The outputs
of the individual output stages are similarly connected to the antenna
terminals via capacitors forming impedance transformers yielding higher
signal voltages at the antenna terminals than at the outputs of the
output stages.

[0006] In an embodiment disclosed in the above mentioned co-pending patent
applications, the transmitter further comprises a receiver circuit
connected to the resonance circuit, which allows the transmitter to
function as a half-duplex transceiver. In the receive mode, the
transmitter can, however, not take advantage of a dynamic change of the
resonance frequency of the resonance circuit, since this would require
knowledge of the instantaneous frequency of the received signal before
receiving it, and this information is obviously only available after
receiving the signal. An alternative method for temporarily increasing
the bandwidth of the resonance circuit and thus of the receiver is to
temporarily add resistive loads to the antenna terminals, e.g. by means
of electronic switch elements. This method is not disclosed in the above
mentioned applications, but is generally known in the prior art, e.g.
from patent application FR 2 911 805. The same method could also be used
for temporarily increasing the bandwidth of the resonance circuit during
transmission, e.g. in order to allow transmission of signals with a
bandwidth exceeding the bandwidths achievable by the dynamic change of
the resonance frequency of the resonance circuit. However, adding
resistive loads directly to the antenna terminals by means of switches
would subject the switches to possibly large signal voltages occurring
across the antenna terminals, which could damage the switches or shorten
their life-time.

DISCLOSURE OF INVENTION

[0007] It is an object of the present invention to provide an RF
transmitter, which allows temporarily altering the bandwidth of the
resonance circuit without the above mentioned disadvantages.

[0008] Further objects are to provide a portable apparatus and a hearing
aid comprising such a transmitter connected to an electrically short
antenna.

[0009] A further object is to provide a use of such a transmitter for
transmission of electromagnetic signals via an electrically short antenna
without the above mentioned disadvantages.

[0010] These and other objects of the invention are achieved by the
invention defined in the independent claims and as explained in the
following description. Further objects of the invention are achieved by
the embodiments defined in the dependent claims and in the detailed
description of the invention.

[0011] The invention is based on an RF transmitter with two antenna output
terminals for connecting to corresponding antenna terminals of an
electrically short antenna for transmission of an electromagnetic signal.
The transmitter comprises a capacitive energy storage, an output stage
and a switching circuit with an open state and a closed state. The
capacitive energy storage is adapted to form with the antenna when
connected thereto a resonance circuit with a resonance frequency and a
quality factor. The output stage is adapted to provide an electric
transmission signal to the resonance circuit. The switching circuit
comprises a first transistor for switching between the open state and the
closed state and is connected to an antenna output terminal through a
capacitance formed by the capacitive energy storage such that the maximum
signal voltage occurring across the switching circuit in its open state
is lower than the maximum signal voltage occurring across the antenna
output terminals. In order to allow temporarily altering the bandwidth of
the resonance circuit, the transmitter is adapted to alter the quality
factor by changing the series resistance of the switching circuit in its
closed state.

[0012] The achieved solution enables temporarily altering the quality
factor without the above mentioned disadvantages of the prior art. In
particular, dedicated loss resistances need not to be added to the
transmitter, and switches do not need to be added where they could be
subjected to high antenna voltages. The achieved solution enables a very
compact transmitter design with relatively high overall power efficiency.

[0013] Preferably, the transmitter is further adapted to change the series
resistance of the switching circuit in its closed state by controlling a
gate voltage applied to the first transistor thereby changing the
intrinsic on-resistance of the first transistor. This allows a
particularly simple way of controlling the quality factor.

[0014] Preferably, the switching circuit further comprises a second
transistor arranged in series with the first transistor, and the
transmitter is further adapted to change the series resistance of the
switching circuit in its closed state by controlling a gate voltage
applied to the second transistor.

[0015] Preferably, the switching circuit further comprises a resistor
connected in parallel with the second transistor. This allows achieving
predefined quality factors.

[0016] Preferably, the transmitter further is adapted to temporarily alter
the resonance frequency by switching between the open and closed states
of the switching circuit.

[0017] Preferably, the switching circuit is comprised in the output stage.
Preferably, the first transistor is an output transistor of the output
stage.

[0018] Preferably, the transmitter further comprises a bandwidth
controller adapted to control the series resistance of the switching
circuit in its closed state.

[0020] Preferably, the calibration means are adapted to have the output
stage provide an electric transmission signal to the resonance circuit
and to determine the quality factor in dependence on the electric
transmission signal. This allows a simple and robust way of calibrating
the bandwidth controller.

[0021] Preferably, the calibration means comprises current measurement
means for determining a current through an output transistor of the
output stage and is adapted to determine the quality factor in dependence
on the determined current. This allows for accurate calibration.

[0022] The transmitter may comprise a receiver circuit connected to
receive signals from the resonance circuit. This allows the transmitter
to operate as a transceiver and further facilitates automatic tuning and
calibration of the transmitter.

[0023] The transmitter may preferably be incorporated in a portable
apparatus, e.g. a hearing device.

[0024] The transmitter may be used for transmitting an electromagnetic
signal via an electrically short antenna.

[0025] In the present context, a "hearing device" refers to a device, such
as e.g. a hearing aid or an active ear-protection device, which is
adapted to improve or augment the hearing capability of an individual by
receiving acoustic signals from the individual's surroundings, modifying
the acoustic signals electronically and providing audible signals to at
least one of the individual's ears. Such audible signals may e.g. be
provided in the form of acoustic signals radiated into the individual's
outer ears, acoustic signals transferred as mechanical vibrations to the
individual's inner ears through the bone structure of the individual's
head and/or electric signals transferred directly or indirectly to the
cochlear nerve of the individual. The hearing device may be configured to
be worn in any known way, e.g. as a unit arranged behind the ear with a
tube leading radiated acoustic signals into the ear canal or with a
speaker arranged close to or in the ear canal, as a unit entirely or
partly arranged in the pinna and/or in the ear canal, as a unit attached
to a fixture implanted into the skull bone, etc. More generally, a
hearing device comprises an input transducer for receiving an acoustic
signal from an individual's surroundings and providing a corresponding
electric input signal, a signal processing circuit for processing the
electric input signal and an output transducer for providing an audible
signal to the individual in dependence on the processed signal.

[0026] A "hearing system" refers to a system comprising one or two hearing
devices, and a "binaural hearing system" refers to a system comprising
one or two hearing devices and being adapted to provide audible signals
to both of the individual's ears. Hearing systems or binaural hearing
systems may further comprise "auxiliary devices", which communicate with
the hearing devices and affect and/or benefit from the function of the
hearing devices. Auxiliary devices may be e.g. remote controls, audio
gateway devices, mobile phones, public-address systems, car audio systems
or music players. Hearing devices, hearing systems or binaural hearing
systems may e.g. be used for compensating for a hearing-impaired person's
loss of hearing capability or augmenting a normal-hearing person's
hearing capability.

[0027] As used herein, the singular forms "a", "an", and "the" are
intended to include the plural forms as well (i.e. to have the meaning
"at least one"), unless expressly stated otherwise. It will be further
understood that the terms "has", "includes", "comprises", "having",
"including" and/or "comprising", when used in this specification, specify
the presence of stated features, integers, steps, operations, elements
and/or components, but do not preclude the presence or addition of one or
more other features, integers, steps, operations, elements, components
and/or groups thereof. It will be understood that when an element is
referred to as being "connected" or "coupled" to another element, it can
be directly connected or coupled to the other element, or intervening
elements may be present, unless expressly stated otherwise. As used
herein, the term "and/or" includes any and all combinations of one or
more of the associated listed items. The steps of any method disclosed
herein do not have to be performed in the exact order disclosed, unless
expressly stated otherwise.

BRIEF DESCRIPTION OF THE DRAWINGS

[0028] The invention will be explained in more detail below in connection
with preferred embodiments and with reference to the drawings in which:

[0029] FIG. 1 shows an embodiment of a transmitter according to the
invention,

[0035] FIG. 7 shows an embodiment of a hearing device comprising the
transmitter of FIG. 1.

[0036] The figures are schematic and simplified for clarity, and they just
show details, which are essential to the understanding of the invention,
while other details are left out. Throughout, like reference numerals
and/or names are used for identical or corresponding parts.

[0037] Further scope of applicability of the present invention will become
apparent from the detailed description given hereinafter. However, it
should be understood that the detailed description and specific examples,
while indicating preferred embodiments of the invention, are given by way
of illustration only, since various changes and modifications within the
spirit and scope of the invention will become apparent to those skilled
in the art from this detailed description.

MODE(S) FOR CARRYING OUT THE INVENTION

[0038] The transmitter 1 shown in FIG. 1 comprises a capacitive energy
storage 2, a power amplifier 3, a modulator 4, a receiver circuit 5, a
demodulator 6, a control unit 7, a bandwidth controller 8, two antenna
output terminals 10, 11 for providing an electric transmission signal, a
clock terminal 12 for receiving a clock signal, a data input terminal 13
for receiving an information signal D (see FIG. 6), a mode input terminal
14 for receiving a mode control signal, a tuning input terminal 15 for
receiving a tuning control signal, a receive data terminal 16 for
providing a receive data signal and a calibration data terminal 17 for
providing a calibration data signal.

[0039] The energy storage 2 comprises 257 storage circuits 20, one of
which is shown in the figure, as well as a number of capacitors 33, 34,
52, 53 that are also comprised in respectively the power amplifier 3 and
the receiver circuit 5. Each storage circuit 20 comprises three tuning
capacitors 21, 22, 23 and two tuning switches 24, 25. The tuning
capacitors 21, 22, 23 are connected in series between the antenna output
terminals 10, 11, thereby defining a first and a second node 26, 27
between them. The first tuning switch 24 is connected between the first
node 26 and signal ground. The second tuning switch 25 is connected
between the second node 27 and signal ground. In each of the tuning
switches 24, 25, a control input, which controls opening and closing of
the respective tuning switch 24, 25, is connected to receive a respective
tuning control signal through a respective tuning control line 28, 29
from the control unit 7. The transmitter 1 has a separate set of tuning
control lines 28, 29 for each storage circuit 20.

[0040] The power amplifier 3 comprises seven identical dual amplifier
circuits 30, one of which is shown in the figure. Each dual amplifier
circuit 30 comprises two digital output stages 31, 32 and two output
capacitors 33, 34. Each digital output stage 31, 32 has a signal input
connected to receive a respective transmit signal from a respective
output of the modulator 4 through a respective signal line 35, 36 and a
control input connected to receive a common quality factor control signal
from the bandwidth controller 8 through a quality factor control line 37.
Each digital output stage 31, 32 further has an output connected to
provide an amplified output signal corresponding to the transmit signal
on its input to a respective one of the antenna output terminals 10, 11
through a respective one of the output capacitors 33, 34. The transmitter
1 has a separate set of signal lines 35, 36 for each dual amplifier
circuit 30. The quality factor control line 37 is common for all dual
amplifier circuits 30. The output capacitors 33, 34 are also comprised in
the capacitive energy storage 2.

[0041] The modulator 4 is connected to receive the clock signal from the
clock terminal 12, the mode control signal from the mode input terminal
14 and a transmit control signal through a transmit control line 45 from
the control unit 7. The modulator 4 is further connected to provide the
transmit signals to each of the signal lines 35, 36 for all of the dual
amplifier circuits 30.

[0042] The receiver circuit 5 comprises a low-noise amplifier 51, two
input capacitors 52, 53 and two mode switches 54, 55. Two complementary
inputs 56, 57 of the low-noise amplifier 51 are each connected to a
respective one of the antenna output terminals 10, 11 through a
respective one of the input capacitors 52, 53. The first mode switch 54
is connected between the first input 56 and signal ground, and the second
mode switch 55 is connected between the second input 57 and signal
ground. In each of the mode switches 54, 55, a control input, which
controls opening and closing of the respective mode switch 54, 55, is
connected to receive the mode control signal from the mode input terminal
14. Two outputs of the low-noise amplifier 51 are connected to provide a
differential receiver output signal to the demodulator 6 through two
receiver output lines 61, 62. The low-noise amplifier 51 is further
connected to receive the mode control signal from the mode input terminal
14. The input capacitors 52, 53 are also comprised in the capacitive
energy storage 2.

[0043] The demodulator 6 is connected to receive the differential receiver
output signal from the receiver output lines 61, 62 as well as a receive
control signal through a receive control line 65 from the control unit 7.
The demodulator 6 is further connected to provide the receive data signal
to the receive data terminal 16 and the calibration data signal to the
calibration data terminal 17.

[0044] The control unit 7 is connected to receive the information signal D
from the data input terminal 13, the mode control signal from the mode
input terminal 14 and the tuning control signal from the tuning input
terminal 15. The control unit 7 is further connected to provide
individual tuning control outputs to each of the tuning control lines 28,
29 for all of the storage circuits 20, the transmit control signal to the
transmit control line 45 and the receive control signal to the receive
control line 65.

[0045] The bandwidth controller 8 is connected to receive the transmit
control signal through the transmit control line 45 from the control unit
7. The bandwidth controller 8 is further connected to provide the quality
factor control signal to the quality factor control line 37 to the dual
amplifier circuits 30.

[0046] A loop antenna 100 external to the transmitter 1 comprises a
conductor with one or more windings and two terminals 101, 102, which are
each connected to a respective one of the antenna output terminals 10, 11
of the transmitter 1. The antenna 100 forms a complex impedance Z between
the antenna output terminals 10, 11, which is modelled by a series
connection of an inductor L and a resistor R. The inductance L, i.e. the
imaginary part of the impedance Z, is primarily determined by the
inductance of the windings of the loop antenna 100, but is also
influenced by parasitic capacitances. The resistance R, i.e. the real
part of the impedance Z, is primarily determined by the inherent
resistance of the conductor and the magnetic induction losses. The
resistance R also comprises the radiation resistance of the antenna 100,
which, however, is relatively small in an electrically short antenna.

[0047] The mode control signal, which must be supplied to the mode input
terminal 14 by an external unit, e.g. a signal processor 704 (see FIG.
7), enables one of a transmit mode, in which an electromagnetic signal TX
(see FIG. 6) may be transmitted by the transmitter 1, a calibration mode,
which allows calibration and tuning of the transmitter 1, and a receive
mode, in which an electromagnetic signal may be received by the
transmitter 1. Each of the modulator 4, the receiver circuit 5, the
demodulator 6 and the control unit 7 receives the mode control signal and
reacts to the enabled mode as described in the following.

[0048] In the transmit mode and in the calibration mode, the modulator 4
is enabled. Furthermore, the mode switches 54, 55 are closed to connect
the inputs 56, 57 of the low-noise amplifier 51 to signal ground, thus
protecting them from high voltages provided by the power amplifier 3 to
the antenna output terminals 10, 11. The mode switches 54, 55 are
implemented as electronic switch elements, e.g. field-effect transistors
(FET), which inherently possess a very small, but nevertheless non-zero
resistance and/or capacitance in their closed state. Each mode switch 54,
55 thus forms a voltage divider with the respective input capacitor 52,
53, which allows the low-noise amplifier 51 to receive signals from the
antenna output terminals 10, 11, however strongly attenuated. In the
calibration mode, the low-noise amplifier 51 is enabled to allow
amplification of the attenuated signals. The demodulator 6 measures the
amplitude and the phase of the amplified signal and provides the
measurement results in the calibration data signal. In the transmit mode,
the low-noise amplifier 51 is disabled so that its outputs as well as the
receive data signal and the calibration data signal from the demodulator
6 are idle.

[0049] In the receive mode, the modulator 4 is disabled so that the
transmit signals to the power amplifier 3 are idle. The outputs of the
digital output stages 31, 32 are thus each tied to a power supply
voltage, so that the output capacitors 33, 34 remain connected as
substantive loads to the respective antenna output terminal 10, 11.
Furthermore, the mode switches 54, 55 are open, so that the low-noise
amplifier 51 may receive weak signals from the antenna output terminals
10, 11. The low-noise amplifier 51 amplifies the received signal and
provides the amplified signals as a differential receiver output signal
to the demodulator 6, which demodulates the differential receiver output
signal in accordance with the information received in the receive control
signal and provides demodulated data in the receive data signal.

[0050] The energy storage 2, which comprises the storage circuits 20 as
well as the output capacitors 33, 34 and the input capacitors 52, 53,
forms an effective storage capacitance C between the antenna output
terminals 10, 11. The effective storage capacitance C further comprises
contributions from parasitic capacitances in the transmitter 1, which may
be caused by e.g. circuits for protection against electrostatic
discharges (ESD). The effective capacitance C depends mainly on the
states of the tuning switches 24, 25 in all of the storage circuits 20
and to a lesser degree on the states of the mode switches 54, 55, since
the input capacitors 52, 53 are chosen relatively small.

[0051] In the capacitive energy storage 2, 255 of the 257 storage circuits
20 are identical and form a linearly-coded--or thermometer-coded--array
200 (see FIG. 2) of storage circuits 20. The function of the
thermometer-coded array 200 is illustrated in FIG. 2. The individual
storage circuits 20 are numbered from 1 to N. In the shown embodiment, N
equals 255. Each storage circuit 20 is illustrated as a rectangle, the
width of which is proportional to the effective capacitance Ci of
the storage circuit 20, i.e. the contribution of the storage circuit 20
to the effective storage capacitance C. The effective capacitance of the
entire thermometer-coded array 200 is thus proportional to the area
covered by the rectangles 20. The effective capacitance Ci of each
storage circuit 20 may be changed by operating, i.e. opening or closing,
one or both of the tuning switches 24, 25 and thus reconfiguring the
circuit formed by the tuning capacitors 21, 22, 23. At any time, the
thermometer-coded array 200 comprises a (possibly empty) set 201 of
storage circuits 20 momentarily having a higher effective capacitance
Ci and a (possibly empty) set 202 of storage circuits 20 momentarily
having a lower effective capacitance Ci.

[0052] FIGS. 2a and 2b show the thermometer-coded array 200 in
respectively a first and a second state. Since the set 201 of storage
circuits 20 having a higher effective capacitance Ci is larger in
the first state than in the second state, the effective capacitance of
the entire thermometer-coded array 200 is also larger in the first state
than in the second state. When changing the effective capacitance of the
entire thermometer-coded array 200 from one state (e.g. the first state)
to another state (e.g. the second state), the control unit 7 operates the
tuning switches 24, 25 such that the effective capacitance Ci of a
minimum of storage circuits 20 changes. This may be achieved by only
adding storage circuits 20 to the set 201 of storage circuits 20 having a
higher effective capacitance Ci when increasing the effective
storage capacitance C, and only removing storage circuits 20 from the set
201 of storage circuits 20 having a higher effective capacitance Ci
when decreasing the effective storage capacitance C. In other words, when
increasing the effective storage capacitance C, none of the storage
circuits 20 in the thermometer-coded array 200 are reconfigured to
decrease their effective capacitance Ci, and vice versa. This allows
keeping the electric noise caused by the switching at a minimum, which
also allows reducing artifacts in the transmitted electromagnetic signal.

[0053] The effective capacitance of the entire thermometer-coded array 200
may be changed with a resolution corresponding to the possible change of
the effective capacitance Ci of a single storage circuit 20. An e.g.
four times finer resolution across the same range could thus be achieved
by using four times the number of storage circuits 20, each with
quartered capacitances of the tuning capacitors 21, 22, 23. Instead--in
order to save space and power in the transmitter 1, two of the 257
storage circuits 20 are implemented with respectively halved and
quartered capacitances of the tuning capacitors 21, 22, 23 compared to
the storage circuits 20 in the thermometer-coded array 200. These two
storage circuits 20 form a binary-coded array of storage circuits 20,
which allows a four times finer resolution at the cost of an only
slightly increased switching noise.

[0054] FIG. 3 shows details of the bandwidth controller 8 and of the
digital output stages 31, 32. The output stages 31, 32 are identical and
the shown output stage 31, 32 is thus representative of any of these. The
output stage 31, 32 comprises an N-channel FET 311, a P-channel FET 312,
two inverters 313, 314 and an output terminal 315. The drain terminal of
the N-channel FET 311 is connected to the output terminal 315, its source
terminal to signal ground and its gate terminal to the output of the
first inverter 313. The drain terminal of the P-channel FET 312 is
connected to the output terminal 315, its source terminal to a common
positive power supply voltage Vdd and its gate terminal to the output of
the second inverter 314. The inputs of the first and the second inverters
are connected to receive the transmit signal from the modulator 4 through
the signal line 35, 36, and their negative power supply terminals are
connected to signal ground. The positive power supply terminal of the
first inverter 313 is connected to the quality factor control line 37
from the bandwidth controller 8, whereas the positive power supply
terminal of the second inverter 314 is connected to the common positive
power supply voltage Vdd. The output terminal 315 is connected to one of
the antenna terminals 10, 11 through the output capacitor 33, 34. The
FETs 311, 312 and the inverters 313, 314 form a digital half-bridge
push-pull output stage.

[0055] The bandwidth controller 8 comprises a microcontroller 81, a
calibration memory 82 and a digital-to-analog converter 83. The
microcontroller 81 is connected to receive the transmit control signal
from the control unit 7 through the transmit control line 45 and to
provide a memory address to the calibration memory 82. The calibration
memory 82 is connected to provide a data output to the digital-to-analog
converter 83, which is connected to provide the quality factor control
signal to the quality factor control line 37.

[0056] A high level of the transmit signal on the signal line 35, 36 will
cause the inverters 313, 314 to apply low voltages, i.e. voltages
slightly above the signal ground potential, to the gate terminals of the
FETs 311, 312, which will cause the N-channel FET 311 to have a high
drain-source resistance, i.e. to be "off", and the P-channel FET 312 to
have a low drain-source resistance, i.e. to be "on", thus providing a
high level on the output terminal 315. In the following, the drain-source
resistance of an FET, when it is in a state with a low drain-source
resistance, is referred to as the intrinsic on-resistance of the FET. The
voltage applied to the P-channel FET 312 is low enough to ensure a
minimum intrinsic on-resistance in the P-channel FET 312. Conversely, a
low level of the transmit signal will cause the inverters 313, 314 to
apply high voltages to the gate terminals of the FETs 311, 312, which
will cause the N-channel FET 311 to be on and the P-channel FET 312 to be
off, thus providing a low level on the output terminal 315. Since the
second inverter 314 is supplied with power from the common positive power
supply voltage Vdd, the gate voltage applied to the P-channel FET 312 is
only slightly below Vdd and thus high enough to ensure that the P-channel
FET 312 has a maximum drain-source resistance. The gate voltage applied
to the N-channel FET 311 is, however, slightly below the voltage on the
quality factor control line 37. Since the intrinsic on-resistance in an
FET depends on the gate-source voltage, the intrinsic on-resistance in
the N-channel FET 311 may thus be controlled by varying the voltage on
the quality factor control line 37.

[0057] In operation, the microcontroller 81 decodes the required quality
factor indicated in the transmit control signal from the control unit 7
(see further below) and computes a corresponding memory address for the
calibration memory 82. The calibration memory 82 outputs the content of
the memory cell with the indicated address to the digital-to-analog
converter 83, which converts the data into a voltage on the quality
factor control line 37 and thus determines the intrinsic on-resistance in
the N-channel FET 311. When the level of the transmit signal is low, the
intrinsic on-resistance in the N-channel FET 311 thus forms a resistive
load to the antenna 100 through the output capacitor 34 and thus affects
the quality factor Q and the bandwidth of the resonance circuit 2, 100.
The bandwidth controller 8 may thus alter the quality factor Q and the
bandwidth of the resonance circuit 2, 100 by varying the voltage on the
quality factor control line 37.

[0058] In the embodiment shown in FIG. 3, the N-channel FET 311 thus forms
a switching circuit 9 with an open state, i.e. when the N-channel FET 311
is off, and a closed state, i.e. when the N-channel FET 311 is on. The
series resistance of the switching circuit 9 in its closed state is thus
varied by varying the intrinsic on-resistance in the N-channel FET 311.

[0059] In the alternative embodiment shown in FIG. 4, the gate terminals
of the FETs 311, 312 are connected to the output of a common inverter
316, which is supplied with power from the common positive power supply
voltage Vdd and signal ground. A further N-channel FET 401 is connected
in series with the N-channel FET 311 such that it forms a variable
resistance between the source terminal of the N-channel FET 311 and
signal ground. The gate terminal of the further N-channel FET 401 is
connected to the quality factor control line 37 from the bandwidth
controller 8. In this embodiment, varying the voltage on the quality
factor control line 37 causes the drain-source resistance in the further
N-channel FET 401 to vary. Similarly to the embodiment shown in FIG. 3,
when the level of the transmit signal is low, the intrinsic on-resistance
in the further N-channel FET 401 forms a resistive load to the antenna
100 through the N-channel FET 311 and the output capacitor 34 and thus
affects the quality factor Q and the bandwidth of the resonance circuit
2, 100. The bandwidth controller 8 may thus alter the quality factor Q
and the bandwidth of the resonance circuit 2, 100 by varying the voltage
on the quality factor control line 37.

[0060] In the embodiment shown in FIG. 4, the switching circuit 9
comprises both the N-channel FET 311 and the further N-channel FET 401.
Again, the open and closed states of the switching circuit 9 correspond
respectively to a state in which the N-channel FET 311 is off and a state
in which the N-channel FET 311 is on. The series resistance of the
switching circuit 9 in its closed state is thus varied by varying the
intrinsic on-resistance in the further N-channel FET 401.

[0061] The embodiment shown in FIG. 4 has the advantage that the load on
the digital-to-analog converter 83 only changes when a change in the
quality factor Q is invoked, whereas in the embodiment of FIG. 3, the
load on the digital-to-analog converter 83 changes every time the
transmit signal changes. This may allow a lower power consumption in the
bandwidth controller 8. Furthermore, instead of using a separate further
N-channel FET 401 for each output stage 31, 32, several output stages 31,
32 may be connected to a common further N-channel FET 401.

[0062] In an alternative embodiment shown in FIG. 5a, the further
N-channel FET 401 is replaced by an electronic switch element 501 in
parallel with a resistor 502, and a control input of the electronic
switch element 501 is connected to allow opening and closing of the
electronic switch element 501 by means of the quality factor control
signal on the quality factor control line 37. When the electronic switch
element 501 is closed, the resistance in series with the N-channel FET
311 is at a minimum. When the electronic switch element 501 is open, the
resistance in series with the N-channel FET 311 equals the resistance of
the resistor 502. In this embodiment, the switching circuit 9 comprises
the N-channel FET 311, the electronic switch element 501 and the resistor
502, and it allows varying the quality factor Q between two predetermined
values. The electronic switch element 501 may e.g. be an FET controlled
by a gate voltage.

[0063] In a further alternative embodiment shown in FIG. 5b, multiple
resistors 502 are connected in parallel with the electronic switch
element 501, each in series with a further electronic switch element 503
such that the resistance provided in series with the N-channel FET 311
when the electronic switch element 501 is open, may be varied by opening
and closing different combinations of the further electronic switch
elements 503. The quality factor control signal is preferably implemented
as a multi-bit digital signal in order to allow individual control of
each of the electronic switch elements 501, 503. In this embodiment, the
switching circuit 9 comprises the N-channel FET 311, the electronic
switch elements 501, 503 and the resistors 502, and the quality factor Q
may be varied between a number of predetermined values depending on the
number and individual resistances of the resistors 502. The multiple
resistors 502 may be identical or they may have different resistances in
order to widen the range of obtainable series resistances. Each of the
further electronic switch elements 503 may e.g. be an FET controlled by a
gate voltage.

[0064] In the embodiments shown in FIGS. 3, 4 and 5, the transmit signal
must be set to a low level for the resistive load to affect the quality
factor Q. This is obviously achievable in the receive mode, where the
output levels of the output stages do not otherwise affect the received
signal. In the transmit mode, however, it is generally desirable that the
resistive load be maintained consistently and thus for both positive and
negative values of the transmit signal. In order to achieve this,
switching circuits (not shown) similar to the switching circuits 9 around
the N-channel FET may be implemented around the P-channel FET 312. The
series resistance of such switching circuits around the P-channel FET 312
may be controlled in manner similar to how the series resistance of the
switching circuits 9 around the N-channel FET is controlled. This does,
however, require the bandwidth controller 8 to output a further voltage
signal for this purpose. As an alternative, two output stages 31, 32 or
two groups of output stages 31, 32 may be operated to function as
full-bridge push-pull output stages, such that at any time, at least one
group has a low-level transmit signal on its input. If identical
switching circuits 9 are implemented in the output stages 31, 32 of both
groups and all of these are connected to the quality factor control line
37, e.g. as shown in FIG. 3, 4 or 5, then the resistive loads of these
groups will alternately control the quality factor Q when the transmit
signal alternates between high and low levels. Thus, the quality factor
is maintained consistently in both the positive and negative phases of
the transmit signal.

[0065] The switching circuits 9 of FIG. 3, 4 or 5 may alternatively or
additionally be implemented in the variable capacitance portion of the
capacitive energy storage 2, e.g. in the storage circuits 20. For this
purpose, similar switching circuits 9 may be applied to each or any of
the tuning switches 24, 25. The series resistance of these switching
circuits 9 may be controlled by the same quality factor control signal or
by a further quality factor control signal provided by the bandwidth
controller 8. A disadvantage of such a configuration is, however, that at
least some of the tuning switches 24, 25 must be closed to allow control
of the quality factor Q, which may impose undesired constrains on the
range of achievable combinations of resonance frequency f0 and
quality factor Q. On the positive side counts that the combined
capacitance of the series capacitors 21, 23 in the variable capacitance
portion of the capacitive energy storage 2 may be substantially larger
than the combined capacitance of the output capacitors 33, 34, e.g. five
times or ten times larger, so that altering the series resistance of the
switching circuits 9 will have a larger effect when the switching
circuits 9 resides in the variable capacitance portion of the capacitive
energy storage 2.

[0066] In order for the series resistance of a switching circuit 9 to have
a substantial effect on the quality factor Q, the switching circuit 9
should preferably be connected to an antenna terminal 10, 11 through a
major capacitance, such as one of the tuning capacitors 21, 23 or one of
the output capacitors 33, 34. The input capacitors 52, 53 are less suited
for this purpose due to their relative small capacitances.

[0067] The transmitter 1 is implemented in an integrated circuit (not
shown). This not only allows reducing the circuit size and production
costs, but also facilitates series production of transmitters 1 with
reproducible properties.

[0068] FIG. 6 shows example signals for the transmit mode and the
calibration mode along a time axis t. An information signal D comprises a
stream of binary data or symbols (0/1) with a sample interval Ts of
e.g. 5 μs. An electromagnetic signal TX is transmitted at an
instantaneous frequency f, which alternates between two transmit
frequencies f1, f2 depending on the information signal D being
respectively 0 or 1. The carrier frequency fc, which is the mean of
the transmit frequencies f1, f2, is 10 times the data rate (200
kHz) of the information signal D, e.g. 2 MHz. The modulation depth is
1/40, so that f1 and f2, respectively, equal 39/40 fc and
41/40 fc, e.g. 1.95 MHz and 2.05 MHz. Consequently, 9.75 signal
periods are transmitted during a 0-sample interval Ts and 10.25
signal periods during a 1-sample interval Ts so that the phase P of
the transmitted electromagnetic signal TX shifts by plus or minus π/2
for each sample interval Ts. This modulation is commonly known as
minimum-shift keying (MSK), which is a type of continuous-phase
frequency-shift keying (CP-FSK). The relative 3 dB-bandwidth of the
transmitted electromagnetic signal TX is about 1.2 times the relative
data rate, i.e. about 1.2×200 kHz/2 MHz=about 12%.

[0069] The energy storage 2 and the antenna 100 form a resonance circuit
with a resonance frequency f0 given by

f0=1/(2π {square root over (LC)}), (1)

a quality factor Q given by

Q=(1/RT+1/R) {square root over (L/C)} (2)

and a 3-dB bandwidth BW given by

BW=f0/Q (3)

wherein L, C and R are defined as described further above, and RT is
a resistance in series with one of the antenna output terminals 10, 11
which models losses in the transmitter 1. The equations are
approximations that are valid for higher values of the quality factor Q.

[0070] The antenna 100 and the transmitter 1 should be dimensioned such
that the range of resonance frequencies f0 obtainable by changing
the effective storage capacitance C by means of the tuning switches 24,
25 comprises the entire frequency band of the electromagnetic signal TX
to be transmitted by the transmitter 1. In the shown transmitter 1, the
range of obtainable resonance frequencies f0 comprises several such
frequency bands, which allows tuning of the transmitter 1 to any one of a
plurality of frequency channels. Alternatively, the range of obtainable
resonance frequencies f0 may be smaller than the range of transmit
frequencies f1, f2, provided that the obtainable pass-bands
comprise the transmit frequencies f1, f2.

[0071] The resonance circuit 2, 100 has a maximum quality factor Q of 50
and thus has a minimum relative 3-dB bandwidth of 2%. If the resonance
frequency f0 was kept at a fixed value, at least one of the transmit
frequencies f1, f2 would thus be outside the minimum 3-dB
pass-band of the resonance circuit 2, 100. To avoid this, the transmitter
changes the resonance frequency f0 whenever the information signal D
changes. The change is effected such that the instantaneous frequency f
remains within the pass-band of the resonance circuit 2, 100. To achieve
this, the resonance frequency f0 is shifted in the same direction as
the shifts in the instantaneous frequency f. Preferably, the change is
effected such that the resonance frequency f0 substantially always
equals the instantaneous frequency f during transmission. The change in
the resonance frequency f0 is effected by changing the effective
storage capacitance C. It should be noted that instead of 3 dB, any other
suitable attenuation level in the resonance circuit 2, 100, e.g. 10 dB,
could be used as criteria for determining whether the instantaneous
frequency f is within the pass-band or not.

[0072] The tuning control signal, which must be provided to the tuning
input terminal 15 by an external unit, e.g. a signal processor 704,
indicates the frequency channel(s) to be used for transmission and
reception as well as the modulation depth and the modulation type to be
used.

[0073] In the receive mode, the control unit 7 computes the receive
frequency, the receive bandwidth and the required quality factor from the
receive channel indicated in the tuning control signal, indicates the
receive frequency, the modulation depth and the modulation type in the
receive control signal to the demodulator 6, indicates the required
quality factor in the transmit control signal to the bandwidth controller
8 and sets the resonance frequency f0 to equal the receive
frequency.

[0074] In the transmit mode and the calibration mode, the control unit 7
computes the transmit frequencies f1, f2, the transmit
bandwidth and the required quality factor from the transmit channel and
the modulation depth indicated in the tuning control signal and indicates
the desired transmit frequency f1, f2 and the required quality
factor in the transmit control signal to the modulator 4 and the
bandwidth controller 8. The desired transmit frequency is either f1
or f2, depending on the state of the information signal D. The
control unit 7 further sets the resonance frequency f0 to always
correspond with the desired transmit frequency f1, f2. The
control unit 7 controls the effective storage capacitance C, and thereby
also the resonance frequency f0, through the tuning control signals
provided to the tuning control lines 28, 29.

[0075] In the shown transmitter 1, the control unit 7 preferably commands
the bandwidth controller 8 to switch between a relatively small bandwidth
in the transmit mode and a relatively large bandwidth in the receive
mode, so that the effective transmit and receive signal bandwidths may be
equal. The bandwidth controller 8 may alternatively be used in a
transmitter without dynamic control of the resonance frequency, in which
case, the bandwidth should be controlled to match the currently desired
transmit or receive bandwidth.

[0076] Alternatively, the control unit 7 may command the bandwidth
controller 8 to temporarily set a lower quality factor Q when switching
from transmit mode to receive mode in order to temporarily achieve a
stronger attenuation of the electric signal in the resonance circuit 2,
100 and thus a faster decay of the transmitted electromagnetic signal TX
after transmission. This allows a faster switch from transmit mode to
receive mode in half-duplex operation. A configuration with switching
circuits 9 in the variable capacitance portion of the capacitive energy
storage 2 may be advantageous for this purpose, since it may allow
achieving a lower quality factor Q due to larger series capacitances 21,
23, and since all tuning switches 24, 25 may be closed during the decay
of the transmitted electromagnetic signal TX, because in this situation,
it is likely not important to maintain a specific resonance frequency
f0 of the resonance circuit 2, 100.

[0077] In an alternative embodiment (not shown), the demodulator 6
comprises a frequency-determining circuit for determining the
instantaneous frequency of the signal received and amplified by the
receiver circuit 5, and the control unit 7 sets the effective storage
capacitance C in dependence on the determined instantaneous frequency.
The instantaneous frequency may alternatively be determined from any
other signal provided to or by the power amplifier 3, such as a modulator
output or the electric transmission signal. In such alternative
embodiments, the modulator 4 may be controlled by, or be part of, an
external unit.

[0078] When a change of the effective storage capacitance C is required to
change the resonance frequency f0, the control unit 7 opens and/or
closes the tuning switches 24, 25 in a subset of the storage circuits 20.
This causes a reconfiguration of the respective storage circuits 20 so
that they alter their effective capacitance C. In order to avoid losing
energy in the energy storage 2 and/or producing glitches during
switching, the control unit 7 controls the timing of switch closings such
that they take place when the respective tuning switch 24, 25 has a
minimum voltage across its terminals. In the disclosed embodiment, the
power supply voltages are asymmetric with respective to signal ground,
e.g. 1.25 V and 0 V respectively, and the tuning switches 24 and 25 of
each storage circuit 20 must thus be closed at different points in time
to achieve this. In embodiments with symmetric power supply voltages, the
control unit 7 may instead close the tuning switches 24 and 25
simultaneously. In the disclosed embodiment, the effective capacitance
Ci of each storage circuit 20 increases when the control unit 7
closes the tuning switches 24, 25 and decreases when it opens the tuning
switches 24, 25. The intermediate capacitor 22 ensures that the voltage
across an open tuning switch 24, 25 does not reach the high levels
occurring on the antenna output terminals 10, 11.

[0079] In the disclosed embodiment, the effective reactance between the
antenna output terminals 10, 11 substantially equals the effective
storage capacitance C and is thus substantially capacitive. Also, the
individual reactance elements, i.e. the tuning capacitors 21, 22, 23,
comprised in the energy storage 2 are substantially capacitive. The
energy storage 2 may, however, additionally or alternatively comprise one
or more inductive reactance elements, such as inductors.

[0080] The modulator 4 uses the clock signal to derive a stable time
reference for the transmit signals provided to the digital output stages
31, 32. The transmit signals are provided as square-wave signals. The
modulator 4 provides the transmit signals so that the instantaneous
fundamental frequency f of the electric transmission signal provided by
the power amplifier 3 to the antenna output terminals 10, 11 equals the
desired transmit frequency indicated in the transmit control signal. The
output of each digital output stage 31, 32 alternates between a power
supply voltage and signal ground in dependence on the transmit signal on
its input. These square-wave output signals are led to the antenna output
terminals 10, 11 through the output capacitors 33, 34, which cooperate
with the other capacitances in the transmitter 1 to form impedance
transformers. This allows the signal voltage at the antenna output
terminal 10, 11 to exceed the signal voltages on the outputs of the
output stages 31, 32.

[0081] The modulator 4 provides the transmit signals such that within each
dual amplifier circuit 30, the digital output stage 31, 32 receive
transmit signals that are inverted with respect to each other. Thus, each
dual amplifier circuit 30 provides an either positive or negative
contribution to the differential voltage across the antenna output
terminals 10, 11. The availability of seven identical dual amplifier
circuits 30 allows controlling the output of the power amplifier 3 to any
one of a number of different levels by controlling the timing of
transitions in the transmit signals, or alternatively or additionally
idling one or more of the transmit signals. The output of the power
amplifier 3 may further be shaped to reduce the amount of radiated
harmonics. To achieve this, the modulator 4 may control the transitions
in the transmit signals on the individual signal lines 35, 36 such that
the electric transmission signal to the antenna 100 is as close to a pure
sine wave signal as possible during each sample interval Ts. The
resonance circuit 2, 100 further functions as a steep band-pass filter,
which suppresses a major portion of the harmonics.

[0082] The demodulator 6 demodulates the differential receiver output
signal according to the receive frequency, the modulation depth and the
modulation type indicated in the receive control signal from the control
unit 7. Preferably, the received signal is modulated with the same type
of modulation as is applied to the transmitted electromagnetic signal TX,
and the data rates used for transmission and reception of signals are
equal. In this case, the receiver 5 and/or the demodulator 6 may comprise
means for improving reception of electromagnetic signals residing outside
the 3-dB pass-band of the resonance circuit 2, 100. Such means may
comprise e.g. a filter adapted to at least partly compensate for the
amplitude and phase changes caused by the resonance circuit 2, 100. Such
means are e.g. described in the co-pending patent application EP 2 367
294. Alternatively, the transmitter 1 may be used for receiving signals
with a data rate smaller than the transmitted data rate. In this case,
means for improving out-of-band reception may be omitted. Alternatively
or additionally, the bandwidth controller 8 may be used to set different
bandwidths of the resonance circuit 2, 100 for the transmit and receive
modes. The demodulator 6 may derive a time base for demodulation from the
received signal or, alternatively, from the clock signal. In the latter
case, the clock signal should also be routed from the clock terminal 12
to the demodulator 6.

[0083] The receiver circuit 5 allows the transmitter 1 to operate as a
half-duplex transceiver by switching between the receive mode and the
transmit and/or the calibration mode. In the calibration mode, the
receiver circuit 5 further allows an external unit, e.g. a signal
processor 704, to monitor the amplitude and/or phase of the electric
transmission signal on the antenna output terminals 10, 11 and thus to
achieve a calibrated transmitter output as well as to tune the resonance
frequency of the resonance circuit 2, 100 by methods already known in the
art. It also allows achieving a calibrated quality factor Q in the
resonance circuit, e.g. by performing measurements comprising frequency
sweeps and/or at discrete frequencies. The transmitter 1 may
alternatively comprise own means for monitoring the amplitude and/or
phase of the electric transmission signal, such as a circuit in the
demodulator 6. The control unit 7 may receive an output of such means and
automatically change the effective storage capacitance C to achieve
maximum amplitude of the electric transmission signal during
transmission. Similarly, the transmitter 1 may comprise own means for
monitoring the amplitude and/or phase of the current through the output
transistors 311, 312 of the output stages 31, 32 for the purpose of
determining the resonance frequency and/or the quality factor Q of the
resonance circuit 2, 100. Automatic adjustment or calibration of the
resonance circuit 2, 100 and/or the bandwidth controller 8 may take place
continuously during transmission, at specific time intervals, e.g. once
per minute, hourly, daily or weekly, or at specific events, such as upon
start-up of the transmitter 1. Instead of a full-power transmitted
signal, transmission of a weaker test signal may be used for calibration.
Calibration results for the quality factor Q may be written to the
calibration memory 82 of the bandwidth controller 8 in known fashion.

[0084] FIG. 7 shows a hearing device 700, e.g. a hearing aid or an active
ear-protection device, comprising a transmitter 1 and a loop antenna 100
configured as described above. The hearing device 700 further comprises a
microphone 701, a preamplifier 702, a digitiser 703, a signal processor
704, a pulse-width modulator 705 and a speaker 706 connected to form an
audio signal path 707. The hearing device 700 further comprises a battery
708 for powering the transmitter 1 and the devices 702, 703, 704, 705 in
the audio signal path 707. The microphone 701 is arranged to receive an
acoustic input signal from an individual's surroundings and provide a
corresponding microphone signal to the preamplifier 702. The preamplifier
702 is adapted to amplify the microphone signal and provide the amplified
microphone signal to the digitiser 703. The digitiser 703 is adapted to
digitise the amplified microphone signal and provide a digitised audio
signal to the signal processor 704, which is adapted to modify the
digitised audio signal in accordance with the purpose of the hearing
device 700, i.e. to improve or augment the hearing capability of the
individual. The signal processor 704 is adapted to provide the modified
audio signal to the pulse-width modulator 705, which is adapted to
provide a corresponding pulse-width modulated signal to the speaker 706.
The hearing device 700 is adapted to be arranged at or in an ear of the
individual, and the speaker 706 is arranged to transmit an acoustic
output signal corresponding to the pulse-width modulated signal to the
individual.

[0085] The signal processor 704 is connected to receive the receive data
signal from the receive data terminal 16 of transmitter 1 and the
calibration data signal from the calibration data terminal 17 of
transmitter 1. The signal processor 704 is adapted to adjust its
modification of the digitised audio signal in response to information
comprised in the receive data signal and/or to provide the modified audio
signal in dependence on an audio signal comprised in the receive data
signal. This allows the hearing device 700 to change its audio signal
processing in response to e.g. commands, status information and/or audio
signals received wirelessly in an electromagnetic signal from a remote
device (not shown), and/or to include such audio signals in the acoustic
signal transmitted by the speaker 706. The remote device could e.g. be a
remote control, a second hearing device located at or in the respective
other ear of the individual or an auxiliary device, e.g. a so-called
audio gateway device, adapted to transmit an audio signal from an
external device, such as e.g. a mobile phone or a TV set, to the hearing
device 700.

[0086] The signal processor 704 further is connected to the clock terminal
12, the mode input terminal 14 and the tuning input terminal 15 of the
transmitter 1 and further is adapted to provide the corresponding signals
for controlling the transmitter 1 as described further above. The signal
processor 704 further is connected to the data input terminal 13 of the
transmitter 1 and further is adapted to provide the information signal D
comprising information such as commands, status information, control
signals and/or audio signals to the transmitter 1 for transmission to a
remote device, which could e.g. be another hearing device for the
opposite ear or an auxiliary device.

[0087] The signal processor 704 further is adapted to calibrate the
bandwidth controller 8. It initially provides appropriate signals to the
transmitter 1 in order to set a desired channel frequency and bandwidth
as well as to invoke the calibration mode. It then has the transmitter 1
perform a frequency sweep of the transmitted electromagnetic signal TX,
simultaneously reads the signal amplitude and phase returned in the
calibration data from the demodulator 6 and computes the achieved quality
factor Q from the read signal amplitudes and phases. If the computed
quality factor Q does not correspond with the desired channel frequency
and bandwidth, it commands the bandwidth controller to change the content
of the calibration memory 82 in order to improve the correspondence. The
tuning control signal and the transmit control signal may comprise
dedicated calibration commands for this purpose. Instead of a frequency
sweep, measurements at discrete frequencies may be made. Furthermore,
instead of reading calibration data from the demodulator 6, currents
through one or more of the output transistors 311, 312 may be measured by
appropriate means (not shown) and used for computing the actual quality
factor Q. The calibration function may alternatively be implemented in
the control unit 7 and/or in the bandwidth controller 8.

[0088] The audio signal path 707 is preferably implemented mainly as
digital circuits operating in the discrete time domain, but any or all
parts hereof may alternatively be implemented as analog circuits
operating in the continuous time domain. Digital functional blocks of the
audio signal path 707 and/or of the transmitter 1 may be implemented in
any suitable combination of hardware, firmware and software and/or in any
suitable combination of hardware units. Furthermore, any single hardware
unit may execute the operations of several functional blocks in parallel
or in interleaved sequence and/or in any suitable combination thereof.

[0089] The hearing device 700 may be part of a binaural hearing system.

[0090] The transmitter 1 may be used in any type of device, most
advantageously in battery-driven and/or portable devices.

[0091] Further modifications obvious to the skilled person may be made to
the disclosed method, system and/or device without deviating from the
spirit and scope of the invention. Within this description, any such
modifications are mentioned in a non-limiting way. The possible
modifications below are mentioned as examples hereof.

[0092] The energy storage 2 and/or the storage circuits 20 may configured
in many alternative ways, while still allowing dynamic change of the
effective storage capacitance C. More specifically, the number of storage
circuits 20 may vary; this applies to the total number of storage
circuits 20, the number of storage circuits 20 in the thermometer-coded
array 200 as well as the number of storage circuits 20 in the
binary-coded array.

[0093] The power amplifier 3 may alternatively comprise analog output
stages. The amplifier circuits 30 may comprise only a single-output
output stage. The number of amplifier circuits 30 may vary, and the
individual amplifier circuits 30 may differ from each other, e.g. in
maximum power.

[0094] The control unit 7, the bandwidth controller 8 as well as the
content and routing of control signals associated herewith may be
implemented in many alternative ways. For instance, the transmit control
signal and the receive control signal may be provided by external units.

[0095] Alternatively to MSK, any form of frequency or phase modulation may
be used for modulating the information signal onto the carrier signal.
Such modulation forms include e.g. analog frequency modulation, analog
phase modulation, phase-shift keying, frequency-shift keying and
combinations hereof. These modulation forms allow both analog and digital
information signals to be modulated onto the carrier signal. Especially
preferred modulation techniques comprise minimum shift keying and other
of the various known modulation techniques that reduce the bandwidth of
the transmitted electromagnetic signal TX and thus allow the modulated
signal to pass the resonance circuit 2, 100 with minimum attenuation
and/or require less dynamic change of the a resonance frequency f0.
The power amplifier 3, the modulator 4, the receiver circuit 5, the
demodulator 6 and the control unit 7 may obviously need to be adapted to
such altered modulation forms.

[0097] The transmitter 1 may be implemented such that it allows the
resonance circuit 2, 100 to have a relative 3 dB-bandwidth less than 10%
(Q>10), less than 5% (Q>20), less than 3% (Q>˜33) or less
than 2% (Q>50), depending on e.g. the carrier frequency fc, the
bandwidth of the information signal D, the desired communication range,
the acceptable bit error rate as well as other requirements for the
communication link in which the transmitter 1 is to be used.

[0098] The transmitter 1 may be implemented such that it allows the
resonance circuit 2, 100 to be tuned to a resonance frequency f0
below 100 MHz, below 30 MHz or below 10 MHz, depending primarily on the
desired range of carrier frequencies fc. A lowering of the frequency
generally causes lower power consumption. The resonance circuit 2, 100
should, however, preferably be tuned to form a narrow band-pass filter
with a resonance frequency f0 above 300 kHz, above 1 MHz or above 3
MHz in order to allow a high data rate (bits per second) in the
communication link. Preferred data rates are above 40 kb/s, above 80 kb/s
or above 160 kb/s in order to allow transmission and reception of
real-time audio signals through the transmitter 1.

[0099] The antenna 100 and/or the transmitter 1 may be dimensioned to have
a largest physical extension of less than 5 cm, less than 2 cm or less
than 1 cm. Small dimensions allow implementation in small devices. The
transmitter 1 may further be dimensioned to having a power consumption
less than 10 mW, less than 3 mW or less than 1 mW. This allows use of the
transmitter 1 in battery-powered devices, such as body-worn hearing
devices 700.

[0100] Stable transmit frequencies f1, f2 may be derived from
e.g. a system clock signal within the device 700 in which the transmitter
1 is comprised. Alternatively, the transmit frequencies f1, f2
may be derived from a dedicated oscillator, e.g. a crystal oscillator or
any other oscillator.

[0101] Some preferred embodiments have been described in the foregoing,
but it should be stressed that the invention is not limited to these, but
may be embodied in other ways within the subject-matter defined in the
following claims. For example, the features of the described embodiments
may be combined arbitrarily, e.g. in order to adapt the system, the
devices and/or the method according to the invention to specific
requirements.

[0102] It is further intended that the structural features of the system
and/or devices described above, in the detailed description of `mode(s)
for carrying out the invention` and in the claims can be combined with
the methods, when appropriately substituted by a corresponding process.
Embodiments of the methods have the same advantages as the corresponding
systems and/or devices.

[0103] Any reference numerals and names in the claims are intended to be
non-limiting for their scope.

Patent applications in class Power control, power supply, or bias voltage supply

Patent applications in all subclasses Power control, power supply, or bias voltage supply