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AUDIO and RF MODULATOR

TRIODE, TETRODE, PENTODE AND BEAM POWER TUBES

Readers, please
note this page is presented for your education, information and guidance only -
it is not intended to be a technical/scientific treatise. The concepts and
ideas presented herein are just as much subjective as objective.

It attempts to
assist those experienced audiophile home constructors who wish to
explore further tube amplifier development options and are prepared "to go
beyond the square" - to challenge the paradigm of "the
status-quo".

This paper refers
only to the characteristics and performance of push-pull tube audio amplifiers
without negative feedback.

For reasons
detailed elsewhere in my website I have no interest whatsoever in single-ended
amplifiers - however the concepts described herein are as equally applicable to
single-ended amplifiers as to push-pull configurations.

For reasons
detailed elsewhere in my website I have no interest whatsoever in the use of
trans-stage negative feedback thus the concepts described herein are intended
to optimise electron tube performance without the use of negative feedback.

For full ratings
and applications of specific tube types in which you are interested please
refer to the manufacturer's catalogue.

Copyright in all
quoted works remains with their owner, author and publisher, as applicable.

Please note that
no warranty is expressed or implied - see footnote notice.

Excepting
where rights in Intellectual Property previously exist, all rights in the
applied engineering concepts expressed in this paper remains exclusively with
the author.

The whole or
part thereof of this paper and/or the designs and design concepts expressed
therein may be reproduced for personal use only without limitation - but must not under any circumstances be
applied or used for commercial gain or reward without the express written
permission of the author, being the intellectual copyright owner.

Historically we
have been restricted by convention to a small set of engineering design
principles that have imposed a barrier to further development, reinforced by
commercial manufacturers staying with the tried and true "safe"
design configurations.

However the
availability of low-cost high-performance solid state amplifiers and
pre-amplifiers, together with advances in recording formats - currently Super
Audio CD and DVD standards - have provided stimulus to tube hi-fi enthusiasts
to improve the performance of their existing equipment.

Many of the old
constraints are longer with us - hiss, noise, wow, flutter and rumble etc. do
not present on CD or DVD - so enable us to open the frequency range and dynamic
range window a little more.

Modern CD's and
DVD's offer a substantially higher dynamic range than traditional analogue
vynil recordings. This dynamic range is compressed when the amplifier is not
capable of reproducing it, resulting in a loss of fidelity and realism.

In the case of
ultra-linear connected TETRODES and BEAM POWER TUBES only the
Screen Grid is connected to the 40-50% turns transformer primary winding tap as
shown above (because there is no separate Suppressor Grid terminal on the tube)

RATIO
of Screen-grid and Suppressor Grid and Cathode Bypass to Plate Bypass
Capacitors

Single
or double stage PI filter in each power supply wherein the filter capacitors
are of equal capacitance value and each inductor filter choke is bypassed
by a reverse biased diode

Decoupling
of the Power Amplifier from the Power Supply by means of a series
connected silicon diode

Decoupling
each amplifier stage from the next following by means of a series
connected silicon diode

Symmetrical
balanced AC signal drive system whereby the central axis of the AC signal
OUTPUT from the Driver stage circuit and the AC signal INPUT of the Driven
stage circuit are at the same common reference voltage potential.

.To optimise the Screen-Grid voltage, first
carefully and accurately measure the physical gap spacing between the Cathode,
Screen-Grid and Plate.

In most cases
this can be easily done without having to destroy a valuable tube. Just measure
the spacing between Cathode andAnode and of the
grid support pins at the top of the tube. In the case of beam power tubes this
may be difficult because there isusually a box-like
assembly covering the Plate structure. But a bit of ingenuity should solve the
problem

If it becomes
necessary to destroy a tube to examine its internals safety precautions must be
observed:

1.
Wrap the tube in a strong cloth2.
Place the tube in a vyce and squeeze the glass bottle slowly until it implodesor3.
Gently hit the glass with a hammer at a point not directly over the electrode
assembly4.
Carefully remove the broken glass and dispose of safely5.
Carefully cut away the electrodes until the elements can be measured

The
"gap" spacing ratio can then be calculated for the particular tube
type.

To determine the
Optimised Screen-Grid voltage, multiply the Plate voltage by the gap ratio.

It is obvious the
Screen-Grid voltage will be substantially less than the Plate voltage -
typically in the range 30 to 50 % but usually 40% - so a separate stable supply
is essential.

The designer has
the option of using:

a
voltage divider network from the Plate Supply (not recommended for hi-fi)

a
stabilised Screen-Grid DC supply derived from the same AC source (power
transformer) as the Plate supply (not recommended for hi-fi)

a
separate Screen-Grid DC supply derived from a separate power transformer
and full-wave rectifier to the Plate supply (tube rectifier not
recommended for hi-fi)

a
separate regulated Screen-Grid DC supply derived from a separate power
transformer and full-wave rectifier to the Plate supply (tube rectifier
not recommended for hi-fi)

What is most important is for the Screen-Grid DC
supply voltage to be reasonably constant between no-signal load and full-signal
load conditions (ie well regulated).

This can be
effectively achieved by using a separate power transformer of generous rating -
ie at least twice the maximum signal DC Screen-Grid current, a full-wave
silicon bridge rectifier and humungous filter capacitors in capacitor input to
filter configuration - preferably with a low DC resistance filter choke
installed too.

For the more
theoretical designer some thought could be given to the "Virtual
Cathode" concept, which suggests that the portion of electron stream
devoted to the negative bias applied to Grid #1 creates a more negative
"Virtual Cathode" in the region of the physical Grid element. This
concept however is difficult to assess because so long as the tube is
conducting some of the electron stream is continuous between real Cathode and
Plate. Maybe only the outer portions of the electron stream are affected by the
negative bias and not the entire thickness of stream.

The perfectionist
- or those having the necessary equipment - looking for the ultimate
optimisation could determine the optimum operating voltage for the Screen-Grids
by means of a Signal Generator, Distortion Meter and variable voltage power
supply to the Screen-Grids.

The optimised AC
load impedance for the Screen-grid will be the ratio of DC Screen-grid volts to
Plate volts x Plate load impedance.If the Screen-grid
load is provided by a fixed resistor the ratio for a Screen-grid - or set of
Screen-grids in one side of a push-pull set, will be the DC voltage ratio x one
quarter Plate to Plate load impedance

Note however that
if the load is provided by a transformer as in ultra-linear or equivalent mode,
the Screen to Screen load impedance will be directly proportional, but the
turns ratio will be the square of the DC voltage ratio.

eg If the DC
ratio is 40% then the turns ratio for the Screen-grid tap on the output
transformer will be 0.4 x 0.4 = 0.16 or 16%

I am indebted to Rudolf
Moers, a distinguished Electrical and Electronics Engineer located in the
Netherlands, who has made available for us his wonderful recent scientific
investigation into the design theory and practice of Ultra-linear audio
amplification.

These papers are posted with
permission from Linear Audio www.linearaudio.net and their author Rudolf
Moers.

Manufacturers'
tube data sheets generally describe Pentodes as having higher measured
distortion than Beam Power Tubes so most audio amplifier designs have focused
on the latter types.

In the case of
recycled or salvaged tubes from yesteryear, it is evident that by WWII, RF
design engineers preferred Beam Power Tubes because of their capability to
operate at higher RF frequencies, hence the true Pentode enjoyed only a short
claim to fame.

One exception to
this is the mighty 803 Pentode, which found widespread use in submarine
and destroyer RF applications because of its ruggedness, stability, reliability
and long-life. Of interest is that the 803 pentode has aligned grids, each
having a ceramic coating to limit electron attraction, thereby improving predictability
and stability of operation, as well as efficiency.

The 837
Pentode tube is also renown as a very reliable oscillator tube for RF
transmitters. The 837 Pentode has a Plate structure similar to that of the
famous TT21 and KT88 Beam Power Tubes.

However Pentodes
do offer user benefits over Tetrodes and have a place in high-fidelity audio
and RF amplifiers - particularly for home constructors who might have a box of
useful Pentodes just waiting to be used in a suitable design.

It is a well established tube
engineering principle that the current in a tube can be regulated using ANY of
the grid elements.

For example, in domestic
radio receivers, the use of multi-grid tubes such as 6A7, 6A8 and 6L7, each
having 5 grids, is a standard application of radio engineering design.

However for audio engineering
purposes, three grids appears to be the practical limit - beyond which no
appreciable benefit is to be realised. In fact. the beam Power Tube utilises
only two grids to control current through the tube.

IMPORTANT NOTE 1: It is the case in many of the popular
PENTODES that the SUPPRESSOR-GRID is internally connected during
manufacture to the CATHODE, so it is not possible to externally access
the Suppresser Grid. In this class of tube the manufacturer controls the
behaviour of the Suppressor-grid and the user cannot do anything to change
that.

IMPORTANT NOTE 2: Tube identification protocols
describe and illustrate by base pinout diagrams Beam Power Tubes as
"Pentodes". These tubes are essentially Tetrodes that have a
beam forming electrode used to both confine the electron stream to a
pre-determined width within the tube, and to focus the electron flow onto a
particular area of the Plate. They usually have aligned Grids to arrange the
electron flow in multiple sheets.

Therefore it is essential to
physically examine tubes of interest to determine their actual mechanical
construction.

DO NOT RELY ON
DESCRIPTIONS OR BASE DIAGRAMS PUBLISHED IN TUBE MANUFACTURERS' CATALOGUES,
MANUALS OR DATA SHEETS.

However those
tubes having a separate base pin connection for the SUPPRESSOR-GRID
offer the designer the option of applying a suitable DC voltage consistent with
the proportional voltage divider effect of the internal electrode gap.

The rules and
requirements for an adequately regulated DC power source are the same as
described above for the Screen-Grid.

The SUPPRESSOR GRID (GRID
#3) regulates the flow of electrons in the tube in the same way as is the
case for Control-grid #1 and Screen-Grid #2.

In conventional
"Pentode" configuration, the Suppressor-grid is directly connected to
the Cathode either internally by the tube manufacturer or by the user.

Having left the Plate as
surplus, randomly travelling electrons, they find their way to the
Suppressor-grid, thence diverted to the Cathode to be absorbed back into the
electron stream. This arrangement obviously creates a short-circuit in respect
of those electrons attracted to the Suppressor-grid.

Thus there is an effective
internal or external circuit (as applicable) created between the
Suppressor-grid and the Cathode, that diverts some of the electrons back to the
Cathode. This current is lost to the output power and therefore reduces
efficiency in the output power stage.

In the case of the
relationship between the Screen-grid and Plate, most experts suggest that the
Plate sees the Screen-grid as the "Cathode", thus if this is so then
the DC potential between Plate and Suppressor Grid will be again determined by
the linear distance between them - unless the Suppressor-grid is purposefully
connected to the Cathode.

It also follows that because
the Suppressor-grid is now positive to the Screen-grid, Plate current will
increase - albeit slightly.

Since both Screen-grid and
Suppressor grids DC voltages will be fixed, it becomes obvious that to control
the electron flow within permissible limits, the negative bias voltage applied
to Grid #1 Control Grid will need to be made MORE NEGATIVE.

It is also obvious that to
limit the DC current flow in Grid #3, and to prevent an AC signal
short-circuit at the Suppressor-Grids, it is essential to load the
Suppressor-grids by installing a Grid-stopper resistor of around 75% of
equivalent Plate to Plate push-pull load impedance.

eg for a Plate to Plate load
of 8,000 Ohms, the transformer will present a load of 2,000 Ohms to each tube
in the push-pull pair.2,000 Ohms x 75% is 1500 Ohms.
This is still a relatively small value so should not present significant
voltage drop or regulation issues.

The optimised AC
load impedance for the Suppressor-grid will be the ratio of DC Suppressor-grid
volts to Plate volts x Plate load impedance.

If the
Suppressor-grid load is provided by a fixed resistor the ratio for a
Suppressor-grid - or set of Suppressor-grids in one side of a push-pull set,
will be the DC voltage ratio x one quarter Plate to Plate load impedance

I have not listed the
EL34/6CA7, its smaller brother the EL84/6BQ5 and cousin 6M5, because the
construction and effect of the Suppressor-grid in these tubes is
"nominal" and not in the same league as those of the transmitting
tubes listed above. Being manufactured from very fine wire, the Suppressor Grid
in these tubes is not capable of handling significant current or power. For high-powered audio amplifiers superior options are
available as shown above.

In this case, a
conventional pentode configured amplifier - where the pentode tube has a
separate externally connected independent Grid 3 pin (typically connected to
the cathode or ground) - may be modified by installing a silicon diode between
the Suppressor-grid pin on the tube socket and Cathode or ground - as
applicable. The arrow must point towards ground - ie the marked terminal on the
diode to the Suppressor-grid pin.

This
configuration allows AC current to flow to ground but blocks DC current into
the tube.

If any
instability or non-linearity occurs, a 1,000 Ohm resistor may be installed
shunting the diode, to create a permanent DC current path to the tube and
anchor the Suppressor Grid to the Cathode or ground.

In the example
cathode-bias is shown, however the same principles apply for a fixed-bias
design.

The notable
feature of this design is that everything is symmetrical and so is easy to
draw, read and understand.

The configuration
of each output tube is considered to be equally distributed about the central
axis of the output stage, (being the B+ and centre-tap of the output
transformer).

The core
electronic design concept assumes (deems) the Plate current in each tube will
be identical at all times and therefore any distortion in the plate circuits is
cancelled out via the push-pull effect. For an explanation of this refer to the
RCA Tube Manuals.

Screen-grids are
not considered to be more than a power output/efficiency enhancing benefit
derived from tetrodes v triodes, and as shown in the example above, are not
connected in any manner other than the most basic available - ie connected
together and supplied from the same power source as the Plates.

So in theory, all
of the components operate in a synchronous manner such that the output stage is
efficient and delivers its power at low distortion.

Unfortunately
things are not always as they appear.

The standard
tolerance on components is usually around + or - 10%, however some components,
including the tubes themselves, are assigned much wider tolerances.

Modern resistors
are typically + or - 5% however + or - 10% tolerance is still used.

Modern capacitors
display tolerances that vary with construction material and style.

Vacuum tubes
offer a tolerance of + or - 20% on transconductance when new. Depending upon
circuit parameters, performance usually varies downwards with use. If one power
tube draws grid current early on then performance of the pair will suffer.

Transformers vary
dramatically. Even if the primary centre-tap is exactly in the centre of
the total number of turns - ie the number of turns either side of the
centre-tap is exactly equal - the transformer may still not deliver an
equal power transformation between primary and secondary if the magnetic
properties of the two halves of the primary are not identical. Also, if the
secondaries are not equally distributed about the primary the induction into
each secondary may vary between halves. Factors such as core design and winding
design will influence the end result.

And what about
the symmetry of the driver stage? This too is subject to tolerance variables
and may deliver an asymmetrical signal drive voltage to the output tubes,
resulting in uneven power output from each half.

So what appears
to be a perfectly symmetrical design is in fact a widely varying practical
configuration.

One obvious
variant is Plate-current.

Many of the
popular tube types do not deliver a linear response between zero signal
conditions and maximum signal conditions. Whilst most guitar amp users would be
familiar with the concept of "re-biasing" after changing an output
tube, it is not the norm in hi-fi equipment. However despite good intent,
guitar amp (and hi-fi amp) users are blissfully unaware that their beautifully
balanced output stage at zero signal is nowhere near balanced at maximum
signal.

In lower-cost/lower
quality transformers it will be found by measurement that the DC resistance of
one half of the primary is somewhat different to that of the other half -
resulting in uneven Plate voltage as Plate-current increases.

Imbalance in a
push-pull output stage results in loss of power and increased distortion.

This condition is
more likely in high-gain tubes like the EL34/6CA7, EL36/6CM5, KT88 and 6550.

SCREEN-GRIDS

Ideally, the
output from the Screen-grids is also identical and therefore cancels out at the
output transformer centre-tap.

But note from the
above design that any voltage appearing at the centre-tap of the output
transformer - whether it be derived from a difference in Plate characteristics,
transformer characteristics, untransformed signal, reflected back emf from the
loudspeaker, or simply hum as a ripple voltage - will appear on one or the
other Screen-gid as either direct injection or feedback.

Obviously
Screen-grid current will normally be a portion of (ideally balanced) Cathode-current,
but if different brand tubes - or tubes of differing design from the same
manufacturer - are used in the output stage pair it is likely that the
Screen-grids will behave differently - resulting in not only different tube
characteristics across the range but also differences in the Screen-grid
performances. Imbalance becomes apparent as distortion.

It follows then
that a useful object of amplifier design would be to eliminate the
Screen-grid as a variable from the system.

This proposition
is very well explained by Renato D. Tancinco of the Philippines, in his 1961 US Patent 3153766

In this patent,
Tancinco presents his design, which aims to eliminate crossover distortion in
tetrodes and pentodes operating in Class B mode - EL34/6CA7 users take note.

Unfortunately
this design only works in Class B mode because it cancels out part or whole of
the opposite polarity alternating signal in the output stage, however it does
offer significant advantage as the patent itself explains.

As noted in my
Screen-Grids paper linked above, reference to tube handbooks shows that in a
typical beam power tube, the Screen Grid current at maximum signal power is
around 20% of Plate current. This ratio of currents appears to be largely
independent of Plate voltage. It would therefore be reasonable to assume that
up to 20% of prospective signal power is lost in the Screen Grid circuit in a
conventional amplifier. (Power = volts x amps. Power supply voltage to the
screen-grids can be up to 100% of plate voltage)

(Note: Two
notable exceptions are the 807 and 814 beam power tubes that incorporate
advanced design technologies to increase tube efficiency and reduce distortion,
however in the overall scheme of things this technology appears to have been
limited to these two tube types - if you are aware of others please let me
know)

Two conventional
options present to overcome this nominal ratio of 20% plate Current:

The first is
ultra-linear connection, where all the electrons collected by the Screen Grids
are fed into the output transformer, but in the process modify the output stage
characteristics.

The second is to
increase the value of the Screen Grid-stopper resistor to a value
sufficiently high to resist the flow of electrons to the Screen Grid.

However the
Grid-stopper resistor must be non-inductive to prevent oscillation. It must
also be capable of handling the Screen current passing through it. One way of
doing this is to use parallel carbon composition resistors (not film
type - to prevent fire) of sufficient number to obtain the required heat
dissipation rating to do the job without excessive temperature rise in the
resistors.

A further problem
here is regulation of the Screen Grid supply. Obviously a Grid-stopper resistor
of say 5,000 - 10,000 ohms will present a significant voltage drop when Screen
Grid current flows - if it does.

The loss of
regulation may be a price we have to pay to obtain a high standard of performance.

Another and
previously unpublished option to creating an operating environment where the
Screen Grids will be at a DC potential sufficiently high enough to attract and
accelerate electrons towards the Plates but - to maximise power output - not to
collect and divert them to earth through the B+ supply, is the humble silicon
diode semi-conductor rectifier.

By inserting a
standard half-wave silicon rectifier diode in series with the Grid Stopper
resistor, an electronic control circuit is created whereby the Screen Grid will
be able to be energised at DC potential attracting and accelerating electrons
towards the Plate - still electrostatically controlling current flow in the
normal way - but blocking the flow of AC current from the Screen Grid back to
the DC source - ie "one way traffic"

This works
because the current flow in the tube is always from the Cathode to the
Anode (Plate). The diode, being a semi-conductor, blocks current flow in
the reverse direction, thus enabling DC current to feed it in the conventional
manner but blocks AC current from passing back through it to a load.

Thus then
there is no output circuit formed between the Screen-grid and the load so no
current can flow in the usual direction.

This causes the
tube to appear at first glance to behave like a triode. Note however that the
tube is still a tetrode, pentode or beam power tube as applicable - it is not a
"super triode" as some folks suggest. Plate-current is still
controlled by Grid #1 and Grid #2 voltages in the usual way - the difference
being that the Screen Grid signal current is not shorted to the B+ supply (AC
ground) so is diverted to the Plate and collected as significantly additional
power output.

The diode is
connected between the B+ supply (line) and the Screen Grid (load) such that the
arrow points towards the Screen Grid. ie forward current is from the line to
the load.

This arrangement
offers huge benefits, because it prevents the Screen Grids from collecting
electrons - thereby diverting all the signal output to the Plates, increasing
tube efficiency, reducing distortion and increasing frequency response, as well
as eliminating the usual effects on changes in Screen Grid voltage on Plate
Current - therefore improving transient response.

One major
benefit is that the diode is not in the signal path and therefore does not
modify the sound.

A secondary
benefit is that there are no bypass or power supply capacitors (eg paper,
polyester or electrolytic) in the Screen-Grid signal path, which is a further
major improvement.

A further
benefit is that the non-linearity described under Fig. 2 above will be less of
a problem for us because the Screen-Grid component of the signal (ie those
electrons normally collected by the Screen-Grid) is diverted to the
Plate.

Thus by
inserting the humble silicon semi-conductor diode in the output circuit, we can
completely break the bonds of traditional audio practice and take a great leap
forward!!

This is not tube
heresy, because the diode is not in the signal path - it merely prevents the
signal from being affected by adverse circuit parameters such as
short-circuited Screen-Grids, fluctuations in Screen-Grid voltage and power
supply filter capacitors.

The relatively
low reverse resistance of the diode appears to adequately satisfy the need for
a low impedance return path to AC earth, so the Screen Grid is not actually
isolated from AC earth - but there is sufficient impedance in the circuit to
discourage electron flow through it.

Note: Having
regard to the EIMAC articles regarding secondary emission in tetrodes referenced
above, it should be the case that the use of a silicon diode in the Screen-Grid
supply will not impede reverse current flow - provided a suitable
bleeder resistor is used between the diode and B+ source - as would be the case
without the diode in the circuit.

Some audiophile
experimenters have used single or series strings of zener diodes in this kind
of circuitry to regulate the DC Screen Grid voltage derived either
directly from the Plate or from the B+ supply, however I have used single
conventional 1A 1000 PIV silicon rectifiers (from B+ supply only) with good
results.

Important
Note: When Zener Diodes are inserted in series with the Screen-Grid supply to
both drop and control Screen-Grid voltage, they are connected in reverse
polarity to the normal rectifier style diode described here, hence their effect
on Screen-Grid behaviour and of "sound" is quite different. A real
danger with the series Zener Diode configuration is that if the diode breaks
down and short-circuits then full supply voltage will be applied directly to
the Screen-Grid. This may destroy the tube in the process.

A further (and
very effective) enhancement is to use a separate diode for each Screen Grid (or
each set in parallel-push-pull) to ensure there is no cross-modulation in the
push-pull activity. This places back to back diodes between the Screen Grids,
which makes each half of the AC push-pull circuit independent to the other.
Less signal averaging and less cross-talk occurs between each half of the
push-pull pair, so the sound is cleaner.

In stereophonic
amplifiers using a common power supply, this system provides significantly
greater channel separation.

Of course,
silicon diodes can be retrofitted to an existing amplifier however the negative
feedback loop should be re-calibrated to suit the changed output circuit
conditions.

It may be
necessary to re-calibrate Grid #1 (Control Grid) bias to ensure Plate Current
and Plate Dissipation are optimised within the tube manufacturer's design
centre ratings.

In already set-up hi-fi
amplifiers, it should not be necessary to change operating conditions because
the Plate current is already determined by the Screen-grid voltage and not the
Plate voltage - but a prudent owner will check in any case to be sure.

If any reader can
shed further light on this breakthrough new concept please email your thoughts.

Important:
Please note this modification is not suited to Class B guitar amplifier
applications (this means most of the "big" guitar amps) because the sound
produced by the silicon diode to Screen-Grids configuration is cleaner and less
distorted, dynamic range (transient response) is substantially improved
and power output is substantially increased - all advantages for hi-fi but not
so good for a guitar amp.

This is because
many guitarists, in order to attain or emulate a particular "sound" -
and thus for them "normal" usage of their guitar amplifier - operate
the amplifier into the severe distortion range by simply driving to full output
or more. Some operate in the sustained overload range continuously, using
reverb, tape echo, electronic echo or acoustic feedback as a musical effect.

In a typical
commercial guitar amplifier - particularly those with tube rectifiers - the
power supply will collapse and simply run out of puff when overloaded,
resulting in substantially reduced B+ supply voltage, lower power output and
increased (severe) distortion as the output signal goes into square-wave like
response. But the use of silicon diodes changes the tube characteristics
insofar as the normal limit on Plate current as controlled by the Screen-Grid
is removed, allowing Plate current to increase in proportion with the signal up
to a maximum "saturation" point where more drive in does not produce
more power out.

Note also that to
maximise output power and minimise distortion in a Class B amplifier, it is
vitally essential to balance each half of the push-pull pair of tubes to
ensure DC current in the output transformer is reasonably equal. The more
tubes in the output stage the harder this is to achieve.

Unfortunately,
most high-power (100 W RMS +) guitar amplifiers do not provide individual
grid-bias adjustments for the output tubes. Under such conditions, to achieve
balanced DC Plate-current it may be necessary to set Grid 1 bias control
voltage at maximum signal - not at quiescent (zero signal) as is
popularly expounded, and to mix and match the tubes either side of the output
transformer to attain reasonably equal balance in total Plate-current per side.
(This may result in some unbalanced DC hum at zero signal).

In an amplifier
having a single bias supply it may be also necessary to modify the circuit to
install a means for balancing AC signal drive voltage into the output tubes
because, as well as being dependent upon primary Grid #1 DC bias voltage, the
Plate Current will also be dependent upon AC input volts to Grid #1. Note
though that the downside to this is that when a tube is replaced, re-biasing is
essential.

For example, I
conducted a test with a Marshall Model 1959 100W super-lead amplifier, with 4 x
EL34 tubes running at 520 VDC B+. When silicon diodes were fitted to the
Screen-Grids instead of the designed 1000 ohm grid-stopper resistor - one to
each Screen-Grid, power output increased from about 90W RMS in OEM
configuration to about 160 W RMS with diodes. Screen-grid current was
negligible, suggesting the tubes behave like triodes, however Plate current
increased to about 180 mA per tube, which increased net plate dissipation at
full output to about 55 Watts - a certain recipe for very short tube life
considering the EL34 has a rated Plate-dissipation of 25 W.

Plate current
then in this situation is primarily controlled by Grid #1 alone.

In this case the
amplifier simply gets louder and louder without the usual breakdown signs
before clipping, enabling the amplifier to be overdriven continuously to
self-destruction. In this particular amplifier, the power transformer is rated
at about 250 VA, and a quick calculation will show this component will also
have a very short life expectancy with silicon-diodes to the Screen-Grids - but
the sound is great!!

For Tetrodes and Beam
Power Tubes just delete the Grid #3 output transformer transformer tap, and
its separate Grid #3 DC supply - a Tetrode or Beam Power Tube will have only a
Grid 2 in the circuit.

Note: Westinghouse declared
the following in their June 1941 user instruction sheet supplied with each 803
tube:

Note the reference to
"Tetrode" connection and its effects.

This design feature offers
a range of options to the DIY designer/constructor.

Important Note: In the case of a Beam Power Tube,
the beam forming plates in a Beam Power Tube are not normally connected
to an active circuit element - ie are usually internally connected within the
tube to its own Cathode by the tube manufacturer. If available as a separate
connection, they should be externally connected to the Cathode in the usual
way.

Note: If a voltage
measurement is taken either side of the silicon diode in an ultra-linear
configuration circuit - ie on the screen-grid side and on the transformer side
- obviously a reading will be evident. The reading on one side will be out of
phase with the reading on the other side. DC current through the diode can be
measured by inserting a small resistor (10 ohms) in series with the diode on
the supply side and reading the current through the resistor.

The reason for the use of a
non-polarised capacitor is simply that polarised electrolytic capacitors offer
asymmetrical impedance to the current flow - depending upon the direction of
the current through the capacitor. Most importantly for high-fidelity
reproduction, polarised capacitors offer symmetrical characteristics to forward
or reverse AC current. This is evidenced by the widespread use of polarised
capacitors in loudspeaker crossover networks (which are purely AC) - even in
the cheapest commercial speaker systems.

The reason for the relatively
large value of capacitor suggested is that it is in series with the load on the
Plate and Grids, forming an LC or RC (depending upon the output stage
configuration) series network. If the value of C is low then peaks or resonance
can occur in the audio range - particularly in the mid to high frequency band
where harmonics are present.

This capacitor serves to
effectively AC short-circuit (or bypass) the DC power supply and thus eliminate
the power supply and its components from the AC signal path, and to ensure that
any shortcomings in polarised electrolytic capacitor performance are eliminated
- but in such a way that the signal is not significantly aurally affected.

In the case of the Plate
circuit of Triode, Tetrode, Pentode, Beam Power Tube and Ultra-linear wide-band
amplifiers, the non-polarised capacitor should be installed at the centre-tap
of the output transformer.

In the case of amplifiers
having separate Plate, Screen-grid and Suppressor-grid DC power supplies this
requirement also applies to each of the Screen-grid and Suppressor-grid DC circuits
as applicable.

To ensure adequate low
frequency response each of the separate circuits must have as large a value as
is practicable.

A 400 VAC (560 VDCW) Motor
Run Capacitor is ideal for this function and they are readily available in
non-polarised polypropylene construction. The voltage rating must be adequate
to handle the sum of the B+ DC voltage plus the AC rms signal output voltage.

I would recommend a parallel
connected non-polarised capacitor arrangement, to provide 100 to 200 uF total.
For higher B+ voltages the capacitors can be connected in series (800 VAC 1120
VDCW), noting that when seriesed, the effective capacitance is halved.

A further refinement is to
apply the "Rule of Hundredth's", which says that instead of using a
single capacitor - eg a large electrolytic, a bank of capacitors is installed.
Each capacitor is one hundredth the value of its adjoining capacitor. For
example, instead of installing one single 100 uF electrolytic, instal a 100 uF,
1 uF, 0.01 uF, 0.0001 uF etc. wired in parallel. Lead length of the smaller
capacitors must be kept a short as possible to prevent stray capacitance or
inductance. It is also important to ensure all capacitors installed are capable
of coping with the applied voltages without stress.

Of course the bypass
capacitor is installed on the line (source) side of the Screen-Grid-stopper
resistor and/or diode, as the case may be.

Note: This method may result
in resonances in the audio range, particularly to harmonics, resulting in
sibilant accentuation and "hissy" voice. It is thus not suitable in
all amplifiers.

Note 1: Please note that the irrespective of
the value of the electrolytic filter caps - even to many thousands of uF - the
value of the non-polarised cap is still critical in relation to tone, or
spectral balance, across the audio range. To optimise tone experimentation is
essential.Note 2: Modern "fast" capacitors
have a different tone to older oil-filled types. It may be necessary to use
oil-filled paper caps to obtain a smoother, less harsh tone. It all depends
upon the circuit design and componentry used. There is no definitive answer.
Unfortunately the older oil-filled paper caps are physically larger so need
more chassis space. They also do not usually come with flying leads, which makes
wiring more difficult because the terminal lugs are exposed. Ensure voltage
rating is adequate.

5.1.2
Electrolytic Bypass Capacitors to B+ Supply

Please note that as explained
in my Power
Supplies page, notwithstanding the obvious benefits from the use of
non-polarised capacitors as described above, it remains essential to good
transient response to instal large values of electrolytic capacitors to store
adequate power to satisfy the demands for transient peak signals.

The question is "how big
a value"?

In the case of power
requirements, full particulars are provided in my Power Supplies
page.

But what about "sound"?

It has been demonstrated by
early amplifier designs produced in the 1940's and 1950's that the frequency
response - even in the highest quality equipment, tended to roll-off at both
high and low frequencies.

Frequency response usually
deteriorated as power output increased. This was an attribute ("power
response") not usually presented in the glossy sales brochures.

Modern digital recording
techniques and playback media have dramatically increased both low and
high-frequency response for the typical recording, challenging even the very
best of tube audio amplifier designs - particularly when played at high-volume.

But all is not lost!!

Modern capacitors offer
improved performance and reliability over their ancestors. They are also
available in values unheard of in the days of tube rectifiers. (eg 32 uF was
"large" in the 1950's but now 100,000 uF is "medium.")

The thin film caps of today
charge and discharge much faster, enabling larger values to be used in common
applications where large values were previously forbidden - such as interstage
coupling caps.

Modern capacitors offer
reduced unwanted side effects, such as inductance, leakage and resonance, so
offer improved high-frequency performance and audio clarity.

One effect of this is a
reduction in the level of negative loop feedback needed to offset roll-off in
frequency response of an amplifier - a definite advantage.

In the case of low frequency
performance we can retrospectively improve old designs by using higher values
of filter bypass capacitors.

Since the bypass capacitor
forms the return AC circuit for each stage in the amplifier, it follows that
the impedance of the bypass (filter) capacitor at any given frequency will be a
portion of the total impedance of the circuit being bypassed. Now since the
impedance of a capacitor varies in direct proportion with the value of
capacitance at any given frequency, it follows that providing we reduce the
value of the capacitor's impedance to a value that has minimal effect upon the
circuit, then we can attain improved performance from the circuit.

Another way of explaining the
concept is that traditional tube electronic design engineering principles
assume that the B+ rail is at AZ zero (or earth) potential - regardless of the
applied DC voltage.

This is a practical approach
for many applications, but ignores the reality that capacitors - particularly
those of the electrolytic variety - have their own characteristics, which are
injected into the circuit and therefore MUST influence the sound we hear.

Since the capacitive
reactance of a 25 uF capacitor at 30 Hz is nominally only 212 Ohms, it suggests
that the influence of the capacitor may be negligible. However if the
inductance of the capacitor is only 0.1H then the inductive reactance in the
capacitor is 3140 Ohms at 5 kHz - and more at higher frequencies.

So we can see that increasing
capacitive reactance will decrease low frequency performance and decreasing
capacitor induced inductive reactance will increase high-frequency performance.

Since measured inductance and
capacitance values from typical large capacitors - eg above 500 uF - do not
appear to display the characteristics experienced aurally there may be an
alternative explanation. It may be that the bypass capacitor forms an RC or LC
series network with the plate resistor and/or output transformer, forming a
bass boost circuit. This is easier to see with the output transformer, where
either one end (in the case of a single-ended output stage) or the centre-tap
(in a push-pull output stage) forms the terminal at which the output is taken
off. but in the case of driver stage it would seem the series decoupling
resistor causes the formation of the RC network that creates the bass boost effect.

Either way it is desirable to
tune the network to the preferred frequency - eg 40 Hz

The question is what is the
magic number?? How big to we need to go??

To attain extremely good
low-frequency response, from practical experimentation I would suggest the minimum
value of the bypass capacitor for any stage in an amplifier or pre-amplifier,
including the output power stage, may be calculated by dividing the constant
15,000,000 (15 million) by the value of the plate load resistor (or
cathode-follower cathode load resistor) for that stage.

This approach will produce
values of 150 uF for a 100k plate resistor and 2,250 uF for a 6,600 Ohms Plate
to Plate output stage.

If those values appear
frightening, then try a lesser constant - say 10,000,000

If the B+ supply supplies
more than one stage or, in the case of some phase-splitters having more than
one plate resistor, then the value of the resistor should be calculated as the
average of all the resistors in the circuit.

If the B+ supply supplies
a stereo pair of amplifiers then the value of the plate load resistor used to
calculate should be half of the single channel value - ie each bypass capacitor
should be twice the size as for a single channel.

If the output stage uses more
than one pair of tubes from a single transformer, then use the transformer
actual nominal rated load impedance - not the effective impedance as seen by
each pair.

It will be noticed that these
values of capacitance are substantially higher than convention, however this is
what I have determined form extensive critical listening tests.

The general object of this
design approach is to introduce a circuit resonance induced tonal
characteristic that is pleasing.

Too high a value of capacitor
will deliver undefined bass, so some tweaking may be necessary.

It is desirable for all
stages to have the same tonal characteristic, so this formula assists to
achieve that. In other words, each stage should ideally have proportionately
the same tonal or frequency characteristic.

Note 1: This "rule
of thumb" formula deals only with tonal characteristics - power (energy)
requirements for the output stage are not covered by this. However it is easily
seen that the values produced by this design approach will supply adequate
power for most applications.

Note 2: Regardless of the
value of electrolytic capacitors used for bypassing and filtering,
non-polarised bypass capacitors are still essential for good high-frequency
performance and low intermodulation distortion.

Note 3: In the case of guitar
amplifiers the above formula is not applicable because the lowest frequency to
be reproduced is about 80 Hz. In this case a constant of say 10,000,000 or less
would be appropriate.

Note 4: In the case of bass
guitar amplifiers, where the lowest frequency to be reproduced is around 40 Hz,
then for outstanding results the constant needs to be in the region of
20,000,000. This value of capacitance will also deliver adequate power to the
output stage.

Note 4: Large values of
capacitor can be lethal, so discharge resistors should be installed as per
instructions provided in my Power Supplies
page.

RATIO OF SCREEN-GRID AND
SUPPRESSOR-GRID BYPASS TO PLATE BYPASS CAPACITORS

5.2.1
Introduction

Since the advent of
Tetrodes and Pentodes, it has been standard practice to instal a small bypass
capacitor from the Screen-grid to ground (Cathode) in Tetrodes and Pentodes
used in voltage amplifier stages.

This is shown as
C2 in the figure below.

The purpose of
the bypass capacitor is to reduce Grid to Plate capacitance, remove undesirable
audio and high-frequency signals such as RF components from the output before
the load, and to improve the decoupling and stability of the stage.

In voltage
amplifier (driver) stages the value of this capacitor has historically been in
the order of 0.1 to 0.5 uF, the latter value being considered by Radio
Engineers to be adequate for good quality audio purposes. A similar situation
exists in RF power amplifier stages.

It should be
noted though, that this value of capacitor is usually associated with small
tubes in high-impedance circuits, such as the EF86, 6AU6, 6U8, 7199 etc, where
the value of Screen-grid supply resistor may be in the range 100k to 1 Meg Ohms
with very low Screen-grid current, .

For the
theoretically minded, the formula for
calculating the value of the bypass capacitor in a voltage amplifier stage is
given courtesy of the Radiotron Designers Handbook, 3 rd Edition (1940).

5.2.3
High-fidelity Audio POWER AMPLIFIER Applications

Notwithstanding
the above historic convention, in high-fidelity audio POWER AMPLIFIER
output stages a very different situation applies.

If we think for a
moment, it can be easily seen that there is both a DC and an AC signal path
from the negative terminal of the bypass capacitor up through the capacitor to
the Screen-grid external to the tube, thence from the Screen-grid to the
Plate inside the tube. The latter will be the case (even if we do not
want it) because the Screen-grid is negative to the Plate.

As explained
above, the portion of Screen-grid current in a power tube can be quite high -
depending upon output stage configuration and applied voltages to the
Screen-grid and Plate respectively.

Since the
internal impedance of the bypass capacitor will be relatively small and the
Screen-grid is operating independently of Control Grid #1, it follows that the
magnitude of the current flowing from Screen-grid to Plate will depend more or
less entirely upon the actual applied DC voltage between Screen-grid and Plate
and the value of the Plate load - which will be also seen by this secondary
circuit.

That is
"secondary" to the primary Cathode to Plate circuit.

Obviously, the
higher the difference between actual applied DC voltage between Screen-grid and
Plate, the more current will flow.

In Tetrode and
Pentode and Beam Power Tube applications where the Plate and Screen-grid
operate at the same DC voltage, including Triode and Ultra-linear connections,
in the conventional and very common configuration shown above, it is suggested
by most writers that the Plate will function as the primary anode so long as
the Plate signal voltage does not drop below the DC Screen-grid voltage. In
this case, tube manuals show that about 10-20% of Cathode current is lost in
the Screen-grid circuit.

Since the bypass
capacitor C1 is common to both Plate and screen-grid circuits, in terms of
frequency response and dynamic response whatever happens in one will happen in
the other.

This is because a
secondary DC and AC circuit is established between Screen-grid and Plate, with
the Screen-grid forming the negative terminal/element.

In high-voltage
amplifier designs, the voltage between Screen-grid and Plate may be in the
order of several kilovolts (kV).

Since the lower
leg from Cathode to Screen-grid is through the bypass capacitor, and that is
external to the tube, it follows that the limiting factors to current flow in
this secondary circuit will be the Plate load impedance for AC current (the DC
resistance of the output transformer is negligible) and the value of the
Screen-grid resistor if used (either voltage dropping resistor or grid-stopper)
for both AC and DC current.

For explanatory
purposes, the internal tube resistance between Screen-grid and Plate may be
assumed to be zero.

It follows then
that the Cathode to Screen to Plate secondary AC circuit is in parallel with
the Cathode to Plate primary AC circuit.

As explained
above, the value of Screen-grid current - and therefore its contribution to
audio power output, can be significant.

Since the current
in both circuits combine together in the negative to positive Screen-grid to
Plate section of the circuit inside the tube (and thence common return to the
Cathode via the B+ filter capacitor at the output transformer centre-tap) it
follows that any difference in the audio signal between primary and secondary
circuits will be apparent at the output transformer.

For example, if
the Screen-grid bypass capacitor is too small, that portion of audio power
output contributed to by the Screen-grid secondary circuit as described herein,
will not have the same low-frequency response as the primary Plate circuit and
therefore low-frequency power output will be proportionately reduced.

For example, in
the Mullard
High-fidelity Pre-amplifier circuit, typical of conventional design, the
Screen-grid bypass capacitors C9 is 80 times the value of C8, and C17 is 160
times the value of C12.

The value of the
Screen-grid bypass capacitor will also affect the operation of the Plate circuit
B+ bypass capacitor, because the B+ Screen bypass cap and B+ Plate bypass cap
are in series in the signal circuit.

Hence it may be
deduced that:

Where a
separate Screen-grid power supply is provided, it is most important for
full-power hi-fi reproduction at very low frequencies, and for signal balance
within the tube, to ensure the value of the final Screen-grid B+ bypass filter
cap is not less than the value of the Plate circuit B+ bypass filter cap.

In the above
diagram the Plate bypass capacitor is shown as C1 and the Screen-grid bypass
Capacitor as C2.

Note: Even though
the installation of a voltage dropping resistor or grid-stopper resistor to the
Screen-grid (R3 in the voltage amplifier circuit diagram above) may reduce AC
and DC current, the issue of AC impedance remains - hence the grid resistor may
be disregarded for this aspect of hi-fi design.

It will be
readily seen that in a power output stage where the Plate and Screen-grid share
a common AC circuit return at the power supply, and regardless of the use of a
grid-stopper or dropping resistor (R3 in the above diagram) or not, the above
argument does not apply because both Plate and Screen-grid will share a common
B+ bypass capacitor.

BUT - where the
screen-grid is supplied through a filter choke from the centre-tap of the
output transformer at C1, and bypassed by its own electrolytic capacitor C2,
the configuration shown above is still is applicable.

However bear in
mind that the regulation of Screen-grid voltage is also extremely important so
a separate supply should remain an essential design element.

The above
comments also apply to the Suppressor-grid

5.2.4
Cathode Bypass Capacitor

In the case of Tetrode or
Pentode amplifier output stages having Cathode Bias, it is usual to instal a
Cathode-bypass Capacitor as shown in the following diagramme:

. It can be readily seen from the way the circuit is drawn
that the Cathode-bypass Capacitor is in series with the B+ Plate and
Screen-grid supply.

It follows that
if the Cathode-bypass capacitor is too small, that portion of audio frequency
response determined by the Cathode circuit will not have the same low-frequency
response as the primary Plate circuit and therefore low-frequency power output
will be proportionately reduced.

From the above explanations
it can be demonstrated that the value of C3 must be
at least equal to the value of C1.

The more theoretically minded
can calculate the actual values needed for equilibrium in the frequency
response characteristics for each part of the circuit.

Where a separate Screen-grid
power supply is used, as shown below,

it can be readily seen that
the Plate and Screen-grid bypass capacitors C1 and C2 are in AC parallel but
the set of both is in series with the cathode bypass capacitor C3.

Hence it is also essential
that the nominal capacitance value of C3 is equal to the SUM of C1 + C2.

Where separate Cathode-bypass
capacitors are used to each Cathode, the above rule still applies for each
capacitor.

It is usual for Class A
amplifiers to use a single common cathode resistor and no bypass capacitor per
push-pull pair of tubes - this is obviously more AC linear.

5.2.5 Power Factor

Ideal values
for C1 and C2 and C3 can be in the region of up to 5,000 uF per push-pull pair of tubes - see warning re risk of
electrocution.

This extra step
is needed to tune resonances in the output stage circuit to match the
characteristics of the output tubes, output transformer and loudspeakers to the
signal "sound" or "tone" and listening room acoustics. Too
low a resonance may result in a "dull" or "flat" bass sound.
Too high a resonance may result in "bass boom" and loss of
definition.

To optimise the
output stage circuit and deliver maximum power from minimum output impedance
with maximum loudspeaker damping characteristics, changes in power factor
produced by the inductive reactance in the output transformer will ideally be
cancelled out by the capacitive reactance in C1.

To optimise the
value of C1 some degree of practical "trial and error"
experimentation is needed - a frustrating experience but one worth the effort.

Note C2 and C3
need to be suitably modified as described above and hereunder to maintain the
correct ratio to C1 - ie the same capacitance value.

Note also that to
overcome non-linearity in the AC bypass circuit, C1 and C2 may need to be
bypassed by a small stabilising capacitor in the region of 0.5 uF - but since
this small capacitor will directly affect the high-frequency response, it too
must be chosen with care regarding both size and material of construction.

IMPORTANT:

All of the
Plate Circuit output stage power passes through the AC circuit formed by the
output transformer and C1 - and whatever else shunts C1.

Since L1 is
directly connected to the Power Supply, it follows that the Power Supply is in
series with L1 and shunts C1.

This is
illustrated in the diagram below:

So to
minimise the effects of the Power Supply rectification and filter circuitry on
"sound", the value of L1 should be as large as can be practicably
sustained - noting that the DC resistance of L1 will directly reduce the
available B+ voltage to the Plates.

The larger
L1 is, the more AC current will be forced through C1 and less through L1 and
associated circuitry.

Another way
of expressing this is to say that whatever AC power is lost into L1 and its
associated source bypass components - all of which shunt C1 - the less linear
the output stage will be at low frequencies.

An inductance
of about 10 Henries (minimum) is desirable for L1.

This also
holds true for simple one stage filter choke systems, because the
rectifier/filter is always shunting C1 - thus will always affect its
performance and effect on "sound".

IMPORTANT:
To prevent instability in the power supply (which dramatically affects audio
sound as heard through the loudspeaker) it is desirable to ensure C2/L1 and
C3/L2 are of equal value. Where only C2/L1 are used then it is desirable that
C1 and C2 are of equal value. It is thus obvious that a three stage filter
enables a higher value of C1 than for a two stage filter.

Where the
Screen-grid B+ supply is taken from the output of a voltage doubler rectifier,
it may be the case that for convenience the Screen-grid supply is taken from
the mid-point of the two series-connected filter capacitors.

However if we
analyse the effect of this arrangement having regard to the above, it is easy
to see that the external Screen-grid bypass current is passing from negative to
positive through the lower filter capacitor.

It is also easy
to see that return external Screen-grid bypass current is passing from positive
to negative through the lower filter capacitor.

Consequently, it
is evident that the two currents will cancel out.

Bad move!!

So in other
words, this configuration is not desirable for high-fidelity reproduction, even
though it may appear to work satisfactorily in public address amplifiers.

There is also an
issue of the effect of differences in capacitor characteristics in the forward
and reverse directions, as well as the effects of distortion caused by the
output transformer.

There is also the
issue of excessive ripple in the DC supply.

One solution to
both issues may be to instal a 60 mA filter choke in the line (supply) side of
the Screen-grid B+, then add a second filter capacitor to the Screen-grid B+ at
the load side, to provide a separate path for forward and reverse currents.

The value of the
second (final) bypass capacitor should be equal to or greater than the value of
the seriesed pair.

5.2.8
Capacitive Voltage Dividers

Where the
Screen-grid B+ supply is taken from the centre-point of a pair of
series-connected electrolytic capacitors in the Plate B+ circuit, the same
problems arise as explained above.

NOTE: To ensure
equal voltage distribution over time (service life) series connected
electrolytics MUST be stabilised for equal voltage distribution by means of an
external circuit system such as shunt resistors (voltage divider network) or
power supply configuration.

Where
Cathode-bias is employed, in addition to the above it is also essential to
ensure the value of the Cathode-bypass capacitor(s) is not less than the value
of the Plate and screen-grid supply capacitors

Note: Where one
or more filter chokes are used in the B+ supply, the value of capacitors before
the choke(s) may be ignored for this requirement.

5.4
NEGATIVE LOOP FEEDBACK

Where negative loop feedback
is used from the loudspeaker terminals back to an early voltage amplifying
stage, it can be seen from the above that any defects in performance in the
output stage will simply be transferred back to the input, with the result that
the amplifier will always be in a constant state of correcting itself.

In real life we have the
following variables in the power stage system:

output
tubes having unequal and non-linear characteristics over their minimum to
maximum Cathode current range - ie Plate current v Control Grid voltage

the
Control Grid "grid current" varying with AC signal amplitude,

the
Control Grid AC bias voltage varying with AC signal amplitude,

in
ultra-linear systems the Screen Grid voltage varying with AC signal fed
back from the output transformer,

the Screen
Grid current varying with the AC circuit voltage between Screen and Plate,

the Plate,
Screen and bias DC voltages varying with total load on the common power
transformer - ie transformer/power supply "regulation".

unequal
characteristics in each half of the output transformer,

output
transformer frequency power response is less than the full audio
range available at the input terminals,

back EMF
from the loudspeaker reflected (transformed) into the Plate circuit
(loudspeakers have different "push" and "pull" cone
characteristics due to suspension construction and enclosure
characteristics) (Note the cone is tapered in the "pull"
direction and assisted by air pressure however it is cup shaped in the
"push" direction) - ie is acoustically non-linear. (This comment
does not apply to planar loudspeakers)

non-linear
impedance loading by the loudspeaker over the full frequency range -
typical 8 ohm speakers reflect frequency dependent loads between 6 ohms
and 100 ohms - ie reflected Plate circuit load can increase from the
nominal by a factor of 10 to 12 times

In output stages of audio
amplifiers it is common to use FIXED BIAS or BACK BIAS to apply
and control the Grid #1 - Control Grid bias voltage in the output tubes.

Designers always consider the
requirements for direct current operating conditions but often ignore
requirements for alternating current conditions - ie signal voltage.

The Grid #1 Resistor to each
output power tube forms part of the load for the preceding driver stage. Hence
for maximum efficiency and stability, the return path from the output of the
driver stage circuit back to its Cathodes should be direct and have very low
impedance.

Bias, by definition, requires
a voltage potential to be present between the Control Grid and the Cathode.

CATHODE BIAS usually results
in the Control Grid being at nominally 0 VDC and the Cathode at the required
bias voltage being + VDC.

FIXED BIAS usually results in
the Control Grid being set at the required bias voltage being - VDC and the
Cathode at nominally 0 VDC. In the case of FIXED BIAS it is essential to bridge
the difference between the central axis of the AC signal input and the power
tube Cathodes, such that the central axis is at the same AC voltage as the
power tube Cathodes.

It is essential for stable
high fidelity operation to ensure that:

a) the Grid
#1 circuit has a reliable and predictable low-impedance return path for
the AC signal voltage from the preceding stageb) the Grid #1
circuit has a reliable and predictable low-resistance return path for
the DC bias voltage between Grid #1 and the Cathode of the same tube.c) in a
balanced driver stage for Classes AB, AB1, AB2 and B, where the power tubes are
biased towards cut-off and driver stage output signal voltages are
symmetrically balanced about a virtual central axis having a nominal potential
of 0 VAC, it is imperative to ensure the junction of the two Control Grid
resistors of the output tubes be AC grounded to 0 VAC.

Thus in an
RC (resistor/capacitor) coupled amplifier, it is a fundamental requirement that
the Grid #1 Resistor to each power tube provides a return circuit path
to the Cathode of BOTH the preceding driver tube AND the power tube to
which it is connected.

This is usually, but not
always, through the earth or ground terminal of both AC and DC applied
circuits.

Fortunately, in most
amplifiers using FIXED BIAS or BACK BIAS, the Cathodes of the output tubes are
directly earthed (to the chassis), thereby providing a convenient return
circuit at 0 VAC potential.

6.2
Cathode Bias

In most amplifiers using
CATHODE BIAS, the Cathodes of the driver tubes are earthed either through a
bypassed or unbypassed Cathode resistor, hence the Cathode terminal is indirectly
AC earthed to complete the return circuit for the driver tube(s).

This principle applies
particularly to CATHODE BIASING of power tubes, where the Cathode
terminals may well be at say +50 VDC but simultaneously at 0 VAC. . (An
exception is where the output stage is in Class A and the common cathode
resistor is unbypassed to develop negative current feedback in the Cathode
circuit.)

Typical Triode-connected
Pentode output stage configuration with adjustable CATHODE BIAS.Note the absence of a bypass
capacitor on the common Cathode resistor.

This circuit features + or -
adjustment of common bias voltage and also fine adjustment of balance between
the tubes.

6.3
Fixed Bias Adjustment

It is common practice in
FIXED BIAS amplifiers to incorporate an adjustable bias control circuit,
incorporating an adjustable resistive network to enable precision adjustment to
the Control Grid voltage and/or Plate Current of the output power tubes.

Typically, this negative polarity
DC voltage is sourced from a half-wave or full-wave tube or solid-state
rectifier, filtered by a simple resistor/capacitor network.

Note: In this design, the 20
MF (uF) capacitor serves to bypass the AC return circuit from the driver stage
to ground, thereby providing a direct return circuit path to the Cathodes of
the driver tube. This capacitor is connected in such a way as to ensure the
Control Grids are equally earthed at all times through their respective Grid
Resistors, regardless of applied DC bias voltage. It also ensures the central
axis of the balanced signal input is at the same potential as the Cathodes - an
extremely important component.

6.4
Bypass Capacitor Material

In commercial amplifiers with
FIXED BIAS, to minimise cost the bypass capacitor is nearly always of the
polarised electrolytic variety, but this means that the AC return circuit is
not symmetrical. (Electrolytic capacitors have different characteristics in
positive and negative polarity circuits).

Consequently, to ensure symmetrical AC circuit configuration it is
essential to instal a suitable non-polarised bypass capacitor into the circuit
at a point closest to the Grid #1 of the output tubes as is practicable.

Typically this will be at
the junction of the two (or more) Grid Resistors.

If a driver transformer is
used, and its centre-tapped secondary is not directly connected to the power
tube Cathode circuit, then instal the bypass capacitor between the centre tap
of the secondary and earth.

It is therefore absolutely
essential to bypass (or wholly replace) the final polarised electrolytic
capacitor, with a suitable non-polarised high quality mica, polyester,
polypropylene, paper or oil-filled paper capacitor, having a suitable value (of
say 1.0 uF or a 10 uF or more motor start capacitor as above for audio), to
provide an AC bypass at all signal frequencies and under all operating
conditions.

This small capacitor serves
to effectively short-circuit (shunt or bypass) the DC power supply and thus
eliminate the power supply and its components from the AC signal path of
both the driver tube and power output tubes, to ensure that any
shortcomings in polarised electrolytic capacitor performance are compensated -
but in such a way that the signal is not significantly aurally affected.

Importantly, it also provides
an automatic safeguard against the adverse effects of poorly contacting bias
potentiometers and/or adjustable wire-wound resistors.

Normally, the value of the
bias supply capacitor will not exceed 10 uF (to ensure fast charging to full
bias voltage before the output tubes heat up and commence to conduct Cathode
Current) so an extra one or two uF will not significantly affect the charging
circuit performance.

If preferred, to achieve
the same end result as described above, the final Control-grid power supply
filter polarised electrolytic capacitor (and the first as well if so inclined)
can be wholly replaced with a polyester or paper or motor start capacitor of
say 8 uF value and having a suitable DC voltage rating.

Suitable capacitors may also
be reclaimed from unwanted fluorescent lamp-holder assemblies.

For the ultra-fastidious, the
"Rule of Hundredth's" may also be applied.

Note: There is no practical
limit as to the value of the bypass capacitor, provided the bias supply is
capable of charging it quickly to ensure bias voltage is present when the power
tubes warm up and commence conducting. I have successfully used values around
100,000 uF, shunted by suitable non-polarised polypropylene caps.

SINGLE OR DOUBLE STAGE PI
FILTER IN EACH POWER SUPPLY
FILTER CAPACITORS OF EQUAL CAPACITANCE
EACH INDUCTOR FILTER CHOKE BYPASSED BY A REVERSE BIASED SHUNT DIODE (SNUBBER
SYSTEM)

The concept is
shown here:

.The purpose of this configuration is to
stabilise the voltage into the B+ rail. (Note: "Stabilise" means free
from self-oscillation)

Ripple voltage
appears at the input.

The ripple
voltage includes harmonics from the mains and spikes from the rectifiers - both
of which pass through the inductor filter choke and appear at the output
transformer centre-tap (AC earth) - so thereby appear in the output stage
circuit as a signal modulating voltage and, by transformation, audible in the
loudspeaker as distortion.

The shunt diode short-circuits
the choke in the reverse direction

Any unwanted
spurious voltage - such as switching spikes - are either eliminated from the
system or dramatically attenuated

The use of a
small capacitor or capacitors connected in parallel with the first electrolytic
cap provides a path for spikes and any other mains harmonics etc to be diverted
to the negative rail.

The output
voltage is therefore, for practical purposes, spike free

The capacitors
should be of equal value and type so that the two capacitors operate in sync
with the input voltage - ie the current in is matched by an equivalent current
out, synchronised in time. That is to say the charge and discharge rates over
time should be equal. If they are not equal, higher than normal rectified ripple
mains frequency (ie 100 Hz or 120 Hz) ripple spikes can be generated within the
filter system.

The benefit of
the PI filter is that even a small value of choke inductance does wonders for
substantially reducing ripple.

Elimination of
Ripple is vitally important for true hi-fi because the B+ rail is the AC ground
for the signal.

If this voltage
"dances" around with spurious ripple signals and/or the actual
centre-point of the output transformer not being at its nominal zero AC volts
true centre, then the output will be modulated by this voltage. It can either
be bypassed to ground via the filter capacitors or transformed to the
loudspeaker (in which case it will appear in the negative feedback system - if
used)

This system is
very stable and enhances clarity and frequency response

This object
may be easily and economically realised by installing a series connected silicon
diode rectifier into the B+ supply - having its arrow pointing towards the load
- between the Power Supply output terminals and C1 -
ie AFTER the Power Supply and BEFORE C1.

Note that C1
must always remain part of the Amplifier, because it forms a direct path in the
AC output circuit.

So regardless of
whether the Power Supply is of the simple centre-tapped full-wave or full-wave
bridge rectifier or full-wave voltage doubler rectifier, or it has a capacitor
filter, choke filter or some other combination of filter - or even includes a
voltage stabiliser or regulator - to ensure the OPTIMISED VALUE OF C1 is not
altered by shunting the Power Supply circuit, significant improvement to
performance can be made by installing a series silicon rectifier diode as
described.

To ensure cool
operation and reliability, the current rating of the rectifier should be
generous in relation to the power requirements of the amplifier - eg 3A or 6A.
The rectifier can be of the same specification as that used in the Power Supply
B+.

Where a separate
Screen-grid Power Supply is used, then the same principles apply to C2.

Where an inductor - such as
L1 above - is used in the Power Supply, it is preferable for the diode to be installed
after the inductor - to ensure the Power Supply is completely isolated from the
amplifier.

This method makes it easier
to adjust the tonal balance or "tone" of the amplifier power stage -
because fewer components are now in the signal path - therefore there are fewer
interactive variables to deal with when changing component values.

The same principle apply to
L1 and C2.

It will be seen for the above
that in the case of simple filters having just one capacitor after the
rectifier - and nothing else - the requirement described is met. Unfortunately
that simple configuration results in significant hum and ripple and poor
quality direct current (DC) - so is not recommended for high-fidelity
amplifiers.

DECOUPLING OF EACH STAGE OF THE
VOLTAGE/POWER AMPLIFIER
FROM ITS NEXT-FOLLOWING ADJOINING STAGE BY MEANS OF A
SERIES CONNECTED SILICON DIODE

Having described how
decoupling of the amplifier from its power source facilitates and enhances performance,
it follows that the same principles apply to driver stages.

In a conventional amplifier,
the stages are connected in a "cascade" configuration.

That is to say, each stage
"cascades" into the next - just as a stream "cascades" over
its riffles as it runs down a slope.

Conventional theory says that
up to three stages may be connected to the same B+ supply before instability
becomes a problem.

However:

For optimum
performance each stage should be a self-contained discrete circuit - wholly
independent from those before and those after.

The usual method of
decoupling is to instal a series resistor into the B+ supply between stages.
But this produces significant voltage drop when all or any tubes draw current
over the steady state condition.

This convention also allows
signal from one stage being conducted to an adjoining stage when there is a signal
voltage difference between those stages.

So to produce effective
decoupling between stages and to prevent each stage sharing the bypass
capacitors and associated circuitry installed in adjoining stages - before or
after - it is essential to decouple effectively and completely.

A "system" is
defines as a series of "processes", each having an input and an
output. The ideal system is self-adjusting orself-correcting, by means of a
feedback loop (s).

It follows that in such a
system what happens in one part also happens in others.

In the case of an amplifier,
the signal is present on the B+ rail at each individual stage. In some cases
the signal is in phase with other stages, and in other cases the signal is out
of phase with other stages.

Therefore the B+ rail is
common to all stages. Looking at it another way (ie from the B+ line), the B+
rail is the output from a simpleparallel mixer system - each
stage being an input to the mixer.

Clearly then the number of
stages and those in phase and those out of phase will affect the
"sound" of the amplifier.

Conventional design
principles regard the junction between the plate load resistor and the
immediately adjacent filter cap as being at AC "earth" - because the
filter cap is regarded as being a very low (insignificant) impedance path for
the AC in the circuit

However practical tests
reveal a different story

I submit any given
electrolytic cap has different characteristics when comparing its forward and
reverse current flow. One way todemonstrate this is to replace
an electrolytic in the B+ rail with a non-polarised motor start cap - the
difference in tonal quality ishuge.

This proves that the junction
of plate resistor and filter (signal bypass) cap is not at AC earth at all, but
at some point above it - or at best the AC signal circuit will be affected or
influenced by the series connected bypass capacitor's inherent internal
characteristics.

Now since we are talking a
VOLTAGE amplifier we do not need to consider the CURRENT in the B+ rail but the
VOLTAGE

Specifically, the TRANSIENT
PEAK VOLTAGE

When a transient signal
appears on the B+ rail at any point it will appear across the entire rail - not
just at the stage in which itoriginates

So if, for example, the
phase-splitter B+ voltage momentarily sags, the signal from the previous stage
- which does not suffer asmuch sag because it is a low
current stage - will flow to the phase-splitter stage and be mixed with it at
the filter cap

Ohm's Law applies, so
wherever there is a voltage difference you will see current flow - in whatever
direction it chooses

The purpose of the series
diode in the B+ rail is to prevent SIGNAL voltage from transferring forwards
(positive feedback) in thecircuit whenever a voltage
difference in the B+ rail appears

It will not of course reduce
negative feedback between stages through the B+ rail

Provided a suitable diode is
installed to each and every stage in this way, this simple device prevents the
AC in any single-stage circuit from being shared by an adjoining upstream
(line-side) stage - thus ensuring all of the signal current is passed through
the stage discrete bypass (usually "electrolytic") filter capacitor.

That is to say, in a
push-pull amplifier stage, the mid-point central axis of the AC push-pull
signal input voltage MUST be directly connected to the mid-point central axis
of the the AC push-pull signal output voltage - ie the INPUT voltage to the
driven stage.

Transformer
Coupling/Driver System

In a conventional transformer
coupled circuit, the supply terminal of the single-ended driving side, or
centre-tap of the push-pull driving side is earthed via the B+ bypass cap.

In this case, the return
circuit for the AC signal is through the transformer primary back to the driver
cathode via that B+ bypass cap.

Being connected to earth it
follows that the driver cathode bias resistor should be also connected to
earth. In the diagram below that would be via a centre-tap on the driver Triode
AC filament transformer.

In
some Class B designs using zero bias tubes, and in a cathode bias design, the
centre-tap is usually connected to ground as shown.

But in a fixed bias design,
the centre-tap of the transformer secondary is usually connected to the bias
supply.

Thus when the centre tap of
the secondary is connected to the bias supply then the primary and secondary
centre-taps do not share a common connection - except through the bias supply
earth point, which is at positive polarity in respect to the bias voltage.

This can be seen in the
following design, where the single-ended driver is coupled to a push-pull
output stage.

Note the cathode bias
resistor and bypass capacitor in the driver stage are grounded but the
centre-tap of the transformersecondary is connected to the
negative voltage DC bias supply.

But being a transformer none
of that matters, because the primary and secondary are two completely
independent andisolated circuits.

RC Coupled Driver Stage

However when a tube RC
coupled driver circuit is used with either plate or cathode output - then the
two circuits must share a common in and out AC signal axis - or else the sound
will be affected.

This is explained below.

Fig 1: Cathode Biased
Output Stage

In a cathode biased amplifier
the requirement for driving and driven circuits to share a common connection is
normal, conventional design practice. A typical standard configuration is shown
in Fig 1.

Notice how the central axis
of the driving and driven circuits is common. In this case the central axis is
at ground potential.

Since the circuitry in both
halves of the push-pull circuit are symmetrical and exactly equal it follows
that the AC signal voltages in both halves of the push-pull circuit are
symmetrical and exactly equal.

Notice how the Cathode
circuit of the driving stage is not connected to ground - as is conventional
design practice - but to the bias supply for the driven stage.

This is the profound
difference.

Further explanation is
provided below.

Fig 6: Conventional Fixed
Bias System.

Fig 6 shows the AC signal
path in a conventional fixed bias system.

Notice how the AC signal must
pass through the bias supply components - particularly the bypass electrolytic
filter capacitor (usually of the electrolytic variety). In this case, the bias
supply shown is simple, but in most designs the bias supply is complex and
often comprises chokes and other harmonic producing or refining components - in
complex inter-relationships.

Moreover, it is common
practice to use half-wave rectification in bias supplies, so the residual
ripple voltage will appear as a series of pulses in series with the grid
driving circuit.

Consideration must also be
given to the effects of spurious harmonic and transient spike signals injected
into the bias circuit from the bias supply mains transformer.

Since this
part of the driving circuit is common to both halves of the push-pull drive, it
follows that the signal will ALWAYS be modified by the bias supply
characteristics.

If that was not enough to
contend with, it can also be seen that the bias supply negative DC voltage is
permanently in the signal circuit.

Furthermore, the bias voltage
is offset negatively to the centre-line axis of the signal voltage, which is
nominally at ground potential. The greater the applied bias voltage the greater
the offset.

Since the AC signal
alternations in both halves of the push-pull circuit have both positive and
negative polarity, it follows that the bias supply voltage will enhance the
signal in the negative alternation and offset the signal in the positive
alternation.

This DC voltage and current
source will try to support the flow of AC signal sourced energy to the grids
when grid current flows in Classes AB1 and AB2 or B - and also Class A if gas
is present in the power tubes . However to get to the grids it must first pass
through the driving tube circuit. Since the interstage coupling capacitor will
not pass DC it follows that the DC component of grid drive power will be
dissipated into the plate resistor of the driving tube.

But along the way it can
increase the cathode bias voltage of the driving tube (which will attenuate the
signal) whilst simultaneously reducing the plate voltage (which will attenuate
the signal and reduce AC output voltage) - thereby affecting the
capability of the driving stage to respond to the signal input.

Since the plate load resistor
of the driving tube provides an alternative return circuit (to ground) for the
signal voltage it follows that the conventional fixed bias configuration which
adds DC currents into the signal circuit includes elements guaranteeing circuit
instability. When negative loop feedback is added to such a circuit - eg from
the speaker secondary windings of the output transformer back to the driving
stage - then the result must be difficult to predict or control.

Finally, the bias supply
final shunt load resistor is in series with the output tube grid resistor and
forms an integral part of the grid to cathode circuit of the output tubes. It
is common to see relatively high values of bias supply final shunt load
resistor compared with the output tube grid resistor value - eg 47k to 100k.
Sometimes complex bias adjusting networks are also in there. It follows that
the closer the (total) bias supply resistor is to the grid resistor value then
the more effect the bias supply characteristics will have upon the AC signal
behaviour.

Fig 5 shows the symmetrical balanced AC signal drive system whereby the
AC signal OUTPUT from the Driving stage circuit and the AC signal INPUT of the
Driven stage circuit have a common central axis at reference voltage potential.

In this example, the bias is
set at -40 VDC.

Notice how using this system
as shown in Fig 4, the signal path circuits on both sides of the push-pull
circuit are exactly equal and balanced - in a similar manner to that of Fig 1:
Cathode Bias Output Stage.

Moreover, the signal path is
direct between both driving and driven stages and is completely independent of
and therefore unaffected by the bias supply.

Note that the grid to cathode
circuits of the power tubes is the same for both Fig 5 and Fig 6 designs.

This system is very stable
and provides enhanced frequency response and reduced distortion.

It is entirely suitable for a
cathode-follower driver stage - ensure the base of the cathode follower load
resistor is connected to the bias supply as shown in Fig 5.

An added benefit for a
cathode-follower driver system is that the cathode load resistor now spans the
bias voltage as well as the B+ voltage. Consequently the cathode follower load
resistor may be made substantially larger whilst still maintaining the same
plate to cathode voltage in the tube. This can offer increased output voltage
to the driven stage.

Load Resistor

Generally speaking, the
greater the load resistance/impedance to the driving stage the higher the
output capability.

However there are practical
limits to this objective since output tube grid circuit resistance must be held
within manufacturers' published specifications.

To maximise the driver stage
load whilst minimising driven stage grid circuit resistance the RATIO of
driver stage load plate resistor to driven stage grid circuit load resistor
should be held within the range of about 1:1 to 1:2.

There is no
restriction or cost imposition upon the home hobbyist constructor to using
these concepts - the only restriction is on commercial exploitation - so if you
do not like it do not do it.

If you want your
hi-fi to improve its performance at minimal cost to you then experiment. The
concepts presented here do work and cost very little to implement.

However to those
who say that a product is only as good as what you pay for it, then
these concepts are of no value to you because they are free. You would be wiser
to spend a hundred grand on a commercial system and feel better. While you are
so doing, ask the manufacturer to justify the circuit design and component
choices to you - ie why the design is what it is and not some other alternative
approach. That is "why is it better?"

To those who
consider relating these concepts to RF applications - this is an audio focused
site - experiment at your own risk. It may be safer to stay with the tried and
true - waste a little power and live with it.

Intellectual property in the applied engineering concepts
expressed in this paper remains exclusively with the author.

IMPORTANT NOTICE

THE AUTHOR
MAKES NO CLAIM WHATSOEVER AS TO THE VALIDITY OR ACCURACY OF ANY STATEMENT,
INFORMATION OR OPINION CONTAINED IN THESE PAGES AND NO LIABILITY WILL BE
ACCEPTED FOR ANY ERROR OR OMISSION OF ANY KIND WHATSOEVER.

PLEASE
NOTE NO WARRANTY IS EXPRESSED OR IMPLIED AS TO THE WORKABILITY OR PERFORMANCE
OF DESIGN INFORMATION DESCRIBED HEREIN.