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Abstract:

Core loss in an inductor is measured with reduced sensitivity to phase
measurement error by connecting a reactive component to resonate with the
inductor and thus cancel a portion of the reactive voltage on the
inductor; reducing the phase difference between the inductor voltage and
current and making the observed power more resistive. The reactive
component may be a capacitor for sinusoidal excitation or an inductance
such as an air core transformer for arbitrary excitation.

Claims:

1. A method of measuring losses in an inductor or transformer, said
method comprising steps of resonating a winding of said inductor or
transformer with a reactive component in a series resonant circuit with
said winding to cancel a portion of a reactive voltage of a magnetizing
inductance of said inductor or transformer, a value of said reactive
component being chosen such that a phase difference between a measured
voltage across said resonant circuit and a measured current in series
with said resonant circuit is less than 80.degree., and integrating said
measured voltage and said measured current.

2. The method recited in claim 1, wherein said phase difference is less
than 30.degree..

3. The method as recited in claim 1, wherein said reactive component is a
capacitor.

4. The method as recited in claim 3, wherein said capacitor comprises a
plurality of discrete capacitors.

5. The method as recited in claim 3, wherein said capacitor is a variable
capacitor.

6. The method as recited in claim 1, wherein said reactive component is
an inductor

7. The method as recited in claim 6, wherein said inductor comprises a
variable inductor or tunable transformer.

8. The method as recited in claim 1, wherein said reactive component is
an air core transformer.

9. The method as recited in claim 1, including a further step of applying
a DC flux bias to said inductor.

10. A circuit for measuring losses in an inductor or transformer having a
core, said circuit comprising, a current sensor to measure current
through a winding of said inductor or transformer, and a reactive
component connected between said terminals of said sensing winding and
said inductor or transformer and having a reactance value to resonate
with said inductor or transformer such that a phase difference of a
voltage across said sensing winding and said current through said
inductor or transformer is less than 80.degree..

11. The circuit as recited in claim 10, further comprising a sensing
winding wound on said core and having one terminal of said sensing
winding connected to a terminal of said inductor,

12. The circuit as recited in claim 10, wherein said phase difference is
less than 30.degree..

13. The circuit as recited in claim 10, wherein said reactive component
is a capacitor.

14. The circuit as recited in claim 10, wherein said reactive component
is a variable capacitor.

15. The circuit as recited in claim 10, wherein said reactive component
is an inductor.

16. The circuit as recited in claim 10, wherein said reactive component
is an air core transformer.

17. The circuit as recited in claim 10, further including a further
winding on said core, and a DC current source in series with said further
winding.

18. The circuit as recited in claim 10, wherein said inductor or
transformer winding, said sensing winding and said further winding are
formed on two cores and respectively connected in series with said
further windings being inversely coupled to portions of said inductor on
respective cores.

19. The circuit as recited in claim 10, wherein said inductor and said
sensing winding are formed by a bifilar winding on said core.

20. The circuit as recited in claim 10, further comprising a radio
frequency excitation source, and an impedance transformer.

Description:

FIELD OF THE INVENTION

[0001] The present invention generally relates to power electronics
applications such as power supplies and regulators and, more
particularly, to accurate measurement of losses in inductors and
transformers used therein for optimization of their design.

BACKGROUND OF THE INVENTION

[0002] Virtually all currently available devices that includes any
electronic circuits include a power supply circuit to ensure that
appropriate power is delivered thereto for the electronic circuits to
properly perform their intended functions. Since a power supply circuit
does not otherwise contribute to the function of the electronics, its
efficiency is of paramount importance since it directly detracts from the
overall efficiency of the device. Thus, many sophisticated designs for
power supplies, converters and regulators (hereinafter sometimes
collectively referenced by the term power circuits) have been developed
and have achieved relatively high efficiency which is difficult to
increase through circuit design and/or operation.

[0003] For this reason, losses attributable to inductors and transformers
which are generally included in such power circuits are important
concerns. The power circuit cannot be optimally efficient without proper
magnetic cores for such passive elements while accurate measurement of
inductor and transformer losses is necessary for proper inductor and
transformer core design. Further, requirements for so-called
point-of-load power circuits where size and power density are
particularly critical have favored the use of high frequency switching
which complicates loss measurement, as will be discussed in greater
detail below.

[0004] Unfortunately, no satisfactory technique for measurement of core
losses in inductors and/or transformers is currently known. The
techniques currently in use for this purpose, while providing reasonably
useful results, present difficulties and sources of error such as
compensation for winding losses which are difficult to estimate or
extreme sensitivity to measurement discrepancy in regard to parameters
which are subject to errors that may be engendered in whole or in part by
the instrumentation required for the measurement, as will be discussed in
greater detail below. Prior to the present invention, no technique has
been known which is satisfactory for making measurements of core losses
sufficient to support development of optimal core designs for inductors
and transformers of power circuits designed for particular applications.

SUMMARY OF THE INVENTION

[0005] It is therefore an object of the present invention to provide an
apparatus and methodologies for measurement of magnetic core losses in
inductors and transformers operating at high frequencies.

[0006] In order to accomplish these and other objects of the invention, a
method of measuring losses in an inductor or transformer is provided
comprising steps of resonating a winding of the inductor or transformer
with a reactive component in a series resonant circuit with the winding
to cancel a portion of a reactive voltage of a magnetizing inductance of
the inductor or transformer, a value of the reactive component being
chosen such that a phase difference between a measured voltage across the
resonant circuit and a measured current in series with the resonant
circuit is less than 80°, and integrating the measured voltage and
the measured current.

[0007] In accordance with another aspect of the invention, a circuit is
provided for measuring losses in an inductor or transformer having a core
comprising a sensing winding wound on the core and having one terminal of
the sensing winding connected to a terminal of the inductor, a current
sensor to measure current through a winding of the inductor or
transformer, and a reactive component connected between the terminals of
the sensing winding and the inductor or transformer and having a
reactance value to resonate with the inductor or transformer such that a
phase difference of a voltage across the sensing winding and the current
through the inductor or transformer is less than 80°.

BRIEF DESCRIPTION OF THE DRAWINGS

[0008] The foregoing and other objects, aspects and advantages will be
better understood from the following detailed description of a preferred
embodiment of the invention with reference to the drawings, in which:

[0009] FIG. 1 is a schematic diagram of a known method of measuring losses
in a passive device,

[0010]FIG. 2 is a schematic diagram of a known circuit for measuring
losses in an inductor or transformer,

[0011] FIGS. 3A and 3B are a schematic diagram of another known method and
circuit for measuring losses in and inductor or transformer,
respectively,

[0012]FIG. 4 is a graph of loss measurement error induced by 1°
phase error at different phase angles between v2 and vR in the
circuit of FIG. 3B,

[0013] FIGS. 5A and 5B are a schematic diagram and equivalent circuit,
respectively, for core loss measurement in accordance with the invention,

[0014] FIG. 6 is a graph illustrating working waveforms in the circuit of
FIG. 4A,

[0015] FIG. 7 illustrates a graphical comparison of measurements using the
methodology of the present invention and known methodologies with
datasheet information for a commercially available core material,

[0016]FIG. 8 is a graphical comparison of equivalent series resistance of
a resonant capacitor with Rcore as a function of flux density,

[0017]FIG. 9A is a schematic diagram of a circuit useful for
understanding effects of parasitic capacitance,

[0018]FIG. 9B is a graph of measurement error caused by parasitic
capacitance of a sensing winding as a function of flux density,

[0019]FIG. 10 is a schematic diagram of an embodiment of the invention
including an improvement over the circuit and method of FIG. 4A,

[0020] FIG. 11A is a schematic diagram illustrating inclusion of DC flux
bias in the circuit and methodology of FIG. 4A,

[0021] FIG. 11B is a schematic diagram of a circuit for a second
embodiment of the invention usable with arbitrary inductor or transformer
excitation,

[0022] FIGS. 12A and 12B are schematic diagrams of equivalent circuits of
the embodiment of FIG. 11B for an inductor and an air core transformer,
respectively,

[0023]FIG. 13 is a schematic diagram of a complete circuit for the
embodiment of FIG. 11B,

[0024] FIGS. 14A and 14B are graphical depictions of measurement waveforms
and variation of core loss density with flux density.

[0025] FIGS. 15A and 15B are graphical depictions of measurement waveforms
and variation of core loss density with flux density for different
excitation waveforms,

[0026]FIG. 16 is a schematic diagram of the core loss measurement circuit
of FIG. 13 including DC flux bias,

[0027] FIGS. 17 and 18 are simplified depictions of an adjustable parallel
plate capacitor and an adjustable transformer, respectively,

[0028]FIG. 19 is a schematic diagram of a an exemplary inductor loss test
circuit in accordance with the invention, and

[0029] FIG. 20 is a schematic diagram of a an exemplary inductor loss test
circuit including DC flux bias in accordance with the invention

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT OF THE INVENTION

[0030] Referring now to the drawings and more particularly to FIGS. 1-3,
several known measurement methods for obtaining core loss information are
illustrated and will be discussed in turn. It is to be understood that,
while FIGS. 1-3 do not illustrate the invention and are directed to known
measurement methods, the illustrations of FIGS. 1-3 are also arranged to
facilitate conveyance of an understanding of the problems addressed and
overcome by the invention as well as allowing a more complete
appreciation of the meritorious effects of the invention. Therefore, no
portion of any of FIGS. 1-3 is admitted to be prior art in regard to the
present invention.

[0031] One universally applicable and frequency-independent but relatively
crude and indirect known approach to loss measurement is through a
thermal approach in which the basic method is to put the device under
test (DUT) in a thermally isolated chamber through which a fluid is
circulated and to measure the temperature difference of the fluid at the
inlet and outlet of the thermally isolated chamber and the flow rate of
the fluid. The test arrangement is schematically illustrated in FIG. 1.
The heat generated by the DUT can then be calculated from the temperature
difference, flow rate and specific heat of the fluid. However, since this
methodology develops a measurement of total heat generated, it is very
difficult to isolate or exclude resistive winding losses of the inductor
or transformer when the core losses are principally of interest. The
process is also time-consuming since thermal equilibrium or at least
stable, steady-state conditions must be achieved within the thermally
isolated chamber.

[0032] Another loss measurement technique which is applicable to high
frequency inductor or transformer excitation, although also indirect, is
to connect a capacitor in series with the inductor core under test, as
illustrated in FIG. 2. The capacitor is then finely tuned to resonate
with the inductor at the test frequency. Because, at resonance, the
voltages on the inductor and the capacitor completely cancel each other,
the resistance in the RLC network can be easily measured. After
compensating for parasitic resistance (similar to the compensation for
winding resistance mentioned above) and resonant capacitor equivalent
series resistance (ESR), the core loss can be eventually calculated.

[0033] This method is not sensitive to phase discrepancy because phase
angle is not used in the loss calculation. However winding losses cannot
be estimated very accurately and are therefore difficult to compensate.
Further, the resonant capacitor value is critical in order to resonate at
a given frequency and the technique is applicable only to sinusoidal
waveform excitation.

[0034] A generally preferable known approach which is potentially accurate
and far faster is illustrated in FIGS. 3A and 3B. The DUT carries two
windings: an excitation winding and a sensing winding. The current in the
excitation winding and the resulting voltage on the sensing winding are
measured with an oscilloscope. Integrating the voltage and current
waveforms will provide the loss consumed by the core under test. In
principle, this method does not present significant drawbacks and, in
addition to requiring only a relatively straightforward computation, this
technique excludes winding losses; a principal advantage in comparison
with the techniques discussed above.

[0035] However, in practice, this technique is highly sensitive to phase
discrepancies between the measured voltage and current waveforms.
Unfortunately, phase discrepancies sufficient to cause significant
measurement inaccuracy may easily arise from the measurement
instrumentation, itself, as well as from slight inaccuracy in observation
of the measured waveforms.

[0036] To understand this observed high sensitivity to phase discrepancy,
consider the equivalent circuit of FIG. 3B. This method uses the
integration between the sensing winding voltage, v2, and the and the
current sensor voltage vR to obtain the core loss. Because the
impedance of the magnetizing inductor Lm is usually very much larger
than the equivalent core loss resistor, Rcore, the phase difference
between v2 and vR is very close to 90°. Assuming a
sinusoidal excitation, the sensitivity to phase discrepancy is given by:

Δ=tan(φv-i)×Δφ

where φv-i is the angle between the two terms (voltage and
current) for integration. Thus when the voltage and current have a phase
difference near 90°, a small phase discrepancy Δφ will
be greatly amplified by tan(φv-i) and it can be readily
appreciated that a small phase discrepancy can lead to a significant loss
error after the integration. For low frequency measurement, the parasitic
effect in the test circuit is small and the phase discrepancy is
sufficiently small to avoid significant measurement error. However, as
frequency increases, small electrical effects such as the parasitic
effects of a non-ideal current sensing resistor, mismatch between
measurement probes and oscilloscope sampling rate limitation may become
significant. Phase discrepancy generally derives principally from the
parasitic inductance and capacitance of the current sensing resistor and
mismatch between measurement probes and the sampling resolution of the
oscilloscope. For example, a 2Ω current sensing resistor having 1 1
nH parasitic inductance will produce a 0.9° phase discrepancy at 5
MHZ. Similarly, for a 5 GS/s digital oscilloscope, the minimum sampling
period is 200 picoseconds which is 0.36° at 5 MHZ. FIG. 4
graphically illustrates the loss error induced by a 1° phase
discrepancy (less than the sum of the two exemplary phase discrepancies
noted in the previous two sentences) at different phase angles between
v2 and vR. It can be readily seen that the loss error
approaches 100% as the phase shift approaches 90° and that the
percentage error in measured loss varies by nearly four orders of
magnitude from near 0° to near 90° phase difference between
voltage and current. Since these phase discrepancies are inevitable, this
classic technique is not a suitable method by which to accurately test
for magnetic losses at high frequencies.

[0037] Referring now to FIGS. 5A and 5B, the measurement method and
apparatus in accordance with the invention will now be discussed. As with
the classic method discussed above, the method in accordance with the
invention also uses a core having an excitation winding and a sensing
winding. However, one terminal of the sensing winding is connected to the
same polarity terminal of the excitation winding and that node is also
connected to a resonant capacitor Cr. This node connecting the
resonant capacitor and the excitation and sensing windings provides an
additional voltage, v3, to be monitored in addition to v2 and
vR. In such a circuit configuration, v3 is virtually the sum of
the sensing winding voltage and the resonant capacitor voltage.

[0038] The purpose of adding the resonant capacitor Cr is to cancel
the voltage on the magnetizing inductor Lm. If Cr and Lm
are resonant at the input frequency, their voltages are the same in
magnitude but 180° out of phase. As a result, v3 equals the
voltage on the equivalent core loss resistor Rcore, and it is in
phase with the current through excitation winding iR (also in phase
with current sensor voltage vR). Integration of v3*vR will
produce the loss consumed on Rcore because Lm and Cr don't
consume real power. As v3 and vR are in phase, the integration
of their product is not sensitive to phase discrepancy.

[0039] Although Cr is used to cancel the voltage on the inductor, it
is not necessary to completely cancel that voltage. FIG. 2 shows that
when the phase angle is 30° the power error is only 1%, which is
small enough to facilitate improvement in inductor design. (From FIG. 4,
even a phase difference of 80° would limit loss measurement error
for a 1.0° phase discrepancy to under 10% which may be adequate to
support inductor design improvement.) That means the value of the
resonant capacitor Cr could be in a relatively wide range, and does
not have to resonate with Lm at the input frequency. It only needs
to keep the phase angle between v3 and vR far from 90°.

[0040] The phase angle between vR and v3 is described in
equation (1).

Φ v 3 - i R = tan - 1 ( ω L m -
1 ω C r R core ) ( 1 ) ##EQU00001##

By properly choosing the resonant capacitor Cr, the phase angle
between v3 and vR can be moved far away from 90°.

[0041] If the turns ratio is 1:1, the formulas for the flux density and
core loss are

where N2 is the sensing winding turns number, T is the period of the
sinusoidal waveform and Rref is the current sensing resistance. If
the core under test is a thin toroid core, it can be assumed the flux
density that flows in the core is almost uniform. Dividing the core loss
by the equivalent volume Ve, the core loss density at the specific
flux density can be obtained. Thus the core loss density is

P v = 1 TR ref V e ∫ 0 T v 3 v r
t ( 4 ) ##EQU00003##

[0042] It should be noted is that v3 can not be obtained by measuring
v2 and vc separately and then summing them. Because the value
of v3 is much smaller than v2 and vc, a small error
percentage of v2 or v, will produce significant error in v3. So
the cancelled voltage v3 should be measured by a single probe. For
the turns ratio, it could be a different value other than 1:1. However,
it is preferred to choose 1:1 symmetric windings, because this will
automatically guarantee the effective turns ratio to be 1:1, regardless
of the leakage inductance. In addition, a 1:1 turns ratio allows a
bifilar winding to be used which will reduce the parasitic capacitances
of the transformer. For the current sensing resistor, to reduce the phase
discrepancy caused by its equivalent series inductance (ESL), multiple
resistors can be connected in parallel. However, too many resistors in
parallel will cause additional leading parasitic inductance, so only a
few current sensing resistors in parallel are preferred. For the probes
used in the test circuit, they should be the same model so that their
time delays are identical. To further check that the probes match well,
They can be used to test a square wave simultaneously and check if their
rising edges are very close to each other on the oscilloscope. In
addition, v3 and vR's probes should be exchanged during test.
If two groups of data are in adequate agreement, it can be assumed that
their time delay difference will not yield significant error. For the
resonant capacitor, a high quality factor capacitor is preferred. The
loss on the resonant capacitor is included in the calculated loss.
Therefore high Q capacitors like mica or porcelain capacitors are
preferred. During the test, the magnetizing inductance tends to change
due to environment change, such as temperature, DC flux, etc. Therefore
the resonant capacitor should be tuned to keep the phase angle between
v3 and vR far away from 90°.

[0043] To verify the efficacy and accuracy of the above-described method,
the core loss density of the commercial magnetic material 4F1 (NiZn
Ferrite) from Ferroxcube has been tested and compared with its datasheet
value. The test arrangement used a toroidal core of 4F1 material with an
outer diameter of 21 mm, and inner diameter of 20 mm and a thickness of 3
mm sample wound as a transformer and the excitation winding driven by a
commercially available power amplifier (Amplifier Research 25C250A) which
was, in turn, driven by a 10 MHZ sine wave from a commercially available
function generator. The core under test was immersed in thermal oil
heated by a hot plate and its temperature controlled by a thermal coupler
to be 100° C. The waveforms monitored by oscilloscope are shown in
FIG. 6.

[0044] From FIG. 6, it is seen that v2 leads vR by 4.2°
which is far away from 90° and integration of those waveforms will
thus have much reduced phase discrepancy induced error. When the results
of integration and power loss density is determined, the measured power
loss density is in extremely good agreement with the data sheet
information for the commercially available core being tested as
graphically shown in FIG. 7 and in far better agreement with datasheet
information (which is believed to have been determined by the thermal
method alluded to above) than the conventional measurement method
discussed above.

[0045] To summarize, the sources of error in the above measurement method
in accordance with the invention are phase discrepancy, error caused by
the equivalent series resistance (ESR) of the resonant capacitor and
error due to the parasitic capacitance of the sensing winding of the
transformer. As stated above, the phase error is greatly reduced to below
5% if the phase angle is properly adjusted. From FIG. 2, it is seen that
if phase discrepancy is below 1°, a 30° phase angle between
v3 and vR is enough to guarantee that the error in measured
loss is less than 1%. The measured power loss can be converted into the
equivalent core loss resistance Rcore shown in FIG. 5B. From FIG. 8,
it can be seen that Rcore is around 1Ω, and varies with flux
density, from 0.8Ω to 1.6Ω. (The value of Rcore is
converted from the measured power loss, since the power loss and current
are already known. The capacitor ESR is a measured value. Comparison
between these two resistor values is straightforward to estimate the
error.) To obtain the data illustrated in FIG. 8, two 68 pF mica
capacitors (which have very low ESR) were connected in parallel to
resonate with the magnetizing inductor. The total measured ESR using this
arrangement is about 50 mΩ. This value can then be compared with
the measured Rcore in FIG. 8. Therefore, the error in measured loss
due to the ESR of the resonant capacitor is between 3.3% and 6.8%, which
is adequately small even though much larger than the loss measurement
error due to phase discrepancy. However, if the ESR of the resonant
capacitor can be accurately measured or approximated by substitution of
resonant capacitors of different constructions, this contribution to
total loss error measurement can be compensated.

[0046] In regard to the error contribution attributable to parasitic
capacitance of the sensor winding, because the probe's impedance is not
infinite and the transformer has winding parasitic capacitance, there
will be a small current flowing through the sensing winding. Usually the
probe's input capacitance is larger than transformer winding capacitance.
The equivalent circuit including the parasitic capacitor is shown in FIG.
9A. Although v3 and vR probes also have parasitic capacitance,
their loading effect is much smaller than the v2 probe. Because the
magnitude of v3 and vR is much smaller than v2, the
currents flowing through the input capacitances of the probes are much
smaller than that of the v2 probe. Therefore, analyzing the loading
effect of the v2 probe, it can be assumed that the other two probes
are open-circuited.

[0047] In FIG. 9A, Cp is the total parasitic capacitor in parallel,
including the probe's input capacitance and the winding parasitic
capacitance. Because of this capacitance, the assumption that secondary
winding is open fails at sufficiently high frequency. As a result, a
small current flows through Cp. This current will induce a voltage
drop on the sensing winding's leakage inductor L12 and winding
resistor R2. Therefore v2 will have a discrepancy with the
voltage on Lm and Rcore, which is vm. When frequency is
high, this discrepancy cannot be ignored. Furthermore, current measured
by current sensor iR is not equal to the current flow through
magnetizing inductor im.

[0048] To quantify the error caused by the parasitic capacitor, it is
assumed that

The derivation of this approximation equation is unimportant to an
understanding of the invention or its practice, particularly since
simulation proves that this approximation equation can adequately
estimate the error caused by Cp, if the assumptions in equations
(5), (6) and (7) are true.

[0049] For the above 4F1 core measurement example, the following
parameters can be estimated: Cp≈10 pF, Lm=2 μH,
R2=0.2, ω=2π×10 MHz. With the value of Rcore in
FIG. 8, the error percentage for different flux density can be
calculated. Details of this calculation are not important to an
understanding or the successful practice of the invention. The result of
the calculation is graphically illustrated in FIG. 9B.

[0050] Because of the ω2 term in equation (8), the loss error
due to parasitic capacitance of the sensing winding will increase rapidly
with frequency. To reduce this error, sensing winding resistor R2
should be small, the ratio of Lm to Rcore should be small (e.g.
the magnetizing inductor quality factor is low as is usually determined
by material), and parasitic capacitance Cp should be small.
Therefore the parasitic capacitance of the transformer should be well
controlled, and a low input capacitance probe is preferred. Also, to
reduce the loading effect of the probe, an alternative embodiment of the
invention is shown in FIG. 10.

[0051] The circuit of FIG. 10 measures the voltage on the resonant
capacitor v2 rather than measuring the voltage on the sensing
winding. By making this alternative measurement, the current through the
sensing winding is much reduced and Cprobe is merged into the
resonant capacitor Cr. Voltage on the sensing winding can be
calculated by simply subtracting v2 from v3.

[0052] Core loss varies with DC flux bias. Even though DC flux doesn't
generate loss directly, it will determine the working point of the flux
swing. It is therefore necessary to investigate the AC loss under certain
DC flux bias. In the circuit of FIG. 5A and the alternative measurement
method described above (FIG. 10), DC flux bias can be conveniently added
by adding a third winding and injecting DC current through it. There are
two preferred arrangements for doing so. The circuit of FIG. 11A uses one
core material sample and adds the DC current through a third winding. The
arrangement of FIG. 11B uses two core material samples and the DC
windings of the two samples are opposite in polarity. RF chokes are used
to attenuate high frequency current induced in the DC winding. The
purpose of the preferred arrangement of FIG. 11B is to minimize the AC
voltage seen by the RF choke inductors and DC current source so that the
AC current through the DC winding is minimized.

[0053] The principle of operation of the circuits discussed above in
connection with FIGS. 5A, 10, 11A and 11B borrows the concept of
resonance using Cr as the resonant capacitor, to cancel the reactive
voltage on the magnetizing inductor so that the phase difference between
v3 and vR may be set close to 0° (rather than near
90°) and the integration of their product will be far less
sensitive to phase discrepancy. Although this method reduces the phase
sensitivity of the measurement error, it is limited to sinusoidal
excitation because the capacitor voltage can only cancel the inductor
voltage at a single frequency. To cancel the reactive voltage on the
magnetizing inductor for the entire frequency range, the capacitor is
replaced with an inductor and the polarity of the transformer is changed,
as shown in the alternative embodiment of the invention schematically
represented in FIG. 12A. In this embodiment, v3 is the total voltage
of the inductor voltage vL and -vm. That is,

v3=(sL-sLm-Rcore)im,

[0054] If L=Lm, v3 will be the voltage on the equivalent core
loss resistor and in phase with the vR. Integration of the product
of v3 and vR will be the loss on Rcore, and even less
sensitive to phase discrepancy. However, the winding loss on the added
inductor will affect the accuracy of the measured result. To avoid
introducing an additional source of measurement error, the circuit of
FIG. 12B is preferred. The circuit of FIG. 12B uses the secondary side
voltage of an air core transformer to cancel the reactive voltage on the
magnetizing inductor. Because the air core does not exhibit core loss,
its magnetizing inductor can be used to be an effectively ideal inductor
for inductor L in FIG. 12A.

[0055] The complete circuit of the preferred embodiment of the invention
is shown in FIG. 13. Other than the core-under-test/DUT and the air core
transformer, another transformer is used as an impedance transformer to
reduce the voltage drop on the internal impedance of the power amplifier
alluded to above for driving the core-under-test/DUT. Using this circuit,
The core loss can be calculated as

P core = 1 T R ref ∫ 0 T v 3 v r
t ( 10 ) ##EQU00006##

where T is the excitation period and Rref is the current sensing
resistance.

[0056] The principle of operation of the preferred measurement circuit is
that most reactive power in the core-under-test is cancelled by the
reactive power in the air core, leaving only resistive power consumed by
the core-under-test. If the power is more fully resistive, it will be
much easier to be measured. From another perspective, the circuit of FIG.
12B or 13 can be interpreted as comparing the core-under-test with the
reference core, which is air core which does not exhibit core losses. The
difference in the electrical characteristics of the air core transformer
and the core-under-test is then the core loss of the core-under-test.

[0057] In some situations, the air core may not be a good choice for the
reference core, due to its larger parasitics. Nevertheless, when the
inductance value of the core-under-test is high, a large air core
transformer is needed and will introduce non-negligible parasitic
capacitances. In such a case, a core of (known) low loss could be used as
the reference core, and its size and parasitic capacitance can be much
smaller than an air core transformer. If its loss is much smaller than
the core-under-test, the error can be tolerable and/or suitable
compensation made.

[0058] Two experiments have been performed to demonstrate the concept of
this embodiment of the invention. The first uses NiZn ferrite toroid core
(4F1 from Ferroxcube) as the core under test. The excitation signal is
generated by AFG3102 function generator (from Tektronix, and amplified by
25A250A power amplifier (from Amplifier Research. An air core is used as
the reference core, and an air core transformer is used to cancel the
voltage on the magnetizing inductor of the core-under-test. The voltage
on the core-under-test is a 1 MHz or 2 MHz square wave. The measurement
waveform and core loss density is shown in FIGS. 14A and 14B. The second
experiment is a test of sintered low-temperature co-fired ceramic (LTCC)
ferrite material (40011 from ESL ElectroScience). The reference core
material is 4F1, and its core loss is controlled much smaller than the
loss in LTCC core. The test waveforms and core loss density for
sinusoidal and triangular waveforms is shown in FIG. 15.

[0059] To measure the core loss under DC flux bias, it is convenient to
add the third winding to inject DC current as described above in regard
to FIGS. 11A and 11B. Another method to add DC flux is shown in FIG. 16
which directly adds a DC voltage source in the primary winding loop. Thus
there will be a DC current flow through the primary winding and the core
will then be working under a DC flux bias condition corresponding to the
DC component of thee primary loop current. This technique is also
applicable to the embodiments of the invention discussed above.

[0060] When changing the working condition of the core-under-test, the
permeability will change. As a result, the resonant capacitor Cr in
sinusoidal excitation embodiments or the transformer L in arbitrary
waveform excitation embodiments should be tuned in order to properly
cancel the reactive voltage on the magnetizing inductance of the
core-under-test. Tuning the capacitor or transformer is difficult if
discrete components are used since substitution of fixed capacitors or
transformers must be repeatedly mounted and demounted in the test
circuit. Therefore, it is preferred for the capacitor value or the
transformer's magnetizing inductance value to be made variable and
conveniently controlled (or at least easily trimmed with a series of
discrete capacitors or inductors. FIGS. 17 and 18 illustrate two examples
for controlling the value of a capacitor or magnetizing inductance of a
transformer, respectively although numerous other structures for
providing variable capacitance or inductance are known and are suitable
for the practice of the invention. Specifically, FIG. 17 shows the
structure of the tunable parallel plate capacitor. By mechanically
adjusting the distance between the parallel plates, the capacitor value
can be finely tuned. FIG. 18 shows an exemplary structure for a tunable
transformer using two core pieces, each wound with a winding. By
mechanically adjusting the air gap between the two pieces, the inductance
value of the transformer can be conveniently tuned. With these tunable
components, the core loss can be conveniently measured under different
conditions.

[0061] In the interest of completeness, it should be kept in mind that
core loss is one part of the inductor loss. In many practical
applications, designers want to know the total loss (or equivalent
resistance) of the inductor or transformer under practical waveform
conditions, like rectangular voltage waveform and DC current bias.
However, at high frequency or where high frequency components are present
in the excitation waveform, the conventional method still face the same
problems as described above. To avoid those problems, the method and
apparatus of the invention as discussed above can be slightly modified to
satisfy measurement of total inductor loss with good accuracy. An
embodiment of the invention for measurement of total inductor losses is
shown in FIG. 19. Since core loss need not be isolated in this case, the
transformer of the core-under-test in FIG. 13 is replaced by an inductor
wound on the core-under-test. Thus the measured loss is the total loss
consumed on the inductor, including winding loss and core loss.
Similarly, replacing the transformer in FIG. 16 with an inductor wound on
the core-under-test as shown in FIG. 20, the circuit can test the
inductor loss with DC bias. Of course, winding losses can also be
isolated by additionally measuring core losses as discussed above and
subtracting measured core losses from measured total inductor losses;
either or both of such measurements being performed in accordance with
the invention as described above.

[0062] In view of the foregoing, it is clearly seen that the invention
provides for sufficiently accurate measurement of core losses, winding
losses and/or total inductor or transformer losses, even under different
operating conditions sufficient to cause significant change in core
permeability, to support advances in inductor or transformer core design.
Errors in measurement at high excitation frequencies or high frequency
components of excitation waveforms are minimized or avoided and the
magnitude of such measurement error, with or without DC flux bias, can be
quantitatively evaluated.

[0063] While the invention has been described in terms of a several
embodiment that may be preferred for different applications and
measurement and excitation conditions, those skilled in the art will
recognize that the invention can be practiced with modification within
the spirit and scope of the appended claims.

Patent applications by Fred C. Lee, Blacksburg, VA US

Patent applications in class Hysteresis or eddy current loss testing

Patent applications in all subclasses Hysteresis or eddy current loss testing