Sign up to receive free email alerts when patent applications with chosen keywords are publishedSIGN UP

Abstract:

A quadratic amplitude matching system and associated method with an
associated tuning control system is provided for continuously and
automatically tuning a quadratic amplitude matching filter (QAMF) to a
band center of an interfering signal to provide improved rejection of an
interfering signal coupled from a transmission antenna into a local
receive antenna in the presence of local multi-path, thereby providing
improved interference cancellation system performance. The matching
control system is provided as an element of an interference cancellation
system.

Claims:

1. A method for continuously and automatically tuning a quadratic
amplitude matching filter (QAMF) to an inserted signal to allow tracking
of the inserted signal to match a dynamically changing quadratic
amplitude distortion of the inserted signal for improved interference
cancellation system performance, the method comprising: a) forming an
imaginary broadband RF lobed filter having a single quiescent null within
a frequency band of interest; b) dynamically adjusting a delay time (T)
for tuning the single quiescent null of the imaginary broadband RF lobed
filter to effectively block the inserted signal to be tracked; c) forming
a first narrowband RF lobed filter with a quiescent lobe peak centered on
a quiescent null of the inserted signal to be tracked, wherein an output
of the first narrowband RF lobed filter output has a near-linear and flat
amplitude shape to match a dynamically changing quadratic amplitude
distortion of the inserted signal to be tracked; d) forming a second
narrowband RF lobed filter with a quiescent lobe peak centered on the
quiescent null of the inserted signal to be tracked, wherein an output of
the second narrowband RF lobed filter has a downward quadratic amplitude
shape to match a dynamically changing quadratic amplitude distortion of
the inserted signal to be tracked; e) forming an FIR filter with a
quiescent lobe peak centered on the quiescent null of the inserted signal
to be tracked, wherein an output of the FIR filter has an upward
quadratic amplitude shape to match a dynamically changing quadratic
amplitude distortion of the inserted signal to be tracked; f) adjusting
an in-line path delay of the first narrowband RF lobed filter, the second
narrowband RF lobed filter and the FIR filter to have the same throughput
delay; and g) adjusting combining weights of the respective filter
outputs of the first narrowband RF lobed filter, the second narrowband RF
lobed filter and the FIR filter to implement a corrective quadratic
amplitude shaping of the inserted signal, thereby matching a dynamically
changing quadratic amplitude distortion of the inserted signal to be
tracked.

2. The method of claim 1, wherein the respective outputs of the first and
second narrowband RF lobed filters and the FIR filter are dependent upon
the dynamically adjusted delay time (T) for tuning the single quiescent
null of the imaginary broadband RF lobed filter.

3. The method of claim 1, wherein the first and second narrowband RF
lobed filters and the FIR filter are orthogonal to the imaginary
broadband RF lobed filter in a quiescent state.

4. The method of claim 1, wherein said step (g) of adjusting combining
weights of respective filter outputs of the first narrowband RF lobed
filter, the second narrowband RF lobed filter and the FIR filter parallel
filters is performed under control of an external amplitude control
signal.

5. The method of claim 1, wherein the first and second narrowband RF
lobed filters and the FIR filter are comprised of a plurality of lobes
formed within the frequency span of a single lobe of the imaginary
broadband RF lobed tuning filter.

6. The method of claim 5, wherein one of the plurality of lobes of the
narrowband RF lobed filter is peaked on the inserted signal to be
tracked.

7. The method of claim 6, wherein said one of said plurality of lobes
peaked on the inserted signal to be tracked is a quiescent lobe of an
amplitude sloped matching filter (ASMF) function.

8. The method of claim 6, wherein said one of said plurality of lobes is
peaked on the interfering signal to be tracked and is centered at a null
of the broadband RF lobed filter.

9. The method of claim 5, wherein one of the plurality of lobes of the
imaginary broadband RF lobed tuning filter is centered on the inserted
signal to be tracked has an amplitude shape to correct a dynamically
changing quadratic amplitude distortion of the inserted signal to be
tracked.

10. The method according to claim 1, wherein the broadband RF lobed
filter has a null to null bandwidth substantially twice the desired
tuning bandwidth.

11. The method according to claim 1, wherein said step (b) of dynamically
adjusting a delay time is performed via a control process.

12. The method according to claim 11, wherein the control process
comprises: a) measuring an RF energy output from the broadband RF filter;
b) applying a delta increment to the delay time (T); c) re-measuring the
RF energy output from the broadband RF filter; d) comparing the measured
and the re-measured values of RF energy; and e) determining if the
applied delta increment to the delay time (T) improves the tuning as
measured by an energy output from the broadband RF filter; f) if
improved, updating the delay time (T) by adding a positive delta
increment as T=T+ΔT; g) if unimproved, updating the delta delay
time (ΔT) by changing sign of delta delay time increment as
ΔT=-.DELTA.T.

13. The method of claim 1, wherein increasing the control voltage causes
the delay T to be increased resulting in a narrowing of the filter lobes.

14. The method of claim 13, wherein narrowing the filter lobes results in
a downward frequency shift of the individual lobes and nulls of the
filter.

15. The method of claim 1, wherein decreasing the control voltage causes
the delay T to be decreased resulting in a widening of the filter lobes.

16. The method of claim 15, wherein widening the filter lobes results in
an upward frequency shift of the individual lobes and nulls of the
filter.

17. An interference cancellation system, comprising: (A) an adaptively
tuned quadratic control (ATQC) module (200) for providing a tuning
procedure directed to a quadratic amplitude matching filter (QAMF) (100)
control function to the band of interference and for externally
controlling the QAMF control function subsequent to said tuning, said
adaptively tuned quadratic control (ATQC) module (200) comprising: a) an
inline variable lobe filter structure (VLFS) (201) for providing
controlled variable time delay for generating a broadband RF tuning
filter formed by a delay T and a first narrowband inline control filter
for providing a near-flat quadratic control filter lobe formed by a first
delay (T) and a second delay (2nT) yielding a total time delay of (2n+1)T
centered in the null of the broadband RF tuning filter and weighted by an
external amplitude control signal; b) a second narrowband inline control
filter as a slaved lobe filter structure-down (104) for providing a down
quadratic amplitude control filter lobe formed by a delay of (2m+1)T
centered in the null of the RF tuning filter and weighted by an external
amplitude control signal; c) a third narrowband inline control filter as
a slaved lobe filter structure-up (105) for providing a up quadratic
amplitude control filter lobe formed by a simple FIR filter with
inter-tap spacing delay of (2o+1)T centered in the null of the tuning
filter and weighted by an external amplitude control signal; d) an
offline time delay tuning control (TDTC) element (202) for receiving
signal samples output from the inline variable lobe filter structure
(VLFS) 201 to control a first variable time delay element (T) 205 of the
inline variable lobe filter structure (VLFS) 201 to provide said
controlled variable time delay to form said broadband RF tuning filter
and a second variable time delay element (2nT) (208) of the inline
variable lobe filter structure (VLFS) 201 to provide said controlled
variable time delay to generate said slope control filter lobe. (B) an
adaptive control loop (6) for adjusting a complex weighting of the
delayed coupled signal (57) to maximally cancel a propagated reference
signal.

18. The interference cancellation system of claim 17, wherein the inline
variable lobe filter structure (VLFS) 201 comprises: a) said first
variable time delay line (T) (205) for providing broadband tuning of an
imaginary tuning filter lobe; and b) a second variable time delay line
(2nT) (208) for providing more narrowband tuning of the imaginary tuning
filter lobe relative to said first delay element; wherein said first
variable time delay element (205) and second variable time delay element
(2nT) (208) yield a total time delay of (2n+1)T centered in the null of
the tuning filter and skewed by an external slope control signal.

19. The interference cancellation system of claim 17, wherein the second
variable time delay line (2nT) 208 is an integer multiple of said first
time delay line (T) 205.

20. The interference cancellation system of claim 17, wherein the
adaptive control loop (6) comprises: a reference port (9) for receiving
the an antenna signal (30); an auxiliary port (8) for receiving a delayed
and matched coupled signal (57); a complex correlator (66) for generating
error correlation signal (72) an integrator (67) to smooth transients on
the error correlation signal (72) to form the adaptive weight control
signals(73); a complex phase and amplitude weighting device (68) having a
first input and a second input, said first input receiving said delayed
and matched coupled signal (57), said second input receiving a complex
adaptive weight control signal(73)to weight the delayed and matched
coupled signal (57) to produce a weighted delayed coupled signal (65);
and forming a weighted delayed coupled signal (65) as a mirror image of a
propagated reference signal, contained in antenna signal 30; a summing
junction (70) having a first and second input, said first input for
receiving said weighted delayed coupled signal (65) output from said
complex phase and amplitude weighting device (68), said second input for
receiving the antenna reference signal (71) to yield a protected output
signal (58).

21. The interference cancellation system of claim 17 wherein said
propagated reference signal comprises at least a transmission signal (40)
propagated through an uncontrolled path from a first antenna (2) and
received at a second antenna (4).

22. The interference cancellation system of claim 20, wherein forming the
weighted delayed coupled signal (65) as a mirror image of the transmitted
reference signal indicates that it is equal in amplitude and 180.degree.
out of phase with a portion of the transmitted signal (40) in the antenna
reference signal (71).

23. The interference cancellation system of claim 17, wherein said
antenna signal (30) includes both said propagated reference signal and at
least one other signal.

24. The interference cancellation system of claim 23, wherein the at
least one other signal is a desired signal anticipated by a protected
receiver (25).

25. A method for implementing a first-order quadratic correction to the
amplitude of an input signal across its band by the use of a quadratic
amplitude matching filter (QAMF), the method comprising: a) dividing an
input signal into three parallel signal paths; b) dynamically adjusting a
delay time (T) in a first signal path from among said three parallel
signal paths for tuning a first narrowband RF lobed filter with one of
its quiescent lobes peaked on an interfering signal to be tracked; c)
forming a second more narrowband RF lobed filter in a second signal path
from among said three signal paths dependent upon the delay time (T)
wherein one of the quiescent lobes of the second narrowband RF filter is
peaked on the interfering signal to be tracked; d) forming a simple FIR
filter in the third branch dependent upon the delay time, (T), having a
filter configuration in the form of an upward quadratic shape centered on
the interfering signal to be tracked, e) matching each of the first,
second and third signal paths to have a uniform path delay dependent upon
the delay time (T); f) weighting each of the first, second and third
signal paths according to an external control function, and g) combining
the first, second and third signal paths into a single output to allow
facilitate the implementation of a first-order quadratic amplitude
distortion of the input signal via the QAMF to match the delayed coupled
signal to that of the propagation path for improved interference
cancellation of the inserted signal in an interference cancellation
system.

26. The method of claim 25, wherein the first narrowband RF lobed filter
is sufficiently broad enough in bandwidth to implement a first path
characterized by a near-linear and flat amplitude shape.

27. The method of claim 25, wherein the second more narrowband RF lobed
filter is controlled to be more narrow than the first narrowband RF lobed
filter to implement said second signal path for downward quadratic
amplitude adjustment of the inserted signal.

28. The method of claim 25, wherein the FIR filter upward quadratic area
is centered on the interfering signal to be tracked to implement said
third signal path for upward quadratic amplitude adjustment of the
inserted signal.

Description:

FIELD OF THE INVENTION

[0001] The invention relates to the field of radio communication and, in
particular, to the reduction of interference in signals coupled from a
transmission antenna into a local receive antenna in the presence of a
local multipath.

DESCRIPTION OF THE RELATED ART

[0002] Unwanted (i.e., interfering) signals manifest themselves in several
ways. Interference can cause a reduction in the sensitivity of a receiver
(receiver desensitization), masking of a desired signal, tracking of an
undesired interfering signal and loss of the desired signal, and
processing of the unwanted interfering signal instead of the desired
signal. Each of these manifestations of interference limits the
communication capabilities of the radio system afflicted by this problem.
The effects of interference can be some combination of the absence of
usable output from a receiver, false signals from a receiver, and
malfunction of a device which is operated by the receiver. During
emergency situations, the loss and corruption of the desired signal can
be critical.

[0003] Unwanted signal interference is generally caused by modulation of
signals provided to the receiver by the carrier waves, or by the wideband
noise, generated by collocated transmitters. Unwanted signal interference
also occurs when frequency-hopping transmitters are transmitting signals
at frequencies that are substantially close to the frequency of the
desired receiver signal (i.e., co-channel operation). Unwanted signal
interference can also be caused by "pseudo white-noise" generated by
transmitters over a wide band of frequencies on either side of the
transmitter's operating frequency. It is often found in collocated
transceiver systems that this "pseudo white-noise" reaches unacceptable
levels within the operating band of adjacent receivers. Unwanted signal
interference is also attributed to signals (i.e., spurious emissions)
generated by transmitters at odd harmonics of the fundamental frequency
of the transmitter output signal. This is caused by the non-linear
transfer characteristics of amplifiers in the transmitter chain, or by
passive inter-modulation at the transmitter or receiver antenna
connectors.

[0004] In order to substantially reduce and eliminate the undesired
interfering signals while maintaining the spatial benefits afforded by
proximately locating transceivers, especially frequency-hopping
transceivers, several signal processing techniques have been proposed.
These techniques include agile filtering, agile filtering with multi
coupling and interference cancellation.

[0005] When the signal noise and spurious sidebands generated by the
interfering transmitter are strong, broadband, and scenario dependant,
standard interference cancellation is inadequate. Changes in the scenario
surrounding the platform may vary the coupling between the transmitter
and the protected receiver and thus require adjustment of system
parameters in an adaptive process.

[0006] Interference cancellation involves sampling the transmitter output
signal in order to eliminate from the received signal, any interfering
signal having a frequency proximate to the receiver carrier frequency. In
co-site environments, a collocated source usually interferes with the
receiver due to the finite isolation between transmit and receive
antennas. This interference in a co-site environment is a combination of
several factors, desensitization caused by one or more nearby high-power
transmitter carriers and wideband moderate to low-power interference
components associated with those carriers. These interference components
are received by the collocated radio and degrade system operation. The
nearby high-power transmitter carrier signals could simply exist as a
part of the platform signal environment. Further, the interfering signals
may be classified as either cosite or remote interferers. A cosite
interferer is physically collocated with the receiver on a platform
permitting a physical circuit connection from the interference generator
to the receiver. A remote interferer is located far enough from the
receiver to preclude a physical circuit connection.

[0007] A typical Interference cancellation system utilizes a
correlation-based adaptive controller using feedback derived after the
cancellation process. The system takes a sample of an interference signal
and adjusts the magnitude and phase such that the result is equal in
amplitude and 180° out of phase with the interference signal at
the input of the receiver. The vector sum of the two signals will cancel,
leaving only the signal of interest. In practice, however, the two
signals are not identical, due to unwanted distortion in the reference
path, as well as differences in signal path lengths and non-ideal
components in the Tx/Rx signal paths. Cancellation performance is a
function of amplitude and phase match between the interference signal and
the sampled signal. Transmission path distortions include time delay,
magnitude amplitude and phase distortion, linear amplitude and phase
distortion, and quadratic distortion, correction of each adding a level
of performance enhancement but also adding to system complexity and
difficulty in implementing the corrections.

[0008] To suppress a wideband interference signal, the performance of a
cancellation system is directly proportional to the match between the
sampled transmission cancellation signal and the receive path
interference signal across the signal bandwidth. The interference
cancellation system (ICS) compensates for minor corrections and component
drift by controlling a complex weight that implements flat phase and
amplitude controls in the adaptive control loop (ACL) to correct the
magnitude amplitude and phase errors between the two. The receive path
interference signal provided to the ICS is disrupted by signal
distortions in time of arrival, linear and non-linear (i.e., quadratic)
amplitude, and linear and non-linear phase. The sampled transmission
cancellation signal must be adjusted to match this distorted receive path
signal as closely as possible to achieve complete nulling of the received
interference signal. The present disclosure addresses these concerns by
focusing on minimizing mismatch errors caused by first order non-linear
amplitude distortions.

[0009] As is well known, cosite interference cancellation requires
amplitude slope matching across the signal bandwidth to achieve a deep
null across the band. U.S. Pat. No. 6,693,971 "Wideband co-site
interference reduction apparatus" (Kowalski) issued on Feb. 17, 2004 and
assigned to BAE Systems Information and Electronic Systems Integration
Inc. (Greenlawn, N.Y.), incorporated by reference herein in its entirety,
discloses a method of implementing a near-linear correction of the
amplitude slope using an amplitude slope-matching filter. However, a
drawback of the system and method of Kowalski is that it also imparts a
quadratic shape to the amplitude across the band.

[0010] Similarly, the propagation path can also impart a quadratic
amplitude modulation across the band that will be time varying with the
changing environment. Together, these two distortions limit the nulling
performance of the cosite interference cancellation system.

[0011] A need therefore exists for a system and method for continuously
adjusting the quadratic amplitude of a coupled co-sited transmitter
signal before subtracting it from the propagated and received signal with
multipath dispersive distortions to achieve required nulling. Such a
system would also have to be tuned with the transmitter frequency and
adjust to changes in the propagation path distortion.

SUMMARY OF THE INVENTION

[0012] It is therefore an object of the present disclosure to provide a
method and apparatus for reducing the effects of interference between
collocated transceivers.

[0013] It is an object of the present disclosure to provide a method and
apparatus in which proximately located transceivers can simultaneously
transmit and receive independent signals without substantially affecting
the quality of a desired signal reception.

[0014] It is another object of the present disclosure to eliminate the
effects of interference between collocated transceivers utilizing
interference cancellation.

[0015] It is a more particular object of the present disclosure to provide
a method and apparatus for providing a quadratic amplitude matching
capability to an interference cancellation system by implementing a
quadratic amplitude matching filter (QAMF).

[0016] It is a more particular object of the present disclosure to provide
a method and apparatus for automatically tuning a bank of lobed filters
of the QAMF such that the signal tracked is near the center of each
lobing structure to generate quadratic shaping structures in the region
of a tracked signal spectrum.

[0017] It is yet another object of the present disclosure to provide a
method and apparatus for tuning this quadratic amplitude matching filter
(QAMF) over as large of a band as possible without external interface or
control.

[0018] The present disclosure provides a quadratic amplitude matching
filter (QAMF) architecture and a tuning control system as an element of
an interference cancellation system and associated method for
continuously and automatically tuning a quadratic amplitude matching
filter (QAMF) to a band center of an inserted coupled transmitted signal
for improved interference cancellation system performance and adjusting
to match propagation path distortion. More particularly, the QAMF system
provides improved rejection of an interfering signal coupled from a
transmission antenna into a local receive antenna in the presence of
local multipath.

[0019] The tuning control system and associated method of the present
disclosure provide improved signal rejection over other possible tuning
approaches by continuously tuning (adjusting) a lobed filter of the
tuning control system so that the QAMF has a quiescent flat shape in the
region of the tracked signal spectrum.

[0020] In accordance with one embodiment of the present disclosure a
tuning control system is provided for reducing interference in signals
coupled from a transmission antenna into a local receive antenna in the
presence of a local multi-path. The tuning control system interfaces with
a time-delay based lobed filter architecture including delay means for
forming synchronously locked lobed filters for both a tuning filter for
tracking to a predominant interfering signal inserted at an input port
and a bank of filters capable of applying a first order quadratic
amplitude matching to effect the amplitude shape desired for distortion
matching. The system further includes control means, associated with the
delay means, for tuning the QAMF to track the inserted signal and center
it at the center of the filter, thereby eliminating the need to interface
the control means with the transmitter.

[0021] In accordance with one embodiment of the present disclosure, a
method is provided for implementing a first-order quadratic correction to
the amplitude of an input signal across its band by the use of a
quadratic amplitude matching filter (QAMF), the method comprising:
dividing an input signal into three parallel branches, dynamically
adjusting a delay time (T) in the first branch for tuning a first
narrowband RF lobed filter with one of its quiescent lobes peaked on an
interfering signal to be tracked, wherein the first narrowband RF lobed
filter is broad enough in bandwidth to implement a first path with
near-linear and flat amplitude shape, forming a second, more narrowband
RF lobed filter in the second branch dependent upon the delay time (T)
wherein one of the quiescent lobes of the more narrowband RF filter is
peaked on the interfering signal to be tracked but controlled to be more
narrow to implement a second path for downward quadratic amplitude
adjustment of the inserted signal, forming a simple FIR filter in the
third branch dependent upon the delay time, (T), wherein its central form
is shaped to form an upward quadratic shape centered on the interfering
signal to be tracked, and wherein the FIR filter upward quadratic area is
centered on the interfering signal to be tracked to implement a third
path for upward quadratic amplitude adjustment of the inserted signal,
matching each of the first, second and third branches to have a uniform
path delay dependent upon the delay time (T) , weighting each of the
first, second and third branches according to an external control
function, and combining the first, second and third paths into a single
output to allow the QAMF to implement a first-order quadratic amplitude
distortion of the input coupled transmitted signal to match the delayed
coupled signal to that of the propagation path for improved interference
cancellation of the inserted signal in an interference cancellation
system

[0022] Also, in accordance with one embodiment of the present disclosure,
a method is provided for continuously and automatically tuning a
quadratic amplitude matching filter (QAMF) to a band center for improved
interference cancellation system performance, the method comprising: a)
forming a broadband RF lobed filter having a single quiescent null within
a frequency band of interest; b) dynamically adjusting a delay time (T)
for tuning the single quiescent null of the broadband RF lobed filter to
effectively block the inserted signal to be tracked; c) forming a first
narrowband RF lobed filter with a quiescent lobe peak centered on a
quiescent null of the inserted signal to be tracked, wherein an output of
the first narrowband RF lobed filter output has a near-linear and flat
amplitude shape to match a dynamically changing quadratic amplitude
distortion of the inserted signal to be tracked; d) forming a second
narrowband RF lobed filter with a quiescent lobe peak centered on the
quiescent null of the inserted signal to be tracked, wherein an output of
the second narrowband RF lobed filter has a downward quadratic amplitude
shape to match a dynamically changing quadratic amplitude distortion of
the inserted signal to be tracked; e) forming an FIR filter with a
quiescent lobe peak centered on the quiescent null of the inserted signal
to be tracked, wherein an output of the FIR filter has an upward
quadratic amplitude shape to match a dynamically changing quadratic
amplitude distortion of the inserted signal to be tracked; f) adjusting
an in-line path delay of the first narrowband RF lobed filter, the second
narrowband RF lobed filter and the FIR filter to have the same throughput
delay; and g) adjusting combining weights of the respective filter
outputs of the first narrowband RF lobed filter, the second narrowband RF
lobed filter and the FIR filter to implement a corrective quadratic
amplitude shaping of the inserted signal, thereby matching a dynamically
changing quadratic amplitude distortion of the inserted signal to be
tracked.

[0023] According to one aspect of the method described above, dynamic
adjustment of the time-delay element considers both direction and degree
in dependence upon the most recent nulling filter output comparison
result.

[0024] In different embodiments, the system may be implemented in discreet
components or alternatively as a MMIC. Time delays can be implemented as
either a switched delay or a continuously variable delay through an
analog control voltage.

BRIEF DESCRIPTION OF THE DRAWINGS

[0025] These and other objects, features and advantages of the invention
will be apparent from a consideration of the following Detailed
Description Of The Invention considered in conjunction with the drawing
Figures, in which:

[0026] FIG. 1 illustrates the general block diagram of an improved cosite
interference cancellation system, according to one embodiment.

[0027] FIG. 2a is a circuit diagram of the general structure of a lobed
filter for use in an improved cosite interference cancellation system,
according to one embodiment.

[0028] FIG. 2b illustrates the general structure of the lobed filter, for
use in an improved cosite interference cancellation system, according to
one embodiment.

[0029] FIGS. 3a-b are exemplary output waveforms of a lobed filter for
illustrating that subtraction, rather than summation, of the output of
two signal paths of the lobed filter forms an orthogonal filter of the
same repetitive bandwidth as the output of the lobed filter from a basic
delay (T).

[0030] FIG. 4 are exemplary resultant output waveforms of a lobed filter
formed from the summed outputs of a lobed filter having signal paths
characterized by delays which are odd integer multiples of a basic delay
(T), the resultant output waveforms illustrating that a lobe of the
summed outputs is always aligned with a lobe of a lobed filter formed
from the basic delay (T).

[0031] FIG. 5a-c illustrate three different exemplary tuning scenarios of
a generated wideband RF lobed filter that is orthogonal to the lobe of an
imaginary (unformed) wideband RF tuning filter.

[0032] FIG. 6 is a block diagram of a five-tap finite impulse response
(FIR) filter whose pass-band amplitude repeats in the frequency domain,
according to one embodiment.

[0033] FIG. 7a is a plot comprising four curves, a first curve
representing the output of a lobed broadband tuning filter, the second
curve representing a lobed narrowband tuning filter, a third curve
representing a narrower lobed narrowband filter and a fourth curve
representing an FIR filter.

[0035] FIG. 8 is a circuit diagram of a quadratic amplitude matching
filter (QAMF) structure where the tuning control comes from an external
controller, according to one embodiment.

[0036] FIG. 9 is a circuit diagram of an adaptively tuned quadratic
control (ATQC) structure where the time delay tuning control circuit has
been integrated, according to one embodiment.

[0037] FIG. 10 is a circuit diagram of the adaptively tuned quadratic
control (ATQC) of FIG. 9 incorporated into an interference cancellation
system to improve the cancellation of a local transmitter signal that is
received in the receive antenna with a time varying modulation due to
changes in local multipath.

[0038] FIG. 11 illustrates one embodiment of an improved cosite
interference cancellation system 20 for elimination of interfering
signals between three or more co-located transceivers

[0039] FIG. 12 illustrates an improved cosite interference cancellation
system for elimination of interfering signals between a single co-located
transceiver and a plurality of receivers to be protected.

DETAILED DESCRIPTION OF THE INVENTION

[0040] In the following discussion, numerous specific details are set
forth to provide a thorough understanding of the present invention.
However, those skilled in the art will appreciate that the present
invention may be practiced without such specific details. In other
instances, well-known elements have been illustrated in schematic or
block diagram form in order not to obscure the present invention in
unnecessary detail. Additionally, for the most part, details concerning
network communications, electromagnetic signaling techniques, and the
like, have been omitted inasmuch as such details are not considered
necessary to obtain a complete understanding of the present invention and
are considered to be within the understanding of persons of ordinary
skill in the relevant art.

[0041] The present description illustrates the principles of the present
disclosure. It will thus be appreciated that those skilled in the art
will be able to devise various arrangements that, although not explicitly
described or shown herein, embody the principles of the disclosure and
are included within its spirit and scope.

[0042] All examples and conditional language recited herein are intended
for pedagogical purposes to aid the reader in understanding the
principles of the disclosure and the concepts contributed by the inventor
to furthering the art, and are to be construed as being without
limitation to such specifically recited examples and conditions.

[0043] Moreover, all statements herein reciting principles, aspects, and
embodiments of the disclosure, as well as specific examples thereof, are
intended to encompass both structural and functional equivalents thereof.
Additionally, it is intended that such equivalents include both currently
known equivalents as well as equivalents developed in the future, i.e.,
any elements developed that perform the same function, regardless of
structure.

[0044] The functions of the various elements shown in the figures may be
provided through the use of dedicated hardware as well as hardware
capable of executing software in association with appropriate software.
When provided by a processor, the functions may be provided by a single
dedicated processor, by a single shared processor, or by a plurality of
individual processors, some of which may be shared. Moreover, explicit
use of the term "processor" or "controller" should not be construed to
refer exclusively to hardware capable of executing software, and may
implicitly include, without limitation, digital signal processor ("DSP")
hardware, read only memory ("ROM") for storing software, random access
memory ("RAM"), and nonvolatile storage.

[0045] Other hardware, conventional and/or custom, may also be included.
Similarly, any switches shown in the figures are conceptual only. Their
function may be carried out through the operation of program logic,
through dedicated logic, through the interaction of program control and
dedicated logic, or even manually, the particular technique being
selectable by the implementer as more specifically understood from the
context.

Overview

[0046] The present disclosure is directed to a tuning control system and
associated method for continuously and automatically tuning a quadratic
amplitude matching filter (QAMF) to a band center of a reference input
signal for improved interference cancellation system performance in a
cosite interference cancellation system. In some embodiments, the tuning
process may be performed off-line to preclude the interruption of
processing, during an operation stage, with intermediate or final control
signals being transferred to an inline structure to implement the same
control.

[0047] The present disclosure provides an automated system and method that
performs dynamic adjustment of the delay time, tuning a quadratic
amplitude matching filter (QAMF) that centers the QAMF filter on the
frequency of its reference input signal as a pre-requisite to adjusting
the QAMF filter for quadratic amplitude control (i.e., amplitude
matching). More particularly, the present disclosure provides a novel
quadratic amplitude-matching filter (QAMF) to implement dynamic,
real-time correction to the quadratic amplitude mismatch. The present
disclosure further provides a time delay tuning control 202, coupled to
the quadratic amplitude matching filter (QAMF) to provide frequency
tuning to the quadratic amplitude matching filter (QAMF) without the need
for an external tuning control signal (e.g., a tuning control signal from
the transmitter as practiced in the prior art). It should be understood,
however, that quadratic amplitude control is required, as a further
processing step beyond performing dynamic, real-time tuning. As is well
known, Quadratic amplitude control is performed to adjust the quadratic
amplitude-matching filter (QAMF) to the proper weights to match the
quadratic amplitude distortion of a sampled transmission signal to that
of the propagation path.

[0048] Referring now to the drawings, FIG. 1 is a circuit diagram for
illustrating an improved cosite cancellation circuit 20 for eliminating
interfering signals between radio transmitter 21, as an element of
transceiver 1, and receiver 25, as an element of transceiver 5, where
system dynamics cause changes in the coupling between transmit antenna 2
and receive antenna 4, co-located on a platform, according to one
embodiment.

[0049] It should be understood that each of the transceivers 1, 5 function
independent of the other such that they alternate in being viewed as
either the interfering transmitter or protected receiver depending upon
the specific needs of the user. However, the system description will only
address a single functional aspect for ease of explanation. The
transceivers 1, 5 can operate at any RF frequency including, for example,
in the high frequency (HF), very high frequency (VHF) and ultra-high
frequency (UHF) spectrums.

[0050] The improved cosite cancellation circuit 20 for the elimination of
interfering signals between radio transceivers 1, 5, is adapted to be
coupled to transceiver 5, in the illustrative embodiment, or other type
of device, known or envisioned, capable of receiving electronic signals.
The transceiver 1 operating in the transmission mode produces electronic
signals for transmission via antenna 2 of transceiver 1. Substantially
contemporaneously to this signal transmission, other electronic signals
are received by antenna 4 and provided to at least transceiver 5
operating in the receiving mode. As is known to happen, in addition to
the signals intended to be received by antenna 4, the co-located
transmitter 21 also generates noise and distortion signals which
interfere with the electronic signals received by the antenna 4 that are
to be provided to a receiver 5.

[0051] In order to substantially eliminate the effect of the interfering
signals generated from transceiver 1, the novel cancellation circuit 20
is electrically coupled to transmission signal 40. In a preferred form of
the present invention, a directional coupler 7 is operatively coupled to
the output port of transmitter 21. The cancellation circuit 20 receives a
sample of the filtered transmission signal corresponding to the
transmitter 1 to which it is coupled.

Operation

[0052] In operation, transmitter 21 transmits RF transmission signal 40
through antenna 2 which couples spatially 3 either directly or through a
multipath environment into a second antenna 4 connected to a receiver 25
on the same platform as interfering transmitter 21. This coupled energy
interferes with the reception in the receiver 25 of its desired reception
of a distant transmission. The interfering transmitter 21 thus becomes a
collocated source of interference. It is desired to protect the receiver
25 from the interfering transmitter 21. The addition of a simple
Interference Cancellation System (ICS) consisting of only a coupled
adaptive control loop (ACL) 6 can reduce this interference to a limited
extent by sampling the transmission signal 7 and feeding it into the
auxiliary port 8 of the ACL 6 while antenna signal 30, including both the
interfering propagated reference signal and the desired signal, is fed
into the reference port 9 of the ACL 6.

[0053] In an environment clear of reflective obstacles (e.g., no
multi-path sources present), the spatially coupled signal 3 from antenna
2 to antenna 4 would be received unchanged except for the propagation
delay and the quadratic amplitude matching filter (QAMF) 100 would not be
required. However, in a typical multi-path laden environment, the
spatially coupled signal 3 is distorted across the band in a number of
ways, one of them being an undesirable quadratic amplitude distortion
across the band of interest which is constant in a stable environment but
varies with a changing multipath environment of a platform in motion.

Static v. Dynamic Environments

[0054] In a static environment, the cable delay, T.sub.D277, between
sample point 7 and point 8, the input to ACL 6, is ideally adjusted to be
the typical coupling delay through space from source antenna 2 to receive
antenna 4. This delay, T.sub.D277, is implemented to include the delay of
QAMF 100 and any other in-line delays. The next level of correction is
the amplitude slope matching 278 which is ideally adjusted to match the
amplitude slope distortion through space from source antenna 2 to receive
antenna 4. These corrections will change with time in a dynamic
environment but are not the subject of this disclosure. In a dynamic
environment, as environmental conditions change with time in an
unpredictable manner, a variable quadratic amplitude distortion can be
affected upon the propagated signal resulting in an undesirable mismatch
between the coupled transmission (i.e., the signal coupled via path 7 to
8) and the propagated transmission (i.e., the signal coupled via path
2-9) limiting the effectiveness of the applied cancellation.

[0055] To correct a dynamically changing quadratic amplitude mismatch
between the afore-mentioned signals, the present disclosure provides, in
one aspect, a quadratic amplitude matching filter (QAMF) 100, generally
shown in FIG. 1 and illustrated in more detail in FIG. 7, to implement a
dynamic correction to the amplitude slope mismatch between the delayed
coupled signal 57 and antenna signal 30, including both the interfering
propagated reference signal and the desired signal.

[0056] To successfully track and match the distortion introduced by the
dispersive propagated interfering propagated reference signal, contained
in antenna signal 30, interference cancellation circuit 20 must first
continuously and automatically tune a quadratic amplitude matching filter
(QAMF) 100, to the reference input interfering transmitted signal 40 band
center. This continuous and automatic tuning process comprises a key
feature of the invention. In a preferred embodiment, the tuning process
is continuously and automatically performed by a local tuning control
system (i.e., time delay tuning control 202), as a quiescent starting
point for performing subsequent operations such as quadratic amplitude
adjustment.

[0057] It should be understood that the present disclosure is primarily
directed to: (1) an architecture that implements a quadratic amplitude
correction, (2) the tuning of the quadratic amplitude matching filter
(QAMF) 100 as a pre-requisite to performing quadratic amplitude
adjustment, and (3) quadratic amplitude adjustment under control of an
adaptive amplitude controller 225 (see FIG. 1a). It should be understood
that Adaptive amplitude control adjustment 225 uses standard control
algorithms and processes, which are well known in the art, and peripheral
to the teachings of the present disclosure. However, Adaptive amplitude
control adjustment is briefly discussed as follows.

Adaptive Amplitude Adjustment

[0058] As is well known, RF spectral amplitude adjustment may be
implemented by forming a filter of desired shape. Filters of differing
shape can be formed in parallel and a controller can select the best
match for the application but there is often no way of knowing a priori
which filter will best match the application. Another way of selecting a
filter output, or even generating a new filter from a composite of a
number of filters, is to weight and sum each of the filter outputs in a
variable weighting structure. A controller is provided which has a
feedback mechanism such that it can change the weighting and summation
network weight values and then evaluate the change. Adaptive amplitude
control 225 implements this process by monitoring the protected output 58
of ACL 6 (See FIG. 1a) while dithering control lines that adjust the
weights of the quadratic amplitude matching filter (QAMF) under a
sequence determined by its algorithm and loop feedback.

[0059] Referring now to FIG. 2a, a circuit structure is shown for forming
a tunable variable lobed filter 250, according to one embodiment. In this
embodiment, the tunable variable lobed filter 250 is implemented using a
power divider 252, a variable delay 254 and a summing junction 256. The
tunable variable lobed filter 250 is tunable by changing the variable
delay value [T] 254. Tuning the variable delay value [T] 254 causes
expansion and contraction of each lobe from zero and thus a shift of
every lobe, beyond the first, up or down in frequency.

[0060] FIG. 2b is an alternative circuit structure 260 of the tunable
variable lobed filter 250 implemented with a difference hybrid 258 as a
substitute for the summing junction 256. This creates a functionally
similar tunable variable lobed filter 250 as described above but has
orthogonal lobes to the structure of FIG. 2a, providing an important
mathematical relationship to be used in control of the tuning process, as
discussed immediately below and also further below with reference to FIG.
3.

[0061] The inventor has recognized two important mathematical
relationships that together allow tuning over a large bandwidth and
control of a more narrowband filter to provide the desired amplitude
shaping effect. The first important mathematical relationship relates to
the orthogonal nature of the sine and cosine function of two RF filters
simultaneously formed from the same power divider 252 and time delay
structure when combined in either a sum or difference port of the tunable
variable lobed filter 250, as briefly discussed above. The first
recognized mathematical relationship allows the use of a null at one
frequency in a sine filter to align with the lobe of the cosine filter,
or vice versa, and can be used as a sensitive tuning control, as
illustrated in FIG. 3, and described below.

[0062] The second important mathematical relationship is the recognition
that two RF filters, one tuned with time delay T and the other tuned with
a further time delay (2n+1)T, where n is an integer, will always have
lobes co-aligned at the center of the wider band lobe. It is noted that
the relationship is one of the further time delay being an odd multiple
of a basic delay T. The implications of such a relationship are described
in more detail further below with respect to FIG. 4.

[0063] Referring now to FIGS. 3a-3b, there is shown an output of a lobed
filter, such as, for example, the tunable variable lobed filter 250 of
FIGS. 2a and 2b. The output is represented as curve 390 in FIG. 3a (and
further illustrated in expanded form in FIG. 3b).

[0064] Referring to FIG. 3a, the output 390 of tunable variable lobed
filter 250 is shown as a magnitude (cosine) function of the delay
difference in the two paths, i.e., path A and path B, shown in FIG. 2a.
The lobed filter amplitude of output curve 390 of FIG. 3a repeats at a
regular spacing of BWn equal to (2T)-1. As stated above, in an
alternate embodiment, a difference hybrid 258 (See FIG. 2b) can be
substituted for the summing junction 256 (See FIG. 2a) of the tunable
variable lobed filter 250 FIG. 2a. In such an embodiment, the output 390
of the tunable variable lobed filter 250 follows a magnitude (sine)
function, represented as curve 391 in FIG. 3a. Thus, a time delay can be
selected to have the tunable variable lobed filter 250, 260 extend beyond
a band of interest and a corresponding orthogonal filter will have a null
within the tuning bandwidth. For example, by extending tunable variable
lobed filter 250 of FIG. 2a beyond a band of interest it will have a null
390 within the tuning band of interest. As a further example, by
extending tunable variable lobed filter 260 of FIG. 2b beyond a band of
interest, it will have a null 391 within the tuning band of interest.

[0065] It should be appreciated that the null to null bandwidth, BWn
of the lobed tunable variable lobed filter 250 is inversely proportional
to the time delay, T 254, as shown in FIGS. 2a and 2b. Therefore, an
increase in the time delay T 254 reduces the bandwidth BW,, of the
tunable variable lobed filter 250. Further, by changing the time delay to
be an odd multiple of a basic delay T, for example, by (2n+1)T, where n
an integer, the original tunable variable lobed filter 250 is effectively
split into (2n+1) lobes. As this always results in an odd number of
lobes, one lobe 402, necessarily is always centered with the tuning lobe
404 of a broadband tuning filter, as shown in FIG. 4. This single
centered lobe 402 becomes useful in the quadratic amplitude matching
filter (QAMF) structure 100 (See FIGS. 7 and 8) to be weighted by the
adaptive amplitude control 225 controller to shape the coupled signal to
match the propagated interfering propagated reference signal, contained
in antenna signal 30 in an interference cancellation system. The value of
n used to effectively split the output of the filter into 2n+1 lobes can
be adjusted to achieve the desired flatness at quiescent with minimal
propagation path distortion as required for slaved lobe filter
structure-flat 103 but can also be adjusted to the value m to provide the
required quadratic down shaping required for slaved lobe filter
structure-down 104. It is also contemplated to use the values of n and m
as variables for finer tuning control in future envisioned
implementations of the improved cosite interference cancellation system.

[0066] FIGS. 5a-5c illustrate, by way of example, plots of different
exemplary tuning scenarios to further illustrate the concept of
generating a lobed filter orthogonal to the lobe of a broadband tuning
filter. It should be understood that, in accordance with invention
principles, a tuning filter lobe of the broadband tuning filter is not
necessary for actual operation, and is not necessarily formed in actual
operation. It will therefore be referred to hereafter as a so-called
imaginary tuning filter lobe. It should be understood, however that the
quadratic amplitude matching filter (QAMF) will track the center of the
so-called imaginary tuning filter lobe by use of the orthogonal null
formed off-line in the timing delay tuning control (TDTC) 202, as shown
in FIGS. 1, 9 and 10. Herein, inline refers to an action or process that
generates an immediate change, upon signals passing through, at the
output of the circuit where offline refers to action or processes that
may use samples of signals passing through but do not impact the signals
passing through until a result is reached and a change is made to the
inline processes.

[0067] Each of the plots of FIGS. 5a-5c illustrates a common insertion
signal 511 to be tracked. The insertion signal represents the sample of
transmitted signal 40 to be matched to an undesirable multipath signal
received in antenna signal 30 to be cancelled by the improved cosite
cancellation circuit 20 of FIG. 1.

[0068] Referring first to FIG. 5a, four output filter curves are shown
511, 512 NO, 513 NL, 514 NH. Output filter curves 512
NO, 513 NL and 514 NH represent three different filters
tuned with a so-called imaginary tuning filter lobe but orthogonal to the
imaginary tuning filter lobe such that nulls of the orthogonal filter are
aligned with the peak of a lobe of the original filter formed by the same
delay, T. A first filter output curve 512 NO represents the null
portion of a lobed filter, NO, formed by current value of delay T,
orthogonal to the tuning filter tuned on frequency with the imaginary
tuning filter by using the same delay T used to form the tuning filter.
Using the same delay used to form both the first filter output curve 512
NO and the imaginary tuning filter, results in a null of the first
filter output curve 512 NO aligned with the imaginary tuning lobe of
the tuning filter, as shown in FIG. 3.

[0069] A second filter output curve 513 NL represents the null
portion of a lobed filter, NL, formed by delay T+ΔT, an
incremental step of delay time T 254 of the circuit of FIG. 2 tuned low
in frequency with a path delay difference of T+ΔT and results in a
null below, or lower than the current center frequency of the imaginary
tuning lobe of the broadband lobed filter.

[0070] A third filter output curve 514 NH represents the null of the
lobed filter, NH, is tuned high in frequency with a path delay
difference of T-ΔT and results in a null above, or higher than, the
current center frequency of the imaginary tuning lobe of the broadband
lobed filter.

[0071] With continued reference to FIG. 5a, there is shown the condition
in which the filter, NO, is centered at a frequency that is below
the frequency of the insertion signal 511. In this case, the filter
NH allows more of the inserted signal energy of the inserted signal
511 through, than the filter NL thus providing feedback to the
interference cancellation system to move the tuning filter higher in
frequency by decreasing the delay, T.

[0072] FIG. 5b illustrates the case in which the filter, NO, is
centered at a frequency that is above the frequency of the insertion
signal 511. In this case, the filter NL allows more of the inserted
signal energy of the inserted signal 511 through, than the filter NH
thus providing feedback to the interference cancellation system to move
the tuning filter lower in frequency by increasing the delay, T.

[0073] FIG. 5c illustrates the case where conditions when the filter,
NO, is centered on the inserted signal. In this case, the low and
high filters, NH and NL, will pass equal amounts of the
inserted signal energy, thus providing no feedback to change frequency by
changing the delay, T. This state represents a point of stability in
tuning such that, as the null of the orthogonal filter is aligned with
the inserted signal and thus aligned with the peak of the center of the
lobe of the imaginary tuning filter, the inserted signal is thus aligned
with the peak of the quadratic amplitude matching filter (QAMF) center.

[0074] It should be understood that the direction of the null shifts as a
function of the time delay introduced by the interference cancellation
system is inherent to lobed filters which are comprised of a plurality of
nulls originating at zero Hz and repeating at a regular spacing of
(2T)-1. Thus an increase in delay T reduces the effective BWn,
thereby compressing the lobing and shifts the current null to the left,
i.e., lower in frequency.

[0075] Referring again to FIG. 5a, the center null 512 NO is
representative of a filter output null which is orthogonal to the
corresponding filter output formed by the summation of the output of
filter signal paths with path delay differences formed by the inline
delay T.

[0076] The left null 513 NL represents the null of a filter output
having a path delay T+Δ/t, the output exhibiting a slightly more
narrow lobed structure than the output of a filter signal path having a
path delay T, and thus the repetitive lobing shifts to the left, lower in
frequency, moving the null below the nominal location using delay T.

[0077] The right null 514 NH represents the null of a filter output
having a path delay T-Δt, the output exhibiting a slightly wider
lobed structure than the output of a filter signal path having a path
delay T, and thus the repetitive lobing shifts to the right, higher in
frequency, moving the null above the nominal location using delay T.

[0078] It should be appreciated that these two filter output curves 513
NL, 514 NH advantageously allow different amounts of the
incident signal energy to pass through them. In this manner, measurement
of the energy from the respective filter outputs provides information on
a corrective direction in frequency of the tuning lobe orthogonal filter
required for proper tuning.

[0079] With continued reference to FIG. 5a, this figure further
illustrates a set of undesirable image nulls 515. It is appreciated that
these undesirable image nulls 515 are a limitation to the tuning
bandwidth of the tuning control system. They arise by using too large of
a value of delay T, resulting in an excess of narrow lobes for tuning. It
therefore follows that it is desirable to have as large a tuning
bandwidth as possible to preclude the creation of these image nulls. It
is preferred that tuning to the low edge of the frequency tuning band
cannot allow image nulls to approach the high band limit for inserted
signal, or vice versa, or the system may shift lobes of the tuning filter
upon a jump in transmitted signal frequency, and cause significant change
in subsequent filter bandwidths and thus shaping amplitude factors. The
narrowband filter cannot be used for tuning because of this limitation.
This shows the importance of the recognition of the lobe alignment for
filters formed by T and (2n+1)T delays so that the tuning filter lobe can
be very broad for a broad tuning range but still be used to focus a much
more narrow lobe for quadratic amplitude matching filter (QAMF) function.

[0080] As stated above, a primary objective of the tuning control system
of the present disclosure is to continuously and automatically tune a
quadratic amplitude matching filter (QAMF) to an interferer band center
as a quiescent starting point for performing quadratic amplitude control
adjustment. While it is understood that amplitude control adjustment is
not central to the teachings of the present disclosure, it is understood
that it is implemented by controlling the weights of the tuned quadratic
amplitude matching filter (QAMF), tuned in accordance with invention
principles.

[0082] Referring now to FIGS. 7a-7c, there is shown, by way of example, a
plot of four curves. For ease of explanation, each signal has been offset
in level for clarity and each successive plot is an expansion of the
center area of the previous plot, as indicated by the common centerline.

[0083] The first curve 610 is representative of an imaginary broadband
tuning filter formed by a delay interval T, corresponding to an off-line
lobed tuning filter with an orthogonal null to allow it to track an
incoming signal of interest. The second curve 612 is representative of a
lobed filter tracking the tuning filter with a delay interval (2n+1)T,
where n is some integer multiplier of T. In this case, the lobed filter
tracks the off-line broadband tuning filter null, as is true of the first
curve 610 formed with a delay T, however, in the present case, the filter
is more narrow in bandwidth although still nearly flat in the region of
the bandwidth of signal of interest, as generated by Slaved lobe filter
structure-flat 103 and output at 112 (See FIGS. 1 and 8).

[0084] The third curve 614 is representative of the lobed filter tracking
the tuning filter with a multiplier value of m=28 in the present example
such that it tracks the imaginary broadband tuning filter lobe center and
the slaved lobe filter structure-flat 103 (See FIGS. 1 and 8). This lobe
structure is even more narrow in bandwidth than the lobe structure
generated by slaved lobe filter structure-flat 103 and output at 112 (See
FIGS. 1 and 8). The presently described lobe structure has amplitude
shaped as a down quadratic in the region of the bandwidth of signal of
interest, i.e., the "signal bandwidth" region, as generated by slaved
lobe filter structure-down 104 and output at 118.

[0085] The fourth curve is representative of a more complex FIR filter
(e.g., formed using 5 taps by way of example and not limitation). In this
embodiment, weights having values of 1.0, 1.0, -1.0, 1.0, 1.0 are used to
create an upward quadratic curve in the region of the signal of interest
as generated by slaved lobe filter structure-up 106 and output at 122.
Further, the tuning filter is tracked using a tap spacing of T with
multiplier of o (o=14 in this example) such that it tracks the tuning
filter lobe center, the slaved lobe filter structure-flat 103 and the
slaved lobe filter structure-down 104.

[0086] Referring now to FIG. 7c, there is shown a region labeled "Signal
Bandwidth" for illustrating the alternate signal path amplitude filter
shapings, before weighting and combining, implemented upon the coupled
transmitted signal 40 (see FIG. 1a) in the quadratic amplitude matching
filter (QAMF) 100. The coupled transmission signal 40 is desirably shaped
by a cosite cancellation circuit 20 of the protected receiver 25 for the
purpose of matching the distortion introduced in the propagated
interfering propagated reference signal, contained in antenna signal 30
for improved interference cancellation.

[0087] FIG. 8 is a more detailed circuit diagram of the quadratic
amplitude matching filter (QAMF) 100 of FIG. 1. In the presently
described embodiment, a quadratic amplitude adjustment 100 is implemented
as a block of three parallel filters 103, 104, 105, each respectively
formed in standard finite impulse response filters having different
characteristics of amplitude shapings across the band of interest and
each being formed based upon different odd integer multiples of a basic
delay interval T which tunes such structures to a central band of
interest of an interfering signal.

[0088] The three parallel filter blocks include the slaved lobe filter
structure-flat block 103, the slaved lobe filter structure-down block
104, and the slaved lobe filter structure-up block 105. Each filter block
103, 104, 105 uses a common digital control signal WT 129 as a
tuning signal to track a center frequency with different relative
bandwidths, to be described as follows.

[0089] Slaved lobe filter structure-flat 103 includes an equalizing delay
block and a simple filter. The signal enters a time delay ta 111
controlled by the WT 129 but internal circuitry is designed to
adjust the setting of the delay implemented in ta 111 to cause the
total delay through the slaved lobe filter structure-flat 103 to that
matching the simultaneous delays through slaved lobe filter
structure-down 104 and slaved lobe filter structure-up 105. The delayed
signal enters the simple filter at a power divider 108 forming two paths,
one of which feeds directly into one port of a summing junction 110 while
the second path enters a controlled delay line 109 which is set to a
delay (2n+1)T by the same control WT 129 before entering a second
port of the summing junction 110. This delay corresponds to the delay
that would tune a similar lobed filter used for amplitude slope control
to the band of interest and bandwidth such that it is nearly flat in the
region of the signal of interest. The signal exits the summing junction
and enters a weighting device

[0090] Slaved lobe filter structure-down block 104 is the same structure
as slaved lobe filter structure-flat block 103 except that the filter
time constant for Slaved lobe filter structure-down block 104 is
increased to narrow the lobing of the filter to generate a quadratic
shape in the area of the signal of interest.

[0091] Slaved lobe filter structure-down block 104 includes an equalizing
delay block 117 and a simple filter. The signal enters a time delay
tb 117 controlled by the same WT 129. Internal circuitry (not
shown) is included in slaved lobe filter structure-down 104 to adjust the
setting of the delay implemented in tb 117 to cause the total delay
through the slaved lobe filter structure-down block 104 to match the
simultaneous delays through slaved lobe filter structure-flat block 103
and slaved lobe filter structure-up block 105. The delayed signal enters
the simple filter at a power divider 114 forming two paths, one of which
feeds directly into one port of a summing junction 116 while the second
path enters a controlled delay line 115 which is set to a delay (2m+1)T
by the control WT 129 before entering a second port of the summing
junction 116. This delay is based upon the same tuning interval T but has
a multiplier of (2m+1). Multiplier m establishes the relative bandwidth
of the quadratic filter shaping.

[0092] Slaved lobe filter structure-up block 105 includes an equalizing
delay block 121 and a simple FIR filter. The signal enters a time delay
tc 121 controlled by the same WT 129 but internal circuitry is
designed to adjust the setting of the delay implemented in tc 121 to
cause the total delay through the slaved lobe filter structure-up block
105 to that matching the simultaneous delays through slaved lobe filter
structure-flat block 103 and slaved lobe filter structure- down block
104. The delayed signal enters the finite impulse response (FIR) filter
120 of multiple taps spaced at intervals based upon the same tuning
interval T but having a multiplier of (2o+1) and individually weighted to
generate the desired upward quadratic function at the frequency of the
signal of interest. The multiplier o will cause the filter function to
track the signal of interest as the frequency changes and the circuit is
tuned in response.

[0093] The outputs of the three slaved lobe filter shaping structures 103,
104, 105 are each respectively weighted and summed to generate the
amplitude shaping necessary to match a received signal which has been
distorted in a dispersive multipath propagation path from a co-located
transmission antenna 2. Quiescent values of Wp1 113, Wp2 119,
and Wp3 123 will be (1.0, 0.0, 0.0), providing minimal shaping are
modified thereafter by the adaptive amplitude control 225.

[0094] FIG. 9 provides the circuit structure of a tuning control system,
according to one embodiment, for continuously and automatically tuning a
quadratic amplitude matching filter (QAMF) 100 to a band center of a
reference input signal for improved interference cancellation system
performance in a cosite interference cancellation system.

[0095] An Adaptively Tuned Quadratic Control (ATQC) module 200 comprises
two main elements; an inline quadratic amplitude matching filter (QAMF)
100 and an offline Time Delay Tuning Control (TDTC) element 202. The
quadratic amplitude matching filter (QAMF) 100 further includes a
variable lobe filter structure (VLFS) 201, slaved lobe filter
structure-down 104 and a slaved lobe filter structure-up 105. It is noted
that Variable lobe filter structure (VLFS) 201 is a variation of slaved
lobe filter structure-flat 103 of FIG. 1 with the time delay structure
split into two blocks to provide a tuning feed.

[0096] The variable lobe filter structure (VLFS) 201 implements the
functionality of a tuned and quiescent amplitude slope matched filters of
the prior art but is constructed in a novel manner as a variation of a
conventional in-line Lobe Filter Structure such that the delay line
forming the lobed filter is split into two blocks of controlled variable
time delay. Specifically, the delay line is split into a first block 205
with delay T and a second block 208 with delay 2nT, yielding a total
delay of (2n+1)T. The first block 205 is used for broadband tuning and
the second block 208 is implemented as a multiple of the first block 205,
thus making the nearly-flat path more narrowband. As discussed above, a
resulting nearly-flat path filter lobe formed by the (2n+1)T relationship
is centered in the lobe of an imaginary filter orthogonal to the null of
the broadband tuning filter lobe formed by the delay T but is never
actually formed. This establishes one path of the quadratic amplitude
matching filter (QAMF) 100. The other two paths are slaved to the tuning
value T and thus track the inserted coupled transmitted signal. The
weights on the three paths are then adjusted for quadratic amplitude
matching, as taught above and then fed into port 8 of the ICS for
improved interference cancellation, advantageously requiring no control
signals from the transmitter.

[0097] The Timing Delay and Tuning Control (TDTC) module 202 uses signal
samples output from the variable lobe filter structure (VLFS) 201 to
control the first block, T 205 for tuning, which is central to the
teachings of the present disclosure, and the second block, 2nT 208, to
implement the lobed filter function-flat used by the interference
cancellation system, which is a pre-requisite for implementing the
amplitude slope adjustment to match the propagated path of the
interfering signal.

[0098] It should be understood that the tuning filter lobe is referred to
herein as imaginary in the sense that it is never actually formed or used
in actual operation but is instead discussed herein to provide a more
complete understanding of the interrelationships of the various control
signals and the generation of the ASMF filter.

[0099] With continued reference to FIG. 9 reference signal 215 and delayed
signal 216 are sampled and fed into a differencing hybrid 221 to form a
broadband RF filter, a first broadband RF filter having a filter response
223. Thus the offline sine function filter formed using the variable T
205 has a null orthogonal to the center of the imaginary tuning filter
lobe based upon this T, which is never formed.

[0100] In accordance with a method for continuously and automatically
tuning a quadratic amplitude matching filter (QAMF) 100 to a band center
for improved interference cancellation system performance, the energy
passing through the filter is measured by controller 222. The controller
can then dither the digital delay control 232 by a value ΔT, either
positive or negative and re-measure the energy passing through the
filter. In this way, the controller can determine direction to skew the
tuning filter to achieve a null on the input signal 203. This assumes
increasing a control voltage causes the delay T to be increased resulting
in a narrowing of the filter lobes while decreasing the control voltage
causes the delay T to be decreased resulting in a broadening of the
filter lobes. Signs can be easily changed for devices with opposite
control functions. There are a number of search algorithms common in the
art to perform this control that provide for desired rejection of noise
and timely convergence. These include but are not limited to random
searching, gradient searching, and perturbation using orthogonal Walsh
functions, each with advantages and disadvantages. The selected control
algorithm is not part of this present disclosure.

[0101] FIG. 10 is a more detailed circuit diagram of FIG. 1 for
illustrating an improved interference cancellation circuit 20 for
elimination of interfering signals between radio transmitter 21, and
receiver 25 where system dynamics cause changes in the coupling between a
transmit and receive antenna on a platform, according to one embodiment.

[0102] A time delayed, quadratic amplitude matched sample of transmission
signal 40 is output from adaptively tuned quadratic control 200 as the
delayed coupled signal 57 and supplied to auxiliary port 8 of ACL 6.
Interfering propagated reference signal, contained in antenna signal 30
is fed into reference port 9 of ACL 6. A cancellation signal 65 is
generated by ACL 6 via the processes of autocorrelation 66, integration
67, and finally by applying a complex weight 68 of phase and amplitude.
The cancellation signal 65 is provided to summing junction 70. It is
noted that when the cancellation signal 65 is injected into summing
junction 70 it has substantially the same amplitude as the interfering
propagated reference signal, contained in antenna reference signal 71,
however, the cancellation signal 65 is manipulated so that it is
180° out of phase with the interfering propagated reference signal
received by antenna 4 and included in antenna signal 30 so as to
substantially cancel the interfering signal. The adaptive amplitude
control 225 of prior art is still required to adjust the quadratic
amplitude matching filter (QAMF) 100 to the proper weights to match the
quadratic amplitude distortion of the sampled transmission signal to that
of the propagation path. Adaptive amplitude control 225 implements this
process by monitoring ICS protected output 58 while dithering control
lines Wp1 113, Wp2 119, and Wp3 123 under a sequence
determined by its algorithm and loop feedback. As a result, the signal
remaining on the protected output 58 is substantially the same as the
received antenna signal 30 provided by receiver antenna 4 without the
undesired contribution from interfering transmitter 1. ACL 6 is
configured as a Least Mean Square (LMS) analog control loop but those
familiar with the art will realize that many different algorithms,
implemented at RF and digital, can serve this function.

[0103] The use of sum or difference hybrids in the off-line processing and
in-line processing may be switched to design the system for a specific
tuning band and slope control lobed filter width. This embodiment is just
one configuration.

[0104] If the tuning information is available from the transmitter, it
could be used for a table lookup of the starting point for the value of
T. Thus, when the transmitter switched frequency, tuning would start at
approximately the correct value. These stored values may be a one-time
set value at manufacture or may be updated every time the frequency is
visited.

[0105] Referring now to FIG. 11 there is shown four co-located interfering
transmitters 21a-21d, by way of example and not limitation. Four are
shown for ease of explanation. To counteract the multiple interfering
transmitters 21a-21d, and thus reduce or minimize cosite interference,
the improved cosite interference cancellation system 20 includes a common
adaptive amplitude control 225 of prior art operably coupled to a common
summing junction for four Interference Cancellation Systems (ICS) 6a-6d,
and four independent Adaptively Tuned Quadratic Control (ATQC) module
200a-200d comprised of two main elements; an inline quadratic amplitude
matching filter (QAMF) 100 and an offline Time Delay Tuning Control
(TDTC) element 202 operably coupled to a common summing junction for four
Interference Cancellation Systems (ICS) 6a-6d. Four of which are shown
for ease of explanation and not limitation. In this manner, the cosite
interference cancellation process described above with reference to FIG.
10 is independently applied to each interfering transmitter 21a-21d to
protect the single receiver 25. This figure shows a preferred embodiment
with common, shared antenna signal 30, summing junction 70 and antenna
reference signal 71. The function of the Variable Lobe Filter Structure
(VLFS) 201a-201d are in-line and must be independent but the function of
the Time Delay Tuning Control (TDTC) element 202 and adaptive slope
control 225 can be shared through multiplexing techniques implemented in
prior art of adaptive arrays where the correlation and integration
functions were shared. In other embodiments, the ICS summing junctions
are daisy-chained for the use of a standard building block at the cost of
additional potential noise insertions and longer convergence times
because of signal interaction.

[0106] Referring now to FIG. 12 there is shown an improved cosite
interference cancellation system 20 for elimination of interfering
signals between a single co-located transceiver 21 and a plurality of
receivers to be protected. In the presently described embodiment, it is
desired to protect a multiplicity of receivers, 25a-25d, four of which
are shown by way of example and not limitation. To protect the plurality
of receivers 25a-25d, each receiver is coupled to a corresponding
Interference Cancellation Systems (ICS) 6a-6d operably coupled with
associated independent Adaptively tuned quadratic control 200a-200d each
comprises three main elements; a Variable Lobe Filter Structure (VLFS)
201, a Time Delay Tuning Control (TDTC) element 202, and a common
adaptive amplitude control 225 of prior art.

[0107] The foregoing is construed as only being an illustrative embodiment
of this invention. Persons skilled in the art can easily conceive of
alternative arrangements providing a functionality similar to this
embodiment without any deviation from the fundamental principles or the
scope of the invention.