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Abstract:

A method and apparatus for improving channel estimation within an OFDM
communication system. Channel estimation in OFDM is usually performed
with the aid of pilot symbols. The pilot symbols are typically spaced in
time and frequency. The set of frequencies and times at which pilot
symbols are inserted is referred to as a pilot pattern. In some cases,
the pilot pattern is a diagonal-shaped lattice, either regular or
irregular. The method first interpolates in the direction of larger
coherence (time or frequency). Using these measurements, the density of
pilot symbols in the direction of faster change will be increased thereby
improving channel estimation without increasing overhead. As such, the
results of the first interpolating step can then be used to assist the
interpolation in the dimension of smaller coherence (time or frequency).

Claims:

1. A method, comprising: receiving channel estimates for four pilot
symbols in a scattered pilot pattern in time-frequency; calculating the
channel response for the pilot symbols in both a first direction and a
second direction; determining whether the channel changes more slowly in
one direction than the other; interpolating in the direction of slower
channel change.

2. The method of claim 1, further comprising; interpolating in the
direction of faster channel change.

3. The method of claim 2, wherein interpolating in the direction of
faster channel change is performed using the result from the step of
interpolating in the direction of slower channel change.

4. The method of claim 1, wherein the channel changes are calculated by
performing an inner products operation.

5. The method of claim 1, wherein the first direction is a time direction
and the second direction is a frequency direction.

6. The method of claim 1, wherein the first direction is a frequency
direction and the second direction is a time direction.

7. The method of claim 1, wherein the scattered pilot pattern is a
regular diamond lattice.

8. The method of claim 1, wherein the scattered pilot pattern is an
irregular diamond lattice.

10. An OFDM receiver, comprising: one or more receive antennas; the OFDM
receiver being adapted to receive channel estimates for four pilot
symbols in a scattered pilot pattern in time-frequency, calculate channel
changes for the pilot symbols in a first direction and a second
direction, and interpolate in the direction of slower channel change.

11. The OFDM receiver of claim 10, wherein said receiver is further
adapted to interpolate in the direction of faster channel change.

12. The OFDM receiver of claim 11, wherein interpolating in the direction
of faster channel change is performed using the result from interpolating
in the direction of slower channel change.

13. The OFDM receiver of claim 10, wherein the channel changes are
calculated by performing an inner products operation.

14. The OFDM receiver of claim 10, wherein the first direction is a time
direction and the second direction is a frequency direction.

15. The OFDM receiver of claim 10, wherein the first direction is a
frequency direction and the second direction is a time direction.

19. A method of interpolation using a set of four pilot symbols in a
scattered pilot pattern in time-frequency wherein the set of four pilot
symbols comprise first and second pilot symbols on a common sub-carrier
frequency, spaced in time, and third and fourth pilot symbols transmitted
on different sub-carriers on a common OFDM symbol period, the method
comprising: determining a first channel change between the first and
second pilot symbols; determining a second channel change between the
third and fourth pilot symbols; determining which of the first and second
channel change is slower; if the first channel change is slower,
interpolating using the first and second pilot symbols to generate a
channel estimate for the common sub-carrier frequency at the common OFDM
symbol period, and then using the channel estimate in subsequent
interpolations to determine other channel estimates; and if the second
channel change is slower, interpolating using the third and fourth pilot
symbols to generate a channel estimate for the common sub-carrier
frequency at the common OFDM symbol period, and then using the channel
estimate in subsequent interpolations to determine other channel
estimates.

20-33. (canceled)

34. A non-transitory, computer accessible memory medium storing program
instructions, wherein the program instructions are executable by one or
more processors to: receive channel estimates for four pilot symbols in a
scattered pilot pattern in time-frequency; calculate the channel response
for the pilot symbols in both a first direction and a second direction;
determine whether the channel changes more slowly in one direction than
the other; and interpolate in the direction of slower channel change.

35. The non-transitory, computer accessible memory medium of claim 34,
wherein the program instructions are further executable to: interpolate
in the direction of faster channel change.

36. The non-transitory, computer accessible memory medium of claim 35,
wherein interpolating in the direction of faster channel change is
performed using the result from the step of interpolating in the
direction of slower channel change.

38. The non-transitory, computer accessible memory medium of claim 34,
wherein the first direction is a time direction and the second direction
is a frequency direction.

39. The non-transitory, computer accessible memory medium of claim 34,
wherein the first direction is a frequency direction and the second
direction is a time direction.

Description:

CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This application is a continuation of U.S. patent application Ser.
No. 12/064,566, filed on Sep. 4, 2008, which is a National Phase Entry of
International Application No. PCT/CA2006/001380, filed on Aug. 22, 2006,
which claims the benefit of U.S. Provisional Patent Application No.
60/710,527 filed on 5 Aug. 23, 2005 and U.S. Provisional Patent
Application No. 60/722,744 filed on Sep. 30, 2005, all of which are each
hereby incorporated by reference in their entirety, as if fully and
completely set forth herein.

FIELD OF THE INVENTION

[0002] This invention relates to Orthogonal Frequency Division
Multiplexing (OFDM) communication systems, and more particularly to
channel interpolation with the use of pilot symbols.

BACKGROUND OF THE INVENTION

[0003] In wireless communication systems that employ OFDM, a transmitter
transmits data to a receiver using many sub-carriers in parallel. The
frequencies of the sub-carriers are orthogonal.

[0004] Channel estimation in OFDM is usually performed with the aid of
known pilot symbols which are sparsely inserted in a stream of data
symbols. The attenuation of the pilot symbols is measured and the
attenuations of the data symbols in between these pilot symbols are then
estimated/interpolated.

[0005] Pilot symbols are overhead, and should be as few in number as
possible in order to maximize the transmission rate of data symbols. It
is desirable that channel estimation in OFDM be as accurate as possible
without sacrificing bandwidth.

SUMMARY OF THE INVENTION

[0006] In one embodiment, there is provided a method comprising receiving
channel estimates for four pilot symbols in a scattered pilot pattern in
time-frequency; calculating the channel response for the pilot symbols in
both a first direction and a second direction; determining whether the
channel changes more slowly in one direction than the other; and
interpolating in the direction of slower channel change.

[0007] In some embodiments, the method of further comprises interpolating
in the direction of faster channel change.

[0008] In some embodiments, the step of interpolating in the direction of
faster channel change is performed using the result from the step of
interpolating in the direction of slower channel change.

[0009] In some embodiments, the channel changes are calculated by
performing an inner products operation.

[0010] In some embodiments, the first direction is a time direction and
the second direction is a frequency direction.

[0011] In some embodiments, the first direction is a frequency direction
and the second direction is a time direction.

[0012] In some embodiments, the scattered pilot pattern is a regular
diamond lattice.

[0013] In some embodiments, the scattered pilot pattern is an irregular
diamond lattice.

[0014] In some embodiments, the scattered pilot pattern is kite shaped.

[0015] In another embodiment, there is provided an OFDM receiver
comprising: one or more receive antennas; the OFDM transmitter being
adapted to receive channel estimates for four pilot symbols in a
scattered pilot pattern in time-frequency, calculate channel changes for
the pilot symbols in a first direction and a second direction, and
interpolate in the direction of slower channel change.

[0016] In yet another embodiment, there is provided a method of
interpolation using a set of four pilot symbols in a scattered pilot
pattern in time-frequency wherein the set of four pilot symbols comprise
first and second pilot symbols on a common sub-carrier frequency, spaced
in time, and third and fourth pilot symbols transmitted on different
sub-carriers on a common OFDM symbol period, the method comprising:
determining a first channel change between the first and second pilot
symbols; determining a second channel change between the third and fourth
pilot symbols; determining which of the first and second channel change
is slower; if the first channel change is slower, interpolating using the
first and second pilot symbols to generate a channel estimate for the
common sub-carrier frequency at the common OFDM symbol period, and then
using the channel estimate in subsequent interpolations to determine
other channel estimates; and if the second channel change is slower,
interpolating using the third and fourth pilot symbols to generate a
channel estimate for the common sub-carrier frequency at the common OFDM
symbol period, and then using the channel estimate in subsequent
interpolations to determine other channel estimates.

[0017] In yet another embodiment, a method of inserting pilot symbols into
OFDM sub-frames for transmission by a plurality of transmitting antenna,
the OFDM sub-frames having a time domain and a frequency domain, each
OFDM sub-frame comprising a plurality of OFDM symbols, the method
comprising: for each sub-frame, defining a set of at least two OFDM
symbols none of which are consecutive that are to contain pilot symbols;
at each antenna, inserting pilot symbols in each of the set of at least
two OFDM symbols in a scattered pattern that does not interfere with the
scattered pattern inserted by any other antenna.

[0018] Other aspects and features of the present invention will become
apparent to those ordinarily skilled in the art upon review of the
following description of specific embodiments of the invention in
conjunction with the accompanying figures.

BRIEF DESCRIPTION OF THE DRAWINGS

[0019] Preferred embodiments of the invention will now be described with
reference to the attached drawings in which:

[0020] FIG. 1 is a diagram of a single antenna perfect diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention;

[0021]FIG. 2 is a flowchart of a method of performing adaptive
interpolation in accordance with one embodiment of the present invention;

[0046]FIG. 14 is a block diagram of an example base station that might be
used to implement some embodiments of the present invention;

[0047]FIG. 15 is a block diagram of an example wireless terminal that
might be used to implement some embodiments of the present invention;

[0048]FIG. 16 is a block diagram of a logical breakdown of an example
OFDM transmitter architecture that might be used to implement some
embodiments of the present invention;

[0049]FIG. 17 is a block diagram of a logical breakdown of an example
OFDM receiver architecture that might be used to implement some
embodiments of the present invention; and

[0050]FIG. 18 is a block diagram of one embodiment of the present
invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0051] Channel estimation in OFDM is usually performed with the aid of
pilot symbols. More particularly, at an OFDM transmitter, known pilot
symbols are periodically transmitted along with data symbols. The pilot
symbols are typically spaced in time and frequency.

[0052] The variations in phase and amplitude resulting from propagation
across an OFDM channel are referred to as the channel response. The
channel response is usually frequency and time dependent. If an OFDM
receiver can determine the channel response, the received signal can be
corrected to compensate for the channel degradation. The determination of
the channel response is called channel estimation. The transmission of
known pilot symbols along with data symbols allows the receiver to carry
out channel estimation.

[0053] When a receiver receives an OFDM signal, the receiver compares the
received value of the pilot symbols with the known transmitted value of
the pilot symbols to estimate the channel response.

[0054] Since the channel response can vary with time and with frequency,
the pilot symbols are scattered amongst the data symbols to provide a
range of channel responses over time and frequency. The set of
frequencies and times at which pilot symbols are inserted is referred to
as a pilot pattern. In some cases, the pilot pattern is a diagonal-shaped
lattice, either regular or irregular.

[0055] FIG. 1 is an example pilot pattern which can be used in accordance
with one embodiment of the present invention. Pilot and data symbols are
spread over an OFDM sub-frame in a time direction 120 and a frequency
direction 122. Most symbols within the OFDM sub-frame are data symbols
124. Pilot symbols 126 are inserted in a diamond lattice pattern. In the
illustrated example, the diamond lattice pattern in which each encoded
pilot symbols are inserted within the OFDM sub-frame is a perfect diamond
lattice pattern as illustrated by pilot symbols h1, h2, h3
and h4.

[0056] A two dimensional interpolator is used to estimate the channel
response at point h which is between four points of known channel
response, i.e. pilot symbols h1, h2, h3 and h4. Point
h can then be used as an additional point from which the receiver can
carry out channel estimation. The use of point h would, of course, not
add any overhead to the OFDM signal.

[0057] The channel interpolation scheme is adaptive, i.e. it is a scheme
which can adapt to varying conditions in the

h(i,j)=w1(i,j)h1+w2(i,j)h2+w3(i,j)h3+w.sub-
.4(i,j)h4

channel. The following formula presents a particular example of adaptive
two-dimensional (time direction and frequency direction) interpolator to
calculate point h:

[0058] where

w1(i,j)+w2(i,j)+w3(i,j)+w4(i,j)=1

[0059] In this case, the two dimensional channel interpolation can be
viewed as the sum of two one-dimensional interpolations.

[0060] The weights wk(i,j) may be adapted to coherence time and
frequency of the channel. In some embodiments, if the channel coherence
is less in the time direction than it is in the frequency direction, then
h would be calculated using the following formula:

h(i,j)=w1(i,j)h1+w2(i,j)h2+w3(i,j)h3+w.sub-
.4(i,j)h4

[0061] where

[0062] w3(i,j)=0

[0063] w4(i,j)=0 and

[0064] w1(i,j)+w2(i,j)=1

[0065] Alternatively, if the channel coherence is greater in the time
direction than it is in the frequency direction, then h would be
calculated using the following formula:

h(i,j)=w1(i,j)+w2(i,j)h2+w3(i,j)h3+w4(i,j)-
h4

[0066] where

[0067] w1(i,j)=0

[0068] w2(i,j)=0 and

[0069] w3(i,j)+w4(i,j)=1

[0070] In another embodiment, the weights in both directions (time and
frequency) are adaptively changed according to the channel coherence in
the time and frequency directions as follows:

[0076] The method of adaptive interpolation set out above takes advantage
of the fact that interpolated results from the direction of larger
coherence time/frequency is more reliable, and hence is interpolated
first. The calculation of h will effectively increase the density of
pilot symbols in the direction of faster change thereby improving channel
estimation without increasing overhead. As such, the results of the first
interpolating step can then be used to assist the interpolation in the
dimension of smaller coherence time/frequency.

[0077] In general, there are at least three ways to evaluate the channel
change between two pilots, including:

[0078] i. Euclidean distance. One
problem with Euclidean distance, however, is that it is not sensitive to
phase change;

[0079] ii. Phase change. One problem with phase change,
however, is computation complexity; and

[0080] iii. Amplitude change. One
problem with amplitude change, however, is that it is insensitive to
phase change.

[0081] In light of these drawbacks a way to measure channel change so as
to take both amplitude change and phase change into account, while at the
same time keeping the computation complexity to a minimum, is desirable.
According to an embodiment of the invention, therefore, a way of using
the inner products of the two pilot assisted channel estimates as a
measurement of channel change is shown below.

Λfreq=h1,h2=|h1∥h2|cos(θ.-
sub.1,2)

[0082] Λtime denotes channel change in the time direction.

[0083] Λfreq denotes channel change in the frequency
direction.

[0084] The term "<hnhm>" denotes the inner product of
hn and hm.

[0085] The term "|hn|" denotes the magnitude of the vector hn.
If hn=a+bi then |hn|=sqr(a2+b2).

[0086] The term "cos(θ1,2)" denotes the cosine of the difference in
angle between hn and hm:
cos(θn,m)=cos(θn-θm). If hn=a+bi then θn can
be calculated as θn=tan-1(b/a).

[0087] The vector hn can be represented as
h1=|h1|eiθn, or as hn=a+bi, where

a=|hn|cos(θn), and b=|hn|sin(θn).

[0088] When the amplitude changes linearly between the two channel
estimates, the maximum Λ is achieved when |h1|=|h2| in
frequency and |h3|=|h4| in time.

[0089] Hence the more the channel changes, the smaller the Λ,
regardless whether this change is in amplitude or phase. The inner
product is able to solve phase ambiguity. When |θ|>π/2 (which
rarely occurs), cos(θ) becomes negative, and hence smaller. An
inner product may then be computed, which requires two real
multiplications and one real addition, and the result is therefore a real
number.

[0090] Referring again to FIG. 1, what follows is an example of the
adaptive interpolation method.

[0091] Assume:

[0092] h1=0.4423-1.0968i

[0093] h2=-0.0051-0.1484i

[0094] h3=0.1258-0.3413i

[0095] h4=0.3958-0.5883i

[0096] The central point, known from a simulation, has the value of
h=0.2859-0.4224i.

[0097] The inner product is then calculated as follows:

h1h2=0.1605

h3h4=0.2506

where h1h2=denotes the inner product of h1 and h2.

[0098] If h1=a1+ib1 and h2=a2+ib2 then the
inner product can be calculated as

h1h2=a1a2+b1b2.

Alternatively, <h1h2>=|h1|
|h2|cos(θ2-θ1). Since
h1h2<h3h4, the channel changes faster in the
h1/h2 direction. h is then estimated in both the frequency and
time directions:

{tilde over
(h)}h1.sub.,h2=0.5(h1+h2)=0.2186-0.6226i

{tilde over
(h)}h3.sub.,h4=0.5(h3+h4)=0.2608-0.4648i

[0099] Compared with the known h, obviously {tilde over
(h)}h3.sub.,h4 provides a better estimate than {tilde over
(h)}h1.sub.,h2; hence {tilde over
(h)}h3.sub.,h4 can be used to improve the channel
interpolation quality in the h1/h2 direction.

[0100] Note that there is no requirement that h be the middle point
equidistant from h1, h2, h3 and h4.

[0101] In the example above, the interpolation sequence was determined to
be:

[0102] i. interpolate between the two pilots in the time direction
first to calculate h, and

[0103] ii. use h and/or one or both of the two
pilots to interpolate in the frequency direction.

[0104] Of course, if the initial calculation used to determine which
channel direction changes faster determines that the h3/h4
direction changes faster, then the interpolation sequence will be:

[0105] i. interpolate between the two pilots in the frequency direction
first to calculate h, and

[0106] ii. use h and/or one or both of the two
pilots to interpolate in the time direction.

[0107] Once h is calculated, any one of a number of conventional channel
estimation techniques can be used. Such channel estimation techniques
typically consist of two steps. First, the attenuations at the pilot
positions are measured. This measurement is calculated using the formula:

H ( n , k ) ≡ Y ( n , k ) X ( n , k )
##EQU00001##

where X(n,k) is the known pilot symbol, and Y(n,k) is the received pilot
symbol.

[0108] These measurements are then used to estimate (interpolate) the
attenuations of the data symbols in the second step. Persons skilled in
the art will appreciate that such channel estimation techniques include,
but are not limited to, linear interpolation, second order interpolation,
maximum likelihood (least square in time domain), linear minimum square
error and others.

[0109] In another embodiment, a "majority vote" is used to determine the
interpolation sequence for all the "diamonds" across the frequency
domain. This means that there are several calculations performed along
the frequency direction for the channel change. Some results will
indicate there is more change in time, while other results indicate there
is more change in frequency. The "majority vote" option means the choice
whether to interpolate first in the time direction or the frequency
direction is arrived at by assessing the majority of the results. For
example, if the majority of the results indicate that the channel changes
faster in the time direction, then interpolation is first performed in
the frequency direction, and then in the time direction. If the majority
of the results indicate that the channel changes faster in the frequency
direction, then interpolation is first performed in the time direction,
and is then performed in the frequency direction.

[0110] In accordance with an embodiment of the invention, FIG. 3 presents
simulation results for the adaptive interpolation method described above.
The results show the benefit of adaptive interpolation when channel
changes slower in the time direction when UE speed is low, and slower in
the frequency direction when UE speed is high. The curve of "ideal
channel" is of the case with clean known channel, i.e. with no
interpolation loss and additive noise. As shown this approach recoups
most of the interpolation loss. The results were obtained with the
majority vote option described above.

[0111] It is not necessary that there be a regular diamond shaped pilot
pattern in order to use the adaptive interpolation method described
above. For example, an irregular diamond shaped pilot pattern can be used
in accordance with other embodiments of the present invention, such as
the scattered pilot patterns shown in FIGS. 4A to 11. In FIGS. 4A to 11,
the number of OFDM symbols per Transmission Time Interval (TTI) is odd
instead of even. In some embodiments, the scattered pilot patterns can be
generated by more than one antenna such as is shown in FIGS. 5, 7, 9, 10,
11 and 12.

[0112] In general, the adaptive interpolation method works with all
"staggered" pilot patterns which describes all shapes other than a
square, which does not work. A perfect diamond shape, which is the most
favourable shape, is a special case of a staggered pilot pattern. Another
example of a pattern which would work is a "kite" pattern where the
pilots are spread further apart in one direction than the other.

[0113] More generally, in FIGS. 4A to 11, in each sub-frame, pilots are
transmitted by part of the sub-carriers in at least two non-contiguous
OFDM symbols by at least one transmit antenna. The pilot sub-carriers in
the first OFDM symbol and the second OFDM symbol are staggered in the
frequency domain. In FIGS. 5A, 5D, 7A, 7D, 9A and 9D, pilot symbols from
all transmit antennas are transmitted through the same non-contiguous
OFDM symbols. This arrangement will save the terminal power since only
two OFDM symbols are coded to obtain the channel information.

[0114]FIG. 4A is a diagram of a single antenna irregular diamond lattice
scattered pilot pattern which can used in accordance with an embodiment
of the present invention. The overhead associated with this pilot pattern
is 1/28 per antenna. Pilot and data symbols are spread over an OFDM
sub-frame in a time direction 420 and a frequency direction 422. Most
symbols within the OFDM sub-frame are data symbols 424. Pilot symbols 426
are inserted in an irregular diamond lattice pattern. In this embodiment,
an OFDM sub-frame comprises eight sub-carriers 428 and seven OFDM symbols
430.

[0115] As with the scattered pilot pattern in FIG. 1, there is first
performed a calculation of the channel changes in both the time direction
and the frequency direction and a comparison is made as to which
direction the channel changes faster. One-dimensional interpolation is
then performed in the direction with slower channel change.
One-dimensional interpolation is then performed in the direction with
faster channel change.

[0116]FIG. 4B is a diagram of a single antenna irregular diamond lattice
scattered pilot pattern which can used in accordance with an embodiment
of the present invention. Though similar to FIG. 4A, in this case one of
the pilots in each diamond lattice is offset by one OFDM symbol position.
Thus, the adaptive interpolation method does not require that the
scattered pilots line up in either or both of the time direction and the
frequency direction. In the case of staggered pilot patterns where the
pilots do not line up in either the time direction, the frequency
direction, or both, it is more accurate to refer to the "h1/h2
direction" and the "h3/h4 direction" rather than the time
direction and the frequency direction.

[0117]FIG. 5A is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention. The overhead associated with this
scattered pilot pattern is 1/28 per antenna. In this embodiment, an OFDM
frame comprises eight sub-carriers 528 and seven OFDM symbols 530.

[0118] Pilot and data symbols are spread over an OFDM frame in a time
direction 420 and a frequency direction 522. Most symbols within the OFDM
frame are data symbols 524. Pilot symbols 526 are inserted in an
irregular diamond lattice pattern.

[0119] As with the scattered pilot pattern in FIG. 1, there is first
performed a calculation of the channel changes in both the time direction
and the frequency direction and a comparison is made as to which
direction the channel changes faster. One-dimensional interpolation is
then performed in the direction with slower channel change. Using these
measurements, one-dimensional interpolation is then performed in the
direction with faster channel change.

[0120] FIGS. 5B, 5C and 5D are three other examples of scattered pilot
patterns which can be generated according to this embodiment.

[0121]FIG. 6 is a diagram of a single antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0122] FIG. 7A is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0123]FIG. 7B is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0124] FIG. 7C is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0125] FIG. 7D is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0126]FIG. 8 is a diagram of a single antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0127]FIG. 9A is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0128] FIG. 9B is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0129]FIG. 9c is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0130] FIG. 9D is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0131] FIG. 10A is a diagram of a one antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0132] FIG. 10B is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0133] FIG. 11A is a diagram of a one antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0134] FIG. 11B is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0135]FIG. 12A is a diagram of a one antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0136] FIG. 12B is a diagram of a four antenna irregular diamond lattice
scattered pilot pattern which can be used in accordance with an
embodiment of the present invention.

[0137] For the purposes of providing context for embodiments of the
invention for use in a communication system, FIGS. 13-17 will now be
described. As will be described in more detail below, the method of the
present invention can, in one embodiment, be implemented through means of
the channel estimation logic of a conventional OFDM receiver (see channel
estimation 96 in FIG. 17).

[0138]FIG. 13 shows a base station controller (BSC) 10 which controls
wireless communications within multiple cells 12, which cells are served
by corresponding base stations (BS) 14. In general, each base station 14
facilitates communications using OFDM with mobile and/or wireless
terminals 16, which are within the cell 12 associated with the
corresponding base station 14. The movement of the mobile terminals 16 in
relation to the base stations 14 results in significant fluctuation in
channel conditions. As illustrated, the base stations 14 and mobile
terminals 16 may include multiple antennas to provide spatial diversity
for communications.

[0139] A high level overview of the mobile terminals 16 and base stations
14 upon which aspects of the present invention may be implemented is
provided prior to delving into the structural and functional details of
the preferred embodiments. With reference to FIG. 14, a base station 14
is illustrated. The base station 14 generally includes a control system
20, a baseband processor 22, transmit circuitry 24, receive circuitry 26,
multiple antennas 28, and a network interface 30. The receive circuitry
26 receives radio frequency signals bearing information from one or more
remote transmitters provided by mobile terminals 16 (illustrated in FIG.
13). A low noise amplifier and a filter (not shown) may cooperate to
amplify and remove broadband interference from the signal for processing.
Downconversion and digitization circuitry (not shown) will then
downconvert the filtered, received signal to an intermediate or baseband
frequency signal, which is then digitized into one or more digital
streams.

[0140] The baseband processor 22 processes the digitized received signal
to extract the information or data bits conveyed in the received signal.
This processing typically comprises demodulation, decoding, and error
correction operations. As such, the baseband processor 22 is generally
implemented in one or more digital signal processors (DSPs) or
application-specific integrated circuits (ASICs). The received
information is then sent across a wireless network via the network
interface 30 or transmitted to another mobile terminal 16 serviced by the
base station 14.

[0141] On the transmit side, the baseband processor 22 receives digitized
data, which may represent voice, data, or control information, from the
network interface 30 under the control of control system 20, and encodes
the data for transmission. The encoded data is output to the transmit
circuitry 24, where it is modulated by a carrier signal having a desired
transmit frequency or frequencies. A power amplifier (not shown) will
amplify the modulated carrier signal to a level appropriate for
transmission, and deliver the modulated carrier signal to the antennas 28
through a matching network (not shown). Various modulation and processing
techniques available to those skilled in the art are used for signal
transmission between the base station and the mobile terminal.

[0142] With reference to FIG. 15, a mobile terminal 16 configured
according to one embodiment of the present invention is illustrated.
Similarly to the base station 14, the mobile terminal 16 will include a
control system 32, a baseband processor 34, transmit circuitry 36,
receive circuitry 38, multiple antennas 40, and user interface circuitry
42. The receive circuitry 38 receives radio frequency signals bearing
information from one or more base stations 14. A low noise amplifier and
a filter (not shown) may cooperate to amplify and remove broadband
interference from the signal for processing. Downconversion and
digitization circuitry (not shown) will then downconvert the filtered,
received signal to an intermediate or baseband frequency signal, which is
then digitized into one or more digital streams.

[0143] The baseband processor 34 processes the digitized received signal
to extract the information or data bits conveyed in the received signal.
This processing typically comprises demodulation, decoding, and error
correction operations. The baseband processor 34 is generally implemented
in one or more digital signal processors (DSPs) and application specific
integrated circuits (ASICs).

[0144] For transmission, the baseband processor 34 receives digitized
data, which may represent voice, data, or control information, from the
control system 32, which it encodes for transmission. The encoded data is
output to the transmit circuitry 36, where it is used by a modulator to
modulate a carrier signal that is at a desired transmit frequency or
frequencies. A power amplifier (not shown) will amplify the modulated
carrier signal to a level appropriate for transmission, and deliver the
modulated carrier signal to the antennas 40 through a matching network
(not shown). Various modulation and processing techniques available to
those skilled in the art are used for signal transmission between the
mobile terminal and the base station.

[0145] In OFDM modulation, the transmission band is divided into multiple,
orthogonal carrier waves. Each carrier wave is modulated according to the
digital data to be transmitted. Because OFDM divides the transmission
band into multiple carriers, the bandwidth per carrier decreases and the
modulation time per carrier increases. Since the multiple carriers are
transmitted in parallel, the transmission rate for the digital data, or
symbols, on any given carrier is lower than when a single carrier is
used.

[0146] OFDM modulation utilizes the performance of an Inverse Fast Fourier
Transform (IFFT) on the information to be transmitted. For demodulation,
the performance of a Fast Fourier Transform (FFT) on the received signal
recovers the transmitted information. In practice, the IFFT and FFT are
provided by digital signal processing carrying out an Inverse Discrete
Fourier Transform (IDFT) and Discrete Fourier Transform (DFT),
respectively. Accordingly, the characterizing feature of OFDM modulation
is that orthogonal carrier waves are generated for multiple bands within
a transmission channel. The modulated signals are digital signals having
a relatively low transmission rate and capable of staying within their
respective bands. The individual carrier waves are not modulated directly
by the digital signals. Instead, all carrier waves are modulated at once
by IFFT processing.

[0147] In operation, OFDM is preferably used for at least down-link
transmission from the base stations 14 to the mobile terminals 16. Each
base station 14 is equipped with "n" transmit antennas 28, and each
mobile terminal 16 is equipped with "m" receive antennas 40. Notably, the
respective antennas can be used for reception and transmission using
appropriate duplexers or switches and are so labeled only for clarity.

[0148] With reference to FIG. 16, a logical OFDM transmission architecture
will be described. Initially, the base station controller 10 will send
data to be transmitted to various mobile terminals 16 to the base station
14. The base station 14 may use the channel quality indicators (CQIs)
associated with the mobile terminals to schedule the data for
transmission as well as select appropriate coding and modulation for
transmitting the scheduled data. The CQIs may be directly from the mobile
terminals 16 or determined at the base station 14 based on information
provided by the mobile terminals 16. In either case, the CQI for each
mobile terminal 16 is a function of the degree to which the channel
amplitude (or response) varies across the OFDM frequency band.

[0149] Scheduled data 44, which is a stream of bits, is scrambled in a
manner reducing the peak-to-average power ratio associated with the data
using data scrambling logic 46. A cyclic redundancy check (CRC) for the
scrambled data is determined and appended to the scrambled data using CRC
adding logic 48. Next, channel coding is performed using channel encoder
logic 50 to effectively add redundancy to the data to facilitate recovery
and error correction at the mobile terminal 16. Again, the channel coding
for a particular mobile terminal 16 is based on the CQI. In some
implementations, the channel encoder logic 50 uses known Turbo encoding
techniques. The encoded data is then processed by rate matching logic 52
to compensate for the data expansion associated with encoding.

[0150] Bit interleaver logic 54 systematically reorders the bits in the
encoded data to minimize the loss of consecutive data bits. The resultant
data bits are systematically mapped into corresponding symbols depending
on the chosen baseband modulation by mapping logic 56. Preferably,
Quadrature Amplitude Modulation (QAM) or Quadrature Phase Shift Key
(QPSK) modulation is used. The degree of modulation is preferably chosen
based on the CQI for the particular mobile terminal. The symbols may be
systematically reordered to further bolster the immunity of the
transmitted signal to periodic data loss caused by frequency selective
fading using symbol interleaver logic 58.

[0151] At this point, groups of bits have been mapped into symbols
representing locations in an amplitude and phase constellation. When
spatial diversity is desired, blocks of symbols are then processed by
space-time block code (STC) encoder logic 60, which modifies the symbols
in a fashion making the transmitted signals more resistant to
interference and more readily decoded at a mobile terminal 16. The STC
encoder logic 60 will process the incoming symbols and provide "n"
outputs corresponding to the number of transmit antennas 28 for the base
station 14. The control system 20 and/or baseband processor 22 as
described above with respect to FIG. 14 will provide a mapping control
signal to control STC encoding. At this point, assume the symbols for the
"n" outputs are representative of the data to be transmitted and capable
of being recovered by the mobile terminal 16.

[0152] For the present example, assume the base station 14 has two
antennas 28 (n=2) and the STC encoder logic 60 provides two output
streams of symbols. Accordingly, each of the symbol streams output by the
STC encoder logic 60 is sent to a corresponding IFFT processor 62,
illustrated separately for ease of understanding. Those skilled in the
art will recognize that one or more processors may be used to provide
such digital signal processing, alone or in combination with other
processing described herein. The IFFT processors 62 will preferably
operate on the respective symbols to provide an inverse Fourier
Transform. The output of the IFFT processors 62 provides symbols in the
time domain. The time domain symbols are grouped into frames, which are
associated with a prefix by prefix insertion logic 64. Each of the
resultant signals is up-converted in the digital domain to an
intermediate frequency and converted to an analog signal via the
corresponding digital up-conversion (DUC) and digital-to-analog (D/A)
conversion circuitry 66. The resultant (analog) signals are then
simultaneously modulated at the desired RF frequency, amplified, and
transmitted via the RF circuitry 68 and antennas 28. Notably, pilot
signals known by the intended mobile terminal 16 are scattered among the
sub-carriers. The mobile terminal 16, which is discussed in detail below,
will use the pilot signals for channel estimation.

[0153] Reference is now made to FIG. 17 to illustrate reception of the
transmitted signals by a mobile terminal 16. Upon arrival of the
transmitted signals at each of the antennas 40 of the mobile terminal 16,
the respective signals are demodulated and amplified by corresponding RF
circuitry 70. For the sake of conciseness and clarity, only one of the
two receive paths is described and illustrated in detail.
Analog-to-digital (A/D) converter and down-conversion circuitry 72
digitizes and downconverts the analog signal for digital processing. The
resultant digitized signal may be used by automatic gain control
circuitry (AGC) 74 to control the gain of the amplifiers in the RF
circuitry 70 based on the received signal level.

[0154] Initially, the digitized signal is provided to synchronization
logic 76, which includes coarse synchronization logic 78, which buffers
several OFDM symbols and calculates an auto-correlation between the two
successive OFDM symbols. A resultant time index corresponding to the
maximum of the correlation result determines a fine synchronization
search window, which is used by fine synchronization logic 80 to
determine a precise framing starting position based on the headers. The
output of the fine synchronization logic 80 facilitates frame acquisition
by frame alignment logic 84. Proper framing alignment is important so
that subsequent FFT processing provides an accurate conversion from the
time domain to the frequency domain. The fine synchronization algorithm
is based on the correlation between the received pilot signals carried by
the headers and a local copy of the known pilot data. Once frame
alignment acquisition occurs, the prefix of the OFDM symbol is removed
with prefix removal logic 86 and resultant samples are sent to frequency
offset correction logic 88, which compensates for the system frequency
offset caused by the unmatched local oscillators in the transmitter and
the receiver. Preferably, the synchronization logic 76 includes frequency
offset and clock estimation logic 82, which is based on the headers to
help estimate such effects on the transmitted signal and provide those
estimations to the correction logic 88 to properly process OFDM symbols.

[0155] At this point, the OFDM symbols in the time domain are ready for
conversion to the frequency domain using FFT processing logic 90. The
results are frequency domain symbols, which are sent to processing logic
92. The processing logic 92 extracts the scattered pilot signal using
scattered pilot extraction logic 94, determines a channel estimate based
on the extracted pilot signal using channel estimation logic 96, and
provides channel responses for all sub-carriers using channel
reconstruction logic 98. In order to determine a channel response for
each of the sub-carriers, the pilot signal is essentially multiple pilot
symbols that are scattered among the data symbols throughout the OFDM
sub-carriers in a known pattern in both time and frequency. Examples of
scattering of pilot symbols among available sub-carriers over a given
time and frequency plot in an OFDM environment are found in PCT Patent
Application No. PCT/CA2005/000387 filed Mar. 15, 2005 assigned to the
same assignee of the present application. Continuing with FIG. 17, the
processing logic compares the received pilot symbols with the pilot
symbols that are expected in certain sub-carriers at certain times to
determine a channel response for the sub-carriers in which pilot symbols
were transmitted. The results are interpolated to estimate a channel
response for most, if not all, of the remaining sub-carriers for which
pilot symbols were not provided. The actual and interpolated channel
responses are used to estimate an overall channel response, which
includes the channel responses for most, if not all, of the sub-carriers
in the OFDM channel.

[0156] The frequency domain symbols and channel reconstruction
information, which are derived from the channel responses for each
receive path are provided to an STC decoder 100, which provides STC
decoding on both received paths to recover the transmitted symbols. The
channel reconstruction information provides equalization information to
the STC decoder 100 sufficient to remove the effects of the transmission
channel when processing the respective frequency domain symbols.

[0157] The recovered symbols are placed back in order using symbol
de-interleaver logic 102, which corresponds to the symbol interleaver
logic 58 of the transmitter. The de-interleaved symbols are then
demodulated or de-mapped to a corresponding bitstream using de-mapping
logic 104. The bits are then de-interleaved using bit de-interleaver
logic 106, which corresponds to the bit interleaver logic 54 of the
transmitter architecture. The de-interleaved bits are then processed by
rate de-matching logic 108 and presented to channel decoder logic 110 to
recover the initially scrambled data and the CRC checksum. Accordingly,
CRC logic 112 removes the CRC checksum, checks the scrambled data in
traditional fashion, and provides it to the de-scrambling logic 114 for
de-scrambling using the known base station de-scrambling code to recover
the originally transmitted data 116.

[0158] In parallel to recovering the data 116, a CQI, or at least
information sufficient to create a CQI at the base station 14, is
determined and transmitted to the base station 14. As noted above, the
CQI may be a function of the carrier-to-interference ratio (CR), as well
as the degree to which the channel response varies across the various
sub-carriers in the OFDM frequency band. The channel gain for each
sub-carrier in the OFDM frequency band being used to transmit information
is compared relative to one another to determine the degree to which the
channel gain varies across the OFDM frequency band. Although numerous
techniques are available to measure the degree of variation, one
technique is to calculate the standard deviation of the channel gain for
each sub-carrier throughout the OFDM frequency band being used to
transmit data.

[0159]FIG. 18 is a block diagram of one embodiment of the present
invention. In this embodiment, the present invention is shown being
implemented within channel estimation logic 96 of FIG. 17 with the
conventional aspects of channel estimation logic 96 being shown in dotted
outline for ease of reference. Persons skilled in the art will appreciate
that the present invention could be implemented as a separate logical
component as well.

[0160] Shown is time direction channel calculator 127 which performs the
calculation of channel change in the time direction. Frequency direction
channel calculator 129 performs the calculation of channel change in the
frequency direction. As explained above, the preferred calculation is the
inner product of the two pilot assisted channel estimates being compared.
Though time direction channel calculator 127 is shown as being
illustrated to the right of frequency direction channel calculator 129,
this does not mean that the time direction channel calculation is
necessarily to be performed first or that the calculations cannot be
performed simultaneously. Either calculation can be performed first, or
both can be performed simultaneously. Channel direction comparator 131
compares the results of the calculations performed by both direction
channel calculator 127 and frequency direction channel calculator 129 for
the purpose of comparing and ascertaining which channel direction, time
or frequency, changes slower. Channel direction selector 133 selects
which of the two directions changes slower. Block 135 is utilized to
interpolate, first in the direction of slower change, and then in the
direction of faster change, in accordance with conventional means.

[0161] In operation, time direction channel calculator 127 receives two
pilot assisted channel estimates and performs the calculation of channel
change in the time direction. Frequency direction channel calculator 129
performs the calculation of channel change in the frequency direction
though these two calculations can be performed in different order or
simultaneously. Channel direction comparator 131 compares the results of
the calculations performed by both direction channel calculator 127 and
frequency direction channel calculator 129 and compares which channel
direction, time or frequency, changes slower. Channel direction selector
133 selects the direction of slower change and interpolation is then
performed by block 135 in that direction first, and then in the direction
of faster change in accordance with conventional means.

[0162] FIGS. 13 to 18 each provide a specific example of a communication
system or elements of a communication system that could be used to
implement embodiments of the invention. It is to be understood that
embodiments of the invention can be implemented with communications
systems having architectures that are different than the specific
example, but that operate in a manner consistent with the implementation
of the embodiments as described herein.

[0163] Numerous modifications and variations of the present invention are
possible in light of the above teachings. It is therefore to be
understood that within the scope of the appended claims, the invention
may be practised otherwise than as specifically described herein.

Patent applications by Dong-Sheng Yu, Ottawa CA

Patent applications by Hang Zhang, Nepean CA

Patent applications by Hua Xu, Nepean CA

Patent applications by Jianglei Ma, Kanata CA

Patent applications by Ming Jia, Ottawa CA

Patent applications by Peiying Zhu, Kanata CA

Patent applications by Wen Tong, Ottawa CA

Patent applications in class Plural channels for transmission of a single pulse train

Patent applications in all subclasses Plural channels for transmission of a single pulse train