4.6 Automotive EMI Reduction Techniques, Applications, and Solutions

Welcome to this TI training session. The slide set is about automotive EMI, reduction techniques, applications, and solutions. The presentation is put together by myself, Robert Loke-- I'm a member group technical staff located in Santa Clara-- and my colleague, Robert Blattner. He's a system and applications engineer in Santa Clara as well
In the next years, we will see a lot more electronic content going into cars. And in this training, you will learn about some of these new automotive application trends. You will learn about EMI noise sources, and you will hear a lot about E-Field and the E-Field coupling. We will introduce you to some EMI mitigation techniques, especially switch node shaping and E-Field shielding techniques. And we will show you results with EMI measurements and help you to pass CISPR 25 Class 5.
Here is a detailed agenda on this training. First, we talk about ADAS and infotainment applications. And there are some new trend, the change from 400 kilohertz to 2.1 megahertz switching. Some of the switch node noise challenges we will have out of this.
Then, Robert has put together a really nice noise source model for buck regulators. And he will go into detail how the conducted EMI is essential for passing the radiating test. And he has nice details about the differential and common mode noise sources and the reduction techniques.
At the third part, at the end, we will show some results with specific filters like 2-stage filters, filters with common mode choke 3-stage filters. And a lot of E-Field shielding experiments, because we think it's one of the most powerful techniques available.
One of the most fascinating trends in automotive is ADA systems-- Advanced driver-assistance systems. We will get a lot more sensors in the car and electronic systems to process and fuse these datas. Like high speed image cameras, they able will be connected over high speed data links. And with this, we'll see a lot of high speed clocks. All these are a potential threat to EMI.
24 gigahertz and 27 gigahertz radars will introduce new frequency in the whole frequency spectrums and probably give us new limits. Then, with all these new electronic systems, we will get a much more complex wire harness. And we know from previous training sessions that every wire is a potential antenna, or can couple capacitive to other wires. So we are worrying a lot about the signal to noise ratio. So the more electronic systems and noise systems coming into the car, the higher the noise flow is. And the higher the noise flow is, the lower the signal to noise ratio. And to get good data rate, you need high signal to noise ratio. Otherwise, the data is all intermittent or getting slower and slower. And one of the biggest changes, all these are now safety relevant. So autonomous driving requires best in class EMI.
For ADA systems, we will need a lot more camera modules in the car. And to make these camera modules more cost efficient and smaller, a lot of things happened in the past. So for example, the metal enclosure changed to cheaper and lighter plastics. But without the metal enclosure, we have less EMI filtering. Then, these modules, they are forced to get smaller and smaller to better integrate them in the car. And to get this whole solution size smaller, we have a solution for this by using ICs that need smaller EMI filters to pass CISPR. So the solution is go to higher frequencies, like 2.1 megahertz, add new switch node shaping techniques, like spread spectrum to it, and you will have a much smaller EMI filter where you may just need a small ferrite bead.
Another camera category comes. These are smart mirrors and front cameras. So in the existing enclosure, like the rear view mirror, cameras will be integrated together with processing. So we have the potential problem of near field coupling of noise between the clocks and the display wires, or the processor clocks. And the frequency of these processors are going up, and there are multiple clocks in the systems. So we'll have, along the whole frequency spectrum, potential threats. And all those change is set-- all the analog links changing basically to high speed digital links, which introduce new frequencies and new modulations in the EMI field.
Another big change we see in automotive on the infotainment system. The infotainment system has the classic head unit. Then, it has the instrument cluster. Then, the whole audio system is part of it. And then, these new USB power media interfaces everywhere in the car.
The infotainment system has multiple EMI challenges. First, we have such a complex wire harness here of different power cables, display links, speaker cables, USB power or links, RF antennas-- different ones-- and then the automotive CAN bus systems. They all operate on different frequencies and have their own requirements. But the biggest threat is that they can interfere with each other.
So when we look in a classic head unit, we have multiple radio frequency modules-- Bluetooth, LTE, FM, satellite, and GPS. And based on these RF modules, we have certain frequency ranges we need to protect, and need to stay below certain limits. Then, the processor speed went up with new high speed graphic chips and multiple clocks. All these frequencies basically can interfere potentially into these radio frequencies. High resolution displays are introduced now with much higher resolution frequency. Small motors you will find in lower end instrument clusters, for example. And these motors can interfere with other power lines or signal lines.
A new category is eCall systems, or telematic systems, which basically are safety relevant in case of an emergency. A lot of output wires going to the speakers, they are now class D, and they're all modulated. We have new output wires to USB, not only at the head unit, but now in the second row as well, so there are longer wires with it. And we never know what really is on these USB power cables. Different media devices-- phones, tablets, and so on-- can be connected there. And the USB cable is a potential antenna here, and a potential threat for EMI.
Probably one of the biggest worries of the car manufacturers are potential interferences from these external media devices. If they coming close, for example, to an instrument cluster, and that may interfere other information there, or have any kind of disturbance, which is necessary for the driving.
So with every change in these end equipments, you have to ask certain questions, which are EMI relevant. The first important question is-- what is the input power source of this end equipment? Does it come from a battery? What voltage level is-- 12 volt, 24, or 48 volt standard? Or does it come from a generator, or is there any AC components from an AC generator coming.
The next question is-- what kind of cables are going into this end equipment? What type? What gauge? Stranded wire? Is it twisted pair or shielded? They all have a very important relevance to the EMI radiation ladle. And the coupling, potential coupling, between different cables, the total length of the wire is clearly a big question, just because of Lambda from the frequency. And what kind of signals are going over these wires? So the input power source, the cables, and then, of course, what kind of enclosure and connectors are used? Are they EMI shielded? Is there anything which can get out of these connectors, or the enclosure? Never forget to ask, where is the load? Mostly, it's internal, like if there's a processor and a box. But sometimes the load is external, like, for example, in a USB charger.
So each end application has its own EMI challenges and will need its own solutions. We can break them down by PCB and layout, schematic, the use components, and, of course, the kind of ICs or chips used in the solution. When we look at the PCB and the layout, the PCB is mostly given by the end equipment solution size. So a very large PCB will distribute electrical fields much more than a very small board. The number of layers and the stack-up is important. The component placement, the routing of noisy signals, the kind of shielding used, like via stitching. And the position of the connectors is mostly given by the end equipment application as well. So PCB, layout, and the component placement is definitely EMI critical.
One of the biggest influences we have is on the schematic level. What kind of EMI filter we will use. What kind of switch node shaping we can use, like snubber. What kind of noise filter we use on the output. Can we slow down the switch node a little bit with Rboot? What kind of input and output caps? One of the biggest trends we see is the change that we switch in power regulators from 400 kilohertz to 2.1 megahertz. And this is mostly to get a smaller solution size.
There are a lot of different components which perform differently under EMI criteria. There are special metal shield EMI inductors. The use of common mode chokes has a big influence as well. Shield, ferrite beads, caps. They use cables and the connectors, like mentioned before. So the trend-- the overall trend-- on the component layer is basically to get a lower BOM cost.
The IC topology plays a major role in EMI. In the days where we used LDOs, we had much less EMI issues, versus the majority now with more efficient buck regulators. Controllers will behave different than integrated converters. Non-synchronous buck will definitely behave different than synchronous bucks with integrated FETs. Multiphase buck regulators is a very nice technology as well, which we will see more in the future as well, and will reduce ripple noise, mostly in the fundamental. So using an EMI optimized power architecture will be key for the future to pass all the necessary requirements.
One of the major advantages 2.1 megahertz will bring is a smaller solution size. You see here two EVMs side by side, one with 400 kilohertz, the other one with 2.1 megahertz. You can clearly see that the 400 kilohertz solution needs a larger output cap and has a larger inductor. 2.1 megahertz reduces the capacitance by a factor of 5 and inductance by a factor of 5, only if you want to have the same ripple noise. In most applications where they use 400 kilohertz, they reduce Cout, and L as well, with a buy-in of a little bit larger ripple noise. But overall, by theory, 2.1 megahertz will have a certain advantage on lower output capacitance, input capacitance, and used inductance.
This is especially true for a small size solution, like here in the case of the LM53601-Q1, which is a 2.1 megahertz one amp solution. This is mostly used in small camera modules, where every component counts on the filter side, or on the input cap and output cap sides.
2.1 megahertz not only brings it advantages, it has disadvantages as well. And this one, we can see in the harmonics of the frequency spectrum. To see this better, I show here a 400 kilohertz sinusoidal FFT of a standard sinus frequency. As we know and expect, there's just one fundamental here at the 400 kilohertz, and nothing else. No harmonics here.
When we change the shape from sinusoidal to a switch node shape with a certain slew rate, we will see that first harmonics showing up. And because of this slew rate, we will see a 40 dB per decade degradation of the harmonic power level. If we make this switching sharper with faster slew rates, because we want to have a higher efficiency, for example, then we can see that this maximum rise time we will get a minus 20 dB per decade line. So the harmonics will be much higher frequency spectrum. This is 400 kilohertz. And with infinite rise times, we will easily get a lot of dBs above 100 megahertz.
With a slew rate control, we basically can make this more trapezoidal. So with a slower slew rate, these 40 dB line per decade will be shifted to the left, so we will get less harmonics and the higher frequency. If you make the slew rate faster, we basically shift the minus 40 dB per decade line to the right here, and we will get more higher frequency harmonics.
A 400 kilohertz square signal already introduces a lot of harmonics in much higher frequency than the switching frequency itself. If we now switch from 400 kilohertz to 2 megahertz to get a smaller solution size, we will shift basically the 20 dB curve to the right and, as a result, we will get much higher harmonic content in much higher frequencies, just by switching from 400 kilohertz to 2 megahertz. The slew rate control is similar to the 400 kilohertz. The slower the slew rate on a 2 megahertz switching is, the more this minus 40 dB line goes up to the left in the lower frequencies, and its cutting off earlier.
Of course, there are limits, because 2 megahertz is naturally a much lower efficiency than 400 kilohertz switching. We will need very fast rise times to get a higher efficiency. So you will see this arrow here and the 40 dB line moving to the right to get higher efficiencies. But this will introduce a much higher EMI content here in higher frequency, which will need additional filtering later. 2 megahertz switching is calling much higher energy levels in the higher frequencies. One of the techniques to get this down is using spread spectrum. You see here, the result of modulating a 2 megahertz signal is that the band goes down by minus 15 dB in the high frequency range, in the FM range.
To check this out on a scope is very simple. You just probe the switch node. You trigger on a rising edge. You bring the scope in persistence display mode. And then, you look at the following rising edge, how wide this frequency spread is here. On FFT, you see this frequency spread as well. And you will notice that the amplitude of the fundamental-- the energy level-- is not going down by a lot. Spread spectrum is mostly effective in higher frequency bands, and the harmonics-- the needles, basically-- they're getting washed out and schmoozed into the noise flow. This is one of the little disadvantages of spread spectrum, that the low frequency content noise flow always goes up with spread spectrum technologies.
We have shown you that, just by looking at the switch node, how the energy field will look like. I will give now over to Robert Blattner, who will show you how these frequencies will couple in automotive and equipment.
In order to understand noise reduction techniques, a basic model for EMI noise is needed. The model used here includes input noise, voltage ripple conducted from the EUT to its power source, noise coupled off the switch node, and output noise, which is voltage ripple emanating from the EUT and conducted down output wires. Also shown are an input filter-- more about this filter later-- a LISN, also known as an artificial network used for testing, and a ground plane, which simulates a car's metal chassis.
The LISN's job is to allow DC power into the EUT's harness, while stripping out RF noise emanating from the EUT and directing it to RF measurement equipment. The specified RF load is 50 ohms. Most equipment is configured for 50 ohm input. And CISPR 25 artificial networks use 5 microhenries to block RF noise, since 5 microhenries is in the range expected for an input harness. This is different from CISPR 22, non-automotive testing, which utilizes 50 microhenry.
There are two CISPR 25 tests which will be covered in this section. Conducted testing, which measures input ripple on the harness and has limits stated in dB microvolt. Note that the harness is short, allowing a true indication of ripple near the EUT. The second is radiated testing, which measures electric field near the EUT with its harness and has limits stated in dB microvolts per meter. Even though this test is called radiated, it is not a far field test over much of the spectrum over which it is conducted. For this test, the harness is long, allowing it to act as an antenna. This simulates a typical application in a car.
More about conducted EMI testing. On the right is a typical bench setup at the factory for measuring conducted emissions. Shown are the EUT, a load-- in this case, a resistor-- the ANs-- in this case, two separate boxes are used for positive and ground return for the EVM-- a ground plane 5 centimeters below the EVM and harness, and a power supply. If this setup were at a qualified laboratory, the power supply would be a battery, and the setup would be well calibrated. In addition, the conductive table would be more than a copper sheet on a regular table.
Both pictures show a radiated measurement setup at a certified laboratory. The specific setups shown are biconical antenna with horizontal orientation, and a horn antenna with vertical orientation. The full suite of setups include monopole antenna with vertical orientation, along with biconical log and horn antenna, oriented both horizontally and vertically. Note that, other than the horn antenna, which is pointed at the EUT, antennas are centered on the harness. This being a certified setup, a battery is used as a power source. And the harness is longer than the distance between the harness and the antenna.
The diagram on the right shows distances in a CISPR radiated setup. When looking at these distances and the frequencies measure, it can be seen that, over most frequencies of interest when measuring buck converters, this is not actually a measurement of radiation from the setup. Much of what is measured is near field. Below 20 megahertz, this setup can be considered strictly near field, and the harness can be considered short. That means that it can be considered a lump sum element. Above 600 megahertz, most of what is measured can be considered radiated. Between these two frequencies, much of the coupling is near field, but the harness can no longer be considered a lump sum element, and can act as an effective antenna.
It is common knowledge that, if an EUT fails conducted emissions testing, it will typically fail radiated testing as well. The next few slides show a quick calculation, which shows that input filters are needed for almost all bucks. Since low frequencies are covered by this calculation, this calculation is being treated as an electrostatics problem. The first step is to calculate the charge in the harness. In this part of the calculation, the harness is approximated by an infinitely long cylinder. The actual procedure involves use of cylindrical coordinates. The equation and typical results are shown below.
The second step is to adjust the charge found in the first step, to account for harness details. There are actually two wires of finite separation in the harness. This increases the capacitance of the harness as a whole. Note that, if the wires are widely separated and, therefore, independent capacitors, the increase in charge is a factor of 2, which is equivalent to increasing the cylinder's diameter by a factor of 2.7. Typically, diameter should be increased by a factor of 1.2 to 1.6. Both wires are not at the same potential.
If the signal emanating from the EUT is only on the input wire, and none of the signal is on the ground wire, at distance, the harness voltage will appear to be cut in half. This is called unbalanced loading. Note that common mode noise typically dominates above 20 megahertz if the buck is not shielded. In the case of common mode noise, both wires have the same AC potential, so there should be no adjustment for wires of different potential.
The final step to our quick calculation is to find the vertical electric field at the antenna. The spot chosen is in the same plane as the conductor table to prevent the induced charge on the conductive table from changing our calculation. While detailed calculation is not shown on this slide, we can come to the conclusion that the electric field measured in volts per meter is 35 to 50 dB below the conducted voltage on the harness. We can conclude that millivolt level input ripple should be avoided, since it will cause radiated testing to fail, as well as conducted testing to fail.
More detail concerning conducted noise. There are two main types of conducted noise. One is differential mode noise, where the return path is through the ground wire of the harness. The second type of noise is common mode noise. This requires a return path that's not part of the harness. Much of the differential noise emanating from an EUT comes from the input of a buck. The buck can be seen as a current mode waveform generator. This is because we have an inductor and a switch node. The switch node has a high voltage on it. This determines the shape of the current in the inductor. Input ripple and output ripple are small. So the shape of the inductor current waveform determines the shape of the input current waveform.
Once we have the input current waveform for the buck, we can determine the input current waveform for the input capacitor by considering the harness to be a current source. Once we have the input current waveform for the capacitor, we can calculate input voltage ripple, which, for many cases, can be found using this equation. For example, for a 2.1 megahertz buck converting 3.5 volts to 5 volts, and producing 3 amps output, with an 0.75 amp inductor ripple, if we use a 10 microfarad capacitor with 4 milliohm ESR, we'll get about 48 millivolts of input ripple. This far exceeds conducted limits and will cause failure of radiated testing as well.
Since input differential noise emanating from a buck far exceeds the levels needed to pass both conducted and radiated EMI testing, methods must be found to reduce this noise. The first one is simply to increase the input capacitance. This reduces input ripple. In addition, we recommend using a small high frequency cap immediately adjacent to the buck to eliminate high frequency noise. The second method is to insert a filter between the input harness and the buck, which eliminates voltage ripple from voltage found on the input harness.
Step 1 to designing a input filter is to calculate the amount of filtering needed. There are two methods. One is to measure, using an oscilloscope, the input ripple of your buck, and then determining how much filtering is needed. The second is to estimate it using the current waveform and applying the following equation. Once the amount of input ripple reduction that is needed is determined, the filter can be designed. We recommend looking up details in AN-2162.
Next, we cover common mode noise. Common mode noise is typically dominant above 20 megahertz, and is much harder to deal with if you can't use shielding. Typically, with a buck, it comes from capacitive coupling off of the switch node. As a model for calculating how big this ripple should be, we can consider the switch node capacitance to the environment, and the LISN can be considered a 25 ohm resistor. Also note that, since the return path is not through the harness itself, an input EMI filter, such as a PI filter, will have little effect on this noise.
In order to show capacitive coupling off the switch node in action, we show an E-Field scan of an EVM. In this scan, you can clearly see that the vast majority of electric-field is coming from the switch node, including its inductor. The first two methods for reducing common mode noise that we cover involve the switch node itself. The first is to reduce the RF component of the waveform on the switch node using either waveform shaping, which reduces the slew rate on the switch node, or reducing frequency, which reduces the number of edges in a given time on the switch node waveform.
The second method is to reduce the capacitance of the switch node, thereby reducing coupling of the switch node to the environment. This can be done by reducing the physical size of the switch node on the PCB, reducing the physical size of the inductor, which is part of the switch node. Also, align pin 1 of the inductor to the switch node, rather than to output voltage, since some inductors have internal shielding. Finally, avoid routing the switch node on the back of the board, which will couple to the table more effectively than the top of the board.
The next two methods for common mode noise reduction involve changes at the system level. The first is to interrupt coupling of the switch node to the environment using a shield. The shield should be well grounded, so the current captured by the shield is returned to the ground plane, thereby interrupting the circuit. The final method for reducing common mode noise that we cover is inserting a common mode choke in series with the input harness. This breaks the circuit that starts at the switch node, goes through the switch node's capacitance, returns to the table through the LISN, and back through the harness. A disadvantage of the use of common mode chokes is that we've only interrupted the input power connections. There are often other signals which must also be interrupted to prevent common mode noise from becoming a problem.
When considering common mode noise on the input harness, other connections to the EUT must be considered. For example, often voltages are generated across the ground plane. This will show up as common mode noise. In addition, any load, or load simulator, connected to the EUT can capacitively couple to the environment, also introducing common mode noise.
Now, we want to show you some practical examples. On this slide, you see a typical 3-stage filter. At the beginning, at the front, you see the common mode choke. It's usually to suppress the higher frequency noise, or the E-Field it generated noise. Then comes a PI filter, which is a low-pass filter, with CF1, LF1, and CF2. This is mostly to suppress the fundamental, the switch frequency, and the first harmonics. And then, you see a second PI ferrite bead filter here-- CF2, CHF2, and the ferrite bead, which suppress high frequency noise. At the end here is a bulk cap. Sometimes we have a bigger bulk cap on the input side as well.
The 3-stage filter is a very powerful input filter for buck regulators. In this case here, we just used a 2-stage filter, so without the common mode choke. So we can see nicely, this is a 2.1 megahertz 3 amp buck regulator LM53603. And we are supposed to see here a 2.1 megahertz peak of the fundamental. So we can say this filter is very effective-- the low frequency filter is very effective. So the PI is doing a really, really good job here to suppress the fundamental, the second harmonic and third harmonics here. So this whole frequency range, from 150 kilohertz to 30 megahertz, looks very clean, and is way below all the average limits.
On a high frequency, from 30 megahertz to 108 megahertz-- FM range-- we see a lot of needles and harmonics. Especially here in the upper FM range, we see that the average needle lines going way above the average limit. This is the average class 5 limit from CISPR 25, and it's here violated by at least 5 to 8 dB in this frequency range. So there is more filtering needed to pass CISPR 25 class 5 with this kind of configuration.
Now, we are adding a common mode choke to make a 3-stage filter out of this. And what we can nicely see, here, our problem area in the high frequency, went completely down. The whole range, from 80 megahertz to 108 megahertz, is completely pushed down basically by the common mode choke. And it doesn't allow any current flowing back there. So we get a nice 10 to 15 dB margin in this frequency range. And we double check, on the low frequency side, that we don't get any other additional resonances from the common mode choke, also any kind of oscillation. So everything looks nice and clean and down here. So this is a really good solution for this 3 amp 2.1 megahertz buck regulator.
If a common mode choke cannot be used for whatever reason, then there are other alternatives available. Here, we show that we shielded the area with a metal enclosure around the IC, which will shorten directly the electrical field of the switch node close by here. So there will be no current flowing basically out of this system. What you can see is the metal shielding has a very similar effect, if not a greater effect, on the high frequency noise here, like the common mode choke. The common mode choke basically is blocking it to get back in the measurement, and the metal shield is basically closing the E-Field near the switch node, so it cannot be distributed in the system at all.
Another alternative is to use pseudo random spread spectrum. That's a technology which is built in the buck regulator, or can be provided over a synchronized input signal. On the left side, you see a scope shot. And your [? G's ?] kind of looks like a [? jittle ?] on the rising edge. So if you bring your scope in a persistent mode, a display mode, then you can see how wide the frequency is spread. And you can see sometimes a kind of modulation here. In our case, we see five codes every 7.5 nanoseconds, with a spread of a 30 nanoseconds. On FFT, or spectrum analysis, you will see, without spread spectrum, the sharp needles and, with spread spectrum, you would see them wider, and you would see some kind of low frequency modulation, which is an artifact of the spread spectrum modulation.
The second harmonic, you see it's already significantly reduced in third harmonics. And if you go to the higher harmonics, you can see that the noise flow is moved up on the blue one with spread spectrum, and that all the needles are basically gone on the red one. This one blue needle here is an artifact from the how it was recorded. This was the internal CPU frequency of the analog digital converter used here. So it nothing to do with spread spectrum.
So spread spectrum biggest advantage is that all the needles in the high frequency are getting eliminated by shifting up the noise flow a little bit. But in this case here, you see a 1 amp 2.1 megahertz buck regulator with spread spectrum, and you can see that we get a nice 10 to 15 dB margin in the FM range. On the low frequency here, you see that the switch node frequency, the fundamental is very strong. And the main reason is that, in this case, it was just used a ferrite bead filter to make this total solution size extremely small.
So in this case, there's no low frequency PI filter with a wire wound inductor needed, or a common mode choke. You get the same effect, like a common mode choke, here on a high frequency, and the limit here, on the low frequency, is below the class 5 limit. So it will pass a class CISPR 25 class 5. On the negative side, you get here some low frequency artifacts. This comes from the modulation scheme of the used spread spectrum.
An alternative method is E-Field shielding. So we used the same IC with the same board, and turned off the spread spectrum modulation. But we added a metal shield over the solution, so that the electrical field is shorted here to the ground immediately. What you see on the high frequency is beautiful. You basically get a signal which is close to the noise flow. It's not elevated like in a spread spectrum. So you get the largest margin possible here. On the same side, you don't have the modulation artifact anymore from spread spectrum, so you get the lowest possible noise flow. So everything looks really nice and clean here in this whole spectrum, and you even see that the fundamental went a little bit down and gives us a little bit more margin to the average line.
So we wanted to know how E-Field shielding effects on other ICs. And we just choose randomly here another wide [? VIN ?] buck module, which has a switch integrated and inductor integrated already. So what we checked was basically-- we used this module. We put a low frequency PI filter here. Then, we added a metal frame around the solution. And then, we closed the metal with a metal lid dish to have a full E-Field shielding. So it's interesting now how the effect of EMI will be for any generic buck regulator.
When we measure the module with off the shelf components, we can see here all the high frequency noise. We can see in the mid-range, there's a little bit, but not much. The low frequency is very good filtered here. We can see that the fundamental is really nice push down from the low pass PI filter. Then, we added the metal frame. And if we compare these two solutions, and I switch back and forth between the two, you can see how the whole noise flow is coming down, and all the needles. And we're getting at least 5 to 10 dB already, just by putting a metal frame around the solution here. So this will weaken the electrical field here, or partial short at the input side.
When we, then, close the lid, the E-Field is completely shorted within this module. And what you will see is that all the harmonics are gone here. They are basically not detectable with our spectrum analyzer we have in the EMI chamber. And the harmonics are below our detectable noise flow we have. And it has a nice effect on the medium frequencies here as well. All these medium frequency needles are gone. If we compare to the original one, we can clearly see that E-Field shielding is one of the most effective EMI mitigation technique.
With this statement, this training is finished here. And if you are interested to learn more about EMI, then just search under the keyword EMI or automotive EMI on ti.com. Thanks for listening.

Details

Date:
April 16, 2018

Because of the potential havoc that interference can wreak in radio and safety critical systems, automotive electronics are subject to the most stringent EMI standards. In this training, we discuss EMI reduction techniques for increasingly demanding automotive systems like ADAS, cameras, and infotainment.