el%3A20001000

код для вставки на сайт или в блог

ссылки на документ

Fig. 4 shows the contour map of the obtained DTM superimposed to the RCS map evaluated at the points of the same DTM.
The RCS map was calibrated with respect to the image signature
relative to the trihedral corner reflector, the RCS of which is
known. The figures in the labels are relative to the XJ plan. The
corner reflector and the lamp pole are evident.
A conventional optical instrumentation scheme for terrain elevation mapping was used to measure 13 points of the area. By
interpolation, we obtained the DTM shown in Fig. 5. The standard deviation of the elevation difference between the radar measured and the optically measured DTMs, is 0.17m.
20dBm
Transmitted power (Po)
Polarisation
Central frequency U,)
Band (B)
Frequency number (n,)
Linear scan length (L)
Linear scan point number (n,)
Difference between the two antenna heights (b)
Radar elevation over the scenario (12)
vv
5.7GHz
0.6GHz
20 1
0.631n
22
7Scm
18m
Conclusion: We have proposed a radar technique for rapid remote
topographic mapping. The technique could be used both for mapping inaccessible terrain such as active volcanoes or mine fields,
and for monitoring unstable terrain such as subsiding grounds or
landscapes.
The difference between the two antenna heights (ii) has been set
in such a way as to avoid phase ambiguity in the interferogram. In
particular, we used the following condition obtained from eqn. 2:
(3)
where R,,,,is the ininiinuin distance between the radar and the
measurement area, and Az,,,, is the greatest elevation in the measurement area.
The DTM of the measurement area was produced a s described
above, by performing two iterations of the procedure and evaluating the vertical correction at each point, averaged in a 3m radius
circle.
RCS. m
24 -
!
23 22
-
-18
-17.25
-d
-17
-1 7.5
-18
1.6
14
21 -
1.2
2o
-17.5
E 19-
x
18
2
1.a
H
1.o
-17.25
-
17 -
-17
18 May 2000
0 IEE 2000
Electronics Lettecr Online N o : 20000997
D o l : 10.1049/el:20000997
M. Pieraccini, G. Luzi and C. Atzeni (Department of Electronics and
Tel~con2n~unications,University of Florence, viri Sunta Martri 3, 50139
Firenze, ltcilji)
E-mail: pieracciniOdiefi.die.uniii.it
References
MASSONET, o., and RARAUTE. T.: ‘Radar interferometry: limits and
potential’, IEEE Trcins. Geosci. Remote Sen.s., 1993, 31, (2), pp.
455-464
MARECHAL, N.: ‘Tomographic formulation of interferometric SAR
for terrain elevation mapping’, IEEE Trcins. Geosci. Remote Sens.,
1995, 33, (3), pp. 726-739
IIZUKA, K., FREUNDORFER, A.P., WU, K.H., MORI. 13, OGURA, H., atld
NGUYBN, v.K.: ‘Step-frequency radar’, J . Appl. P/iy.s., 1984, 56, (9),
pp. 2572-2583
PIERACCINI, M., TAIICHI, D.. RUDOLF, H., LEVA, D., LUZI, G , and
ATZENI, c : ‘Interferometric radar for remote monitoring of
building deformations’, Electron. Lett., 2000, 36, (6), pp. 569-570
0.8
-1 6.5
0.6
AFC scheme for discontinuous pilot mobile
communication system
x, m
Yan Zhang and XiaoHu Yu
Fig. 4 Contour map and rndur cross-section ( R C S ) inicige
An automatic frequency correction scheme Tor ii spread-spectrum
communication system with a discontinuous pilot channel 1s
proposed. The simulation results show that this scheme is en“ective
and the maximum frequency offset this scheme can overcome is
7 kHz.
Figures are relative to the plan where the radar is positioned
(i) trihedral corner reflector
(ii) lamp pole
24,-
,
I
,
,
,
,
I-,
I
!
i
22 /
,
’
’--
x, m
3rd August 2000
the carried frequency inay be higher than 2 GHz. For W-CDMA
systems, the downlink carried frequency is 2.1 1-2.17GHz [l]. The
frequency stability of a mobile station’s local crystal oscillator
approximates to 3PPM [l], which means that the maximum offset
between the frequency of the downlink carrier and local oscillator
can be up to 6.5 I kHz. This is too large for a RAKE receiver to
work normally.
In [2] Li, Yu and Cheng presented an AFC scheme in which a
transmission model with a continuous pilot channel is adopted.
However, this scheme cannot be employed in discontinuous pilot
channel systems. On the other hand, the maximum frequency offset that the scheme can correct is only 1kHz, far lower than the
required 6.510kHz.
Estimation of Jiequency bius: The transmission model (shown in
Fig. 1) proposed in this Letter includes a pilot channel and a data
channel which are time multiplexed and QPSK modulated. The
model is similar to W-CDMA [I]. The pilot channel employs an
unmodulated complex pseudo-noise (PN) code. The received signal that is downconverted to the baseband and sampled at the
1106151
Fig. 5 Contour mcip of optically measured DTM
Figures are relative to the plan where the radar is positioned
(i) trihedral corner reflector
(ii) lamp pole
ELECTRONICS LETTERS
Introduction: Ln third-generation mobile communication systems,
Vol. 36
No. 16
1417
and
where P,, is the transmitting power of the pilot channel, PClis the
transmitting power of the data channel, P,,(t) is the PN code used
by the pilot, N is the spreading factor (SF), T, is the chip interval,
T;,is the symbol interval T, = NT,, Sn(t)is the transmitting signal
of the nth data channel, L is the number of multipaths, ACOis the
frequency offset, n(k) is complex additive white Gaussian noise
(AWGN), and a;,8, and q are random variables, representing the
Ith path gain, phase and delay. respectively.
(5)
1=1
where n(m) and q(n) are the remaining noise (interference). In
deriving eqns. 3 and 4, the influence of noise is ignored for clarity.
From eqn. 5, the frequency offset estimation can be obtained a s
follows:
data channel K
"'data bits
AFC loop; The structure of the automatic frequency correction
loop is shown in Fig. 3. Having been estimated in eqn. 6, the frequency offset is quantised to two levels: ' I ' or '-l', corresponding
to the sign of its input. The loop filter is an accumulator the input
of which is x(m) and output of which is y(nz). The Z transforms of
x(m) and y(m) are X(z) and Y(z),respectively. We obtain
data channel 2
Ndatabits
pilot channel
2*Npilot bits
data channel 1
0.625ms, 2*40bits
slot 1
slot m
slot 2
slot ...
y(7n) =
az(m)+ y(rn
-
1)
H ( 2 ) = Y ( 2 )X(2)
a
~
1-2-1
(7)
where CL is the frequency fine-tuning step of the AFC in one slot.
The AFC scheme uses a digital filter (accumulator) instead of an
analogue filter. When the pilot interval is over, we set: x(m) = 0,
y(m) = y(m - 1). It can thus work in a discontinuous pilot communication system.
Fig. 1 Frume structure f o r trunsviission model (QPSK)
offset estimator
I
L
I
T
I
I
&$-qqqgjp~
hard decision
1624131
Fig. 3 Structure
o j AFC loop
"'pilot-1
Fig. 2 Structure o j j k g u e n c y bius estimation
The coherence time of the channel is much larger than the symbol duration c,,hence the channel parameters are considered to
be coilstant over the observation interval.
The structure of the frequency offset estimation is shown in
Fig. 2. The estimation values q(n) of the Ith channel can be written as
sin(N
Q(n)=
e - (~A w q -01
4 5 sill(&Tc/2)
e 4~AUT,( 2 N n+ N
-
1)
TZ
1418
)
+ nl(n)
= 0,1, ...,Npzlot- 1
(2)
Sinzulatiun results: In our experiment, the parameters were TI, =
64Tc = 15.625p, processing gain N = 64, U = 10, N,,,,, = 8 and
Nrrrrro
= 32. The multipath fading channel was a typical ITU-R
M. 1225 vehicular channel model [3]. The maximuin Doppler frequency shift was 300Hz. Channel coding adopted 1/3 convolutional coding with constraint length 9. Fig. 4 depicts the
performance of the RAKE receiver with the frequency offset set
from 1 to 7kHz. Without an AFC loop, the receiver cannot work
at 1kHz frequency offset. With an AFC loop, the performance of
the receiver is slightly improved when the SNR is less than or
equal to 7dB. If the SNR is no less than 9dB, the performance of
the RAKE receiver is almost no longer affected by the frequency
offset, and is comparable to the case of zero frequency offset.
Analysis; In theory, our scheme has two benefits. (i) The maximum frequency offset this AFC scheme can overcome is large.
The discernible frequency offset IAffl = IAw/2z/ is determined by
eqn. 3, from which it must be satisfied that -IC < AwT, < z, i.e.
IAffl < 1/(2Th).In our simulation, IAfIlnn,was -32kHz. (ii) The
output X(m) is obtained as a hard decision, which means the
scheme possesses good anti-noise (interference) ability.
ELECTRONICS LETTERS
3rd August 2000
Vol. 36
No. 16
The deficiency of this scheme is the long time the AFC needs to
track the frequency offset, e.g. if the frequency offset is 7kHz, at
least 0.625 * 7/10 = 0.4375s tracking time is required.
5
6
7
8
9
11
10
SNR, dB
1624/41
Fig. 4 Relationship between BER and S N R
fre uency offset AFC
-&
0.0 kHz OFF
-XI.OkHz ON
-+-
-%-
-e-
3.0kHz ON
5.OkHz ON
7.0kHz ON
l.OkHz OFF
....o...
Introduction; In DS/CDMA cellular systems, when higher bit rate
services are introduced into conventional low bit rate systems, the
high bit rate terminals occupy almost all resources, which means
that the capacity for the low bit rate terminals is extremely
reduced. Of course, the number of supported high bit rate terminals could be restricted by using a method such as a higher air
charge. However, even if the number of high bit rate terminals is
1, if the bit rate is 100 times higher than that of the low bit rate
terminals, the system would allocate no resources to 100-channel
low bit rate terminals. From the point of view of service provision,
reduced capacity for low bit rate terminals is not preferable
because low bit rate traffic, especially voice traffic, is dominant,
especially in multimedia communication systems. Interference suppression techniques are therefore necessary to overcome the problem.
An adaptive array antenna is a very popular technique for solving this problem, because it can create a null in the direction of
high-power interference for the reception of low-power terminals.
However, there are two major problems associated with the adaptive array antenna application. First, the computational load
required to individually create an antenna beam pattern for all the
connected terminals is very high, and secondly, it takes a longer
time to converge to the optimum beam pattern when the desired
signal level is extremely low, or the interference level is extremely
high. To solve these problems, we propose a beam pattern selection diversity technique combined with transmit power control
using a delay profile measurement based beamforming technique.
uplink
from terminal 1
Conclusions: In this Letter, an AFC scheme for use in discontinuous pilot communication systems is proposed. The simulation
results show that a large frequency offset can be eliminated, the
performance of the RAKE receiver can be improved greatly and
the AFC scheme is a more effective and credible scheme.
0 IEE 2000
Electronics Letters Online No: 20001000
DOI: IO.1049/el:20001000
i M
from terminal M
I
adaDtive
h
14 June 2000
i
Yan Zhang and XiaoHu Yu (National Mobile Communications
Research Laboratory, Southeust University, Nanjing. People 2 Republic
convergenced
delay profile
estimation unit
random signal
generator
K
x
L
w
I/ IN
I/
I
of China)
References
LI
1 ETSI SMG2 UMTS ITU expert group: ‘Description of the
terrestrial radio access system’. SMG2 UMTS-ITU 26/98
2 LI, YAN, and YU, xmoHu: ‘Automatic frequency corrcction scheme
for spread spectrum coherent RAKE receiver’, Electron. Lett.,
1998, 34, (9), pp. 844-845
3 China Wireless Telecommunication Standard (CWTS) Working
Group l(WG1) Nodc B Radio Transmission and Reception,
Annex B.2 Channel models TS C402 v.3.0.0 (1999-10)
Beam pattern selection diversity technique
for DSKDMA reverse link
K. Watanabe, S. Sampei and N. Morinaga
A beam pattern selection diversity technique combined with
transmit power control VPC) is proposed for suppressing high
level interference from high bit rate terminals to low bit rate
terminals in the uplink of DS/CDMA systems. The proposed
system employs a delay profile measurement based beamforming
technique to calculate the optimum beamform for the reception of
signals from terminals. To reduce the amount of computation
required to control the beam directivity, the proposed system
calculates the optimum beam directivity for only a limited number
of terminals located far from the base station, and signals from
the rest of the terminals are demodulated by selecting one of the
most reliable adaptive array combined signals. TPC is employed
to compensate for the performance degradation caused by such
suboptimal control. Computer simulation confirms that the
proposed technique is effective in suppressing high level
interference with a relatively low signal processing load.
ELECTRONICS LETTERS
3rd August 2000
Vol. 36
I.
\r
I reference
1652/11
Fig. 1 Adaptive beam-forming unit
Beamforming using measured delay proj‘ile: In the proposed
scheme, we employ the delay profile measurement based beainforming technique proposed in [I]. Fig. 1 shows its configuration,
where K is the number of antenna elements and M is the number
of terminals in each cell. Because CDMA systems are designed to
detect a pilot channel with a higher signal-to-noise power ratio
(SNR) than that for traffic channels, the employed adaptive beamforming unit, first of all, measures the delay profile at each
antenna element for all terminals using the pilot channel, and the
quasi-received signal for each terminal received at each antenna
element is generated by calculating the convolution between the
measured delay profile and a known QPSK symbol sequence. The
optimum antenna weight is then calculated using the generated
quasi-received signal. By using this quasi-received signal generation process as well as applying the RLS algorithm, we can accurately and quickly calculate the optimum antenna weights for a
target terminal even if the interference signal level is very high or
the desired signal level is very low, which is a very important
advantage of the delay profile measurement based beamforming
technique. However, when the weight calculation is performed
over all terminals in each cell, the signal processing load becomes
very high, even if the system employs this scheme. We therefore
propose a beam pattern selection diversity technique combined
with TPC.
Beam pattern selection diversity technique combined with TPC:
Even when a main beam is directed to a target low power terminal
No. 16
1419