Patent application title: SYMMETRIC LOAD DELAY CELL OSCILLATOR

Abstract:

An oscillator includes a control circuit and a ring of symmetric load
delay cells. Each delay cell includes two novel symmetric loads. Each
load involves a level shift circuit and a diode-connected transistor
coupled in parallel with a current source-connected transistor. The
control circuit converts an oscillator input signal into bias control
signals that in turn control the effective resistance of the symmetric
loads such that delays through the delay cells are a function of the
input signal. The control circuit uses a symmetric load replica in a
control loop to control the level shift circuits of the delay cells such
that the oscillating delay cell output signals have a constant amplitude.
In a first advantageous aspect, due to the constant amplitude, the
oscillator is operable over a wide frequency range. In a second
advantageous aspect, the oscillator input signal to output signal
oscillation frequency has a substantially linear relationship.

Claims:

1. An oscillator comprising:a delay cell comprising:a first symmetric load
comprising a current source-connected transistor, a diode-connected
transistor, and a level shift circuit, wherein a drain of the current
source-connected transistor is coupled to a drain of the diode-connected
transistor; anda control circuit that controls the level shift circuit
such that the level shift circuit adjusts a gate voltage of the
diode-connected transistor, wherein the adjusting of the gate voltage
causes a lower swing limit of an output signal of the delay cell to be
substantially constant as an oscillating frequency of the oscillator
changes.

2. The oscillator of claim 1, wherein the lower swing limit changes by
less than five percent over an oscillating frequency range of more than
three hundred percent.

3. The oscillator of claim 1, wherein the control circuit receives a
frequency control input signal, wherein changes to the frequency control
input signal have a substantially linear relationship with respect to
corresponding changes in the oscillating frequency of the oscillator, and
wherein the oscillating frequency varies less than ten percent from a
perfectly linear oscillating frequency to frequency control input signal
relationship over an oscillating frequency range of more than three
hundred percent.

4. The oscillator of claim 1, wherein the delay cell further comprises
second symmetric load, the second symmetric load including a current
source-connected transistor, a diode-connected transistor, and a level
shift circuit, wherein a drain of the current source-connected transistor
of the second symmetric load is coupled to a drain of the diode-connected
transistor of the second symmetric load, and wherein the control circuit
controls the level shift circuit of the second symmetric load.

5. The oscillator of claim 4, wherein the delay cell further comprises a
tail current source transistor, wherein the control circuit supplies a
first bias signal onto a gate of the current source-connected transistor
of the first symmetric load and onto a gate of the current
source-connected transistor of the second symmetric load, and wherein the
control circuit supplies a second bias signal onto a gate of the tail
current source transistor.

6. The oscillator of claim 5, wherein the delay cell further comprises a
first switching transistor and a second switching transistor, the first
switching transistor having a drain coupled to the first symmetric load
and having a source coupled to the tail current source transistor, the
second switching transistor having a drain coupled to the second
symmetric load and having a source coupled to the tail current source
transistor, wherein a gate of the first switching transistor is a first
input node of the delay cell and the drain of the first switching
transistor is a first output node of the delay cell, and wherein a gate
of the second switching transistor is a second input node of the delay
cell and the drain of the second switching transistor is a second output
node of the delay cell.

7. The oscillator of claim 1, wherein the level shift circuit comprises:a
first transistor having a gate, a source and a drain, wherein the gate of
the first transistor is coupled to the drain of the current
source-connected transistor and to the drain of the diode-connected
transistor, wherein the drain of the first transistor is coupled to a
supply voltage node, and wherein the source of the first transistor is
coupled to a gate of the diode-connected transistor.

8. The oscillator of claim 7, wherein the first transistor has a threshold
voltage of approximately zero volts.

9. The oscillator of claim 7, wherein the level shift circuit further
comprises:a second transistor having a gate, a source and a drain,
wherein the drain of the second transistor is coupled to the source of
the first transistor of the level shift circuit, wherein the source of
the second transistor of the level shift circuit is coupled to a ground
node, and wherein the gate of the second transistor is coupled to receive
a lower swing limit control signal (LSLCS) from the control circuit.

10. The oscillator of claim 1, wherein the control circuit comprises:a
replica symmetric load, wherein the replica symmetric load includes a
current source-connected transistor, a diode-connected transistor, and a
level shift circuit, wherein a drain of the current source-connected
transistor of the replica symmetric load is coupled to a drain of the
diode-connected transistor of the replica symmetric load; anda feedback
control circuit that has a first input node, a second input node, and an
output node, wherein the first input node is coupled to the drain of the
current source-connected transistor of the replica symmetric load and to
the drain of the diode-connected transistor of the replica symmetric
load, and wherein the output node is coupled to the level shift circuit
of the replica symmetric load and to the level shift circuit of the first
symmetric load of the delay cell.

11. The oscillator of claim 10, wherein the feedback control circuit is a
differential amplifier, and wherein the second input node of the feedback
control circuit is coupled to receive a reference voltage.

12. The oscillator of claim 1, wherein the delay cell is one of the
plurality of substantially identical delay cells, and wherein the delay
cells of the plurality of substantially identical delay cells are coupled
together into a ring.

13. A symmetric load circuit comprising:a supply voltage node;a current
source-connected transistor having a gate, a source, and a drain, wherein
the source of the current source-connected transistor is coupled to the
supply voltage node;a diode-connected transistor having a gate, a source
coupled to the source of the current source-connected transistor, and a
drain coupled to the drain of the current source-connected transistor;
anda level shift circuit that adjusts a gate-to-source voltage (VGS) of
the diode-connected transistor.

14. The symmetric load circuit of claim 13, wherein the level shift
circuit includes a transistor having a gate, a source, and a drain,
wherein the gate of the transistor of the level shift circuit is coupled
to the drain of the current source-connected transistor and to the drain
of the diode-connected transistor, and wherein the source of the
transistor of the level shift circuit is coupled to the gate of the
diode-connected transistor.

15. The symmetric load circuit of claim 13, wherein the symmetric load
circuit is a part of an oscillator having a plurality of stages, and
wherein a node coupled to the drain of the diode-connected transistor and
to the drain of the current source-connected transistor is an output node
of one of the stages.

16. The symmetric load circuit of claim 13, wherein the symmetric load
circuit has an effective resistance between the supply voltage node and
the drain of the current source-connected transistor, and wherein a
control signal on the gate of the current source-connected transistor is
changed to change the effective resistance.

17. An oscillator comprising:a current source-connected transistor of a
symmetric load;a diode-connected transistor of the symmetric load,
wherein a drain of the current source-connected transistor is coupled to
a drain of the diode-connected transistor; andmeans for adjusting a gate
voltage of the diode-connected transistor such that a lower swing limit
of an oscillating signal of the oscillator remains substantially constant
as an oscillating frequency of the oscillator changes.

18. The oscillator of claim 17, wherein the means in addition to adjusting
the gate voltage of the diode-connected transistor is also for outputting
a first control signal and a second control signal, wherein the first
control signal is supplied to a gate of the current source-connected
transistor of the symmetric load, and wherein the second control signal
is supplied to a tail current source transistor, wherein the symmetric
load and the tail current source transistor are parts of a delay cell of
the oscillator.

19. The oscillator of claim 17, wherein the means includes a transistor
having a gate coupled to the drain of the current source-connected
transistor and to the drain of the diode-connected transistor, and also
has a source coupled to a gate of the diode-connected transistor of the
symmetric load.

20. The oscillator of claim 17, wherein the means is a part of the
symmetric load.

21. The oscillator of claim 17, wherein the means includes a feedback
control loop that is not a part of the symmetric load.

22. A method comprising:(a) controlling a gate-to-source voltage of a
diode-connected transistor of a first symmetric load such that a swing
limit voltage of an oscillating signal on a node of the first symmetric
load is controlled to have a substantially constant value, wherein the
first symmetric load includes a current source-connected transistor,
wherein a source of the diode-connected transistor is coupled to a source
of the current source-connected transistor, and wherein a drain of the
diode-connected transistor is coupled to a drain of the current
source-connected transistor.

23. The method of claim 22, wherein the swing limit voltage changes by
less than five percent over an oscillating frequency range of the
oscillating signal of more than three hundred percent.

24. The method of claim 22, further comprising:(b) generating a control
signal using a feedback control loop; and(c) supplying the control signal
to a level shift circuit such that the level shift circuit controls the
gate-to-source voltage in (a).

25. The method of claim 22, wherein the first symmetric load, a second
symmetric load, a first switching transistor, a second switching
transistor, and a tail current source transistor are parts of a delay
cell, wherein a drain of the first switching transistor is coupled to the
first symmetric load, wherein a source of the first switching transistor
is coupled to the tail current source transistor, wherein a drain of the
second switching transistor is coupled to the second symmetric load, and
wherein a source of the second switching transistor is coupled to the
tail current source transistor.

26. The method of claim 25, further comprising:(b) controlling a
gate-to-source voltage of a diode-connected transistor of the second
symmetric load such that a swing limit voltage of an oscillating signal
on a node of the second symmetric load is controlled to have a
substantially constant value.

27. A method comprising:receiving a frequency control input signal onto an
oscillator, wherein a change in the frequency control input signal causes
a corresponding change in an oscillating frequency of an oscillating
signal in the oscillator, wherein the oscillating signal is present on a
node of a symmetric load of a delay cell in the oscillator;
andcontrolling a gate-to-source voltage of a diode-connected transistor
of the symmetric load such that the oscillating frequency of the
oscillating signal varies less than ten percent from a perfectly linear
oscillating frequency to frequency control input signal relationship over
an oscillating frequency range of more than three hundred percent.

Description:

BACKGROUND INFORMATION

[0001]1. Technical Field

[0002]The disclosed embodiments relate to symmetric load delay cells such
as are found in voltage-controlled oscillators (VCO) and
current-controlled oscillators (ICO).

[0003]2. Background Information

[0004]Current-Controlled Oscillators (ICOs) see use in many circuits such
as, for example, Phase-Locked Loops (PLL) within clock signal generators
that supply clock signals to digital processors. ICOs sometimes also see
use in local oscillators of radio receivers and radio transmitters and in
other circuits. FIG. 1 (Prior Art) is a simplified diagram of one example
of a simple PLL 1. PLL 1 includes a Voltage-Controlled Oscillator (VCO) 2
that in turn includes a voltage-to-current converter 3 and an ICO 4. The
local oscillator (LO) signal output by ICO 4 is frequency divided by a
loop divider 5. The resulting divided-down feedback signal 6 is
phase-compared with respect to a reference clock signal XO by a phase
detector 7. Phase detector 7 outputs an error signal that is processed by
a charge pump 8 and loop filter 9 to generate a signal 10 that controls
VCO 2. The feedback control loop operates to phase-lock the feedback
signal 6 to the reference clock signal XO. By setting the frequency
control value by which loop divider 5 divides, the frequency of the local
oscillator signal LO can be set to have a desired frequency over a
frequency tuning range.

[0005]FIG. 2 (Prior Art) is a simplified diagram of ICO 4 of FIG. 1. ICO 4
actually includes two current-controlled oscillators ICO#1 11 and ICO#2
12 because a single current-controlled oscillator of this type would not
have an adequately wide tuning range.

[0006]ICO 4 also includes two VCO buffer circuits 13 and 40 and a
two-to-one digital logic multiplexer 41. Each VCO buffer circuit converts
a differential ICO output signal into a single-ended digital signal. In
the example of FIGS. 1 and 2, the overall ICO 4 is tunable to generate an
output signal LO that ranges from approximately 400 MHz to 1.3 GHz.

[0007]ICO#1 11 and VCO buffer circuit 40 are used when the LO signal is to
be of a frequency in the 400 MHz to 800 MHz range, whereas ICO#2 12 and
VCO buffer circuit 13 are used when the LO signal is to be of a frequency
in the 800 MHz to 1.3 GHz range. Depending on the frequency of the
desired output signal LO, one of the ICO 11 or 12 is enabled and other is
disabled. Multiplexer 41 is controlled to output the output signal of the
appropriate VCO buffer circuit as the single-ended local output signal
LO.

[0008]FIG. 3 (Prior Art) is a more detailed diagram of ICO#1 11 of FIG. 2.
This particular ICO circuit is ring oscillator and involves a bias
control circuit 14, and a plurality of delay cells 15-19. The delay cells
have differential signal inputs and differential signal outputs. ICTL
signal 37 is input control signal ICTL 20 received via transistor 38 of
FIG. 2 from the voltage-to-current converter 3 of FIG. 1.

[0009]FIG. 4 (Prior Art) is a diagram that illustrates bias control
circuit 14 and delay cell 15 in further detail. All the delay cells of
FIG. 3 are of similar topology. Delay cell 15 includes a pair of circuits
21 and 22 referred to as "symmetric loads". The first symmetric load 21
includes a current source-connected transistor (CSCT) 23 and
diode-connected transistor (DCT) 24. The second symmetric load 22
includes a current source-connected transistor (CSCT) 25 and
diode-connected transistor (DCT) 26. Delay cell 15 also includes a tail
current source transistor 28 and two current steering switching
transistors 29 and 30.

[0010]Tail current transistor 28 pulls a substantially fixed control
current ICTL from node N3.

[0011]As the differential input signal (VIP minus VIN) between inputs
nodes 31 and 32 switches, current ICTL is steered to flow through the
first symmetric load, and then through the second symmetric load, and
back again. The differential output signal VOP minus VON present between
nodes N2 34 and N1 33 is output via leads 36 and 35 to the next delay
cell in the ring of delay cells.

[0012]FIG. 5 (Prior Art) illustrates operation of delay cell 15 during a
first portion of a switching cycle. Switching transistor 29 is relatively
conductive and switching transistor 30 is relatively nonconductive. The
voltage on node N1 is pulled down to be lower than the voltage on node
N2. The voltage of the output signal therefore increases as capacitor 39
charges. Resistor symbol R1 represents the parallel equivalent effect
resistance of the first symmetric load 21. Resistor symbol R2 represents
the parallel equivalent effect resistance of the second symmetric load
22.

[0013]FIG. 6 (Prior Art) illustrates operation of delay cell 15 during a
second portion of the switching cycle. Switching transistor 29 is
relatively nonconductive and switching transistor 30 is relatively
conductive. The voltage on node N2 is therefore pulled down to be lower
than the voltage on node N1. The voltage of the output signal therefore
decreases as capacitor 39 discharges.

[0014]FIG. 7 (Prior Art) is a simplified waveform diagram that illustrates
how the differential output signal between nodes N2 and N1 oscillates up
and down as the switching transistors 29 and 30 are controlled to be on
and off from cycle to cycle. The lower swing limit of the output signal
is referred to as the Lower Swing Limit Voltage (LSLV). The upper swing
limit of the output signal is approximately the high supply voltage VDD
of the circuit. For additional information on a delay cell of this type
involving symmetric loads see the article "Low-Jitter Process-Independent
DLL and PLL Based on Self-Biased Techniques", IEEE Journal of Solid-State
Circuits, Vol. 31, No. 11, November 1996, by John G. Maneatis.

[0015]The frequency of oscillation of the ring oscillator is changed by
changing the input control current ICTL. The circuit is therefore
referred to as a current-controlled oscillator.

[0016]If control current ICTL supplied to bias control circuit 14 is
increased, then the current ICTL pulled through the tail transistor 28
increases. Also, the PBIAS control voltage decreases. The decrease in the
PBIAS control signal reduces the effective resistance REFF of the first
and second symmetric loads 21 and 22. The reducing of the effective
resistance REFF reduces the RC delay through the delay cell, thereby
increasing the frequency of oscillation FOSC. Similarly, decreasing
control current ICTL supplied to bias control circuit 14 causes a
decrease in the frequency of oscillation FOSC.

[0017]Although a ring oscillator of this type works well in certain
applications, it may have an undesirably narrow frequency tuning range.
As the input control current ICTL increases, the voltage swing of the
output signal between nodes N2 and N1 increases. The lower swing limit
voltage LSLV drops lower and lower as ICTL increases and increases.

[0018]Because the lower voltage limit of the output signal has a lower
voltage limit, the frequency tuning range of the ring oscillator circuit
is effectively limited.

[0019]FIG. 8 (Prior Art) is a chart that illustrates how ICO#1 is only
usable over a limited frequency tuning range up to 800 MHz because at the
upper input control current limit of ICTLMAX the voltage swing of its
output signal has reached its maximum allowable voltage swing. Note in
FIG. 8 that the dashed line representing the voltage swing has reached
the MAX ALLOWABLE VOLTAGE SWING value. As a result, if the frequency
tuning range of the overall ICO 4 of FIG. 1 is to extend above 800 MHz,
then a second ICO#1 must be provided. The circuit of FIG. 2 therefore
includes ICO#2. In addition, as FIG. 8 illustrates, the change in
frequency of oscillation FOSC is significantly non-linear with respect to
changes in input control current ICTL. Note that the solid line
representing the relationship of FOSC to ICTL is not a straight line, but
rather is curved. At the upper end of frequency operation of an ICO of
this type, the input control current ICTL must be increased a relatively
large amount in order to increase the oscillation frequency FOSC a
relatively small amount. An improved circuit is desired.

SUMMARY

[0020]A novel Current Controlled Oscillator (ICO) includes a novel bias
control circuit and a number of novel symmetric load delay cells. The
novel symmetric load delay cells are coupled together in a ring to form a
ring oscillator. An oscillator input control current (ICTL) supplied to
the bias control circuit controls the oscillation frequency (FOSC) of an
oscillator output signal output by the ring of delay cells.

[0021]Each delay cell includes a pair of novel symmetric load circuits.
Each novel symmetric load circuit includes a current source-connected
transistor, a diode-connected transistor, and a novel level shift
circuit. The novel level shift circuit can adjust the gate-to-source
voltage (VGS) of the diode-connected transistor in response to a control
signal referred to as the "Lower Swing Limit Control Signal" (LSLCS).
First and second switching transistors within the delay cell steer a
control current to flow from a voltage supply node (VDD node) through one
or the other of the symmetric loads, then through a conductive one of the
switching transistors, and then through a tail current source transistor
to a ground node (GND node). The effective resistance of the symmetric
loads is determined at least in part by a control signal PBIAS supplied
to the current source-connected transistor, by a control signal NBIAS
supplied to the tail current source, and by the LSLCS control signal
supplied to the level shift circuits of the delay cell. The delay of the
delay cell, and therefore the frequency of oscillation of the ring of
delay cells, is controlled by using the control signals PBIAS, NBIAS and
LSLCS to control the effective resistance of the symmetric loads. The
novel bias control circuit generates the PBIAS, NBIAS and LSLCS control
signals such that the effective resistance of the symmetric loads in the
delay cells changes to change the ring oscillator frequency as a function
of the input control current ICTL.

[0022]The novel bias control circuit also includes a replica circuit. The
replica circuit includes a replica of the novel symmetric load, a replica
of a switching transistor, and a replica of a tail current source
transistor coupled together as these circuits are coupled together in one
of the delay cells. The voltage dropped across the replica symmetric load
in this replica circuit is the same as the voltage dropped across one of
the symmetric loads in a delay cell when the switching transistors in the
delay cell are switched such that all the tail current flows through one
of the symmetric loads. The voltage on a node of the symmetric load in
the delay cell in this condition is the lower limit of the voltage of the
output signal of the delay cell (the "lower swing limit voltage"). This
lower limit is the lower swing limit voltage of the oscillating signal as
output from the delay cell. An operational amplifier in the novel bias
control circuit is used in a feedback control loop to set the lower swing
limit of the delay cells of the loop. The operational amplifier compares
the voltage on the node of the replica symmetric load to a desired
reference voltage (VREF). The operational amplifier outputs an error
control voltage that is supplied to the replica level shift circuit such
that the effective resistance of the replica symmetric load is such the
voltage on the node has the desired reference voltage value VREF. Because
the replica level shift circuit in the bias control circuit is of
identical construction to the level shift circuits in the delay cells,
supplying the control voltage from the operational amplifier to the level
shifting circuits in the delay cells causes the lower swing limit voltage
of the output signals of the delay cells to be fixed to VREF. Setting the
lower swing limit voltage of the delay cells prevents the amplitude of
the oscillating output signals of the delay cells from increasing as a
function of increasing oscillation frequency. In a first advantageous
aspect, fixing output signal amplitude in this way increases the
frequency tuning range of the ICO. In a second advantageous aspect,
fixing output signal amplitude causes the input control current (ICTL) to
output signal frequency (FOSC) relationship of the ICO to be
substantially linear as compared to conventional JCOs involving
conventional symmetric loads.

[0023]The foregoing is a summary and thus contains, by necessity,
simplifications, generalizations and omissions of detail; consequently,
those skilled in the art will appreciate that the summary is illustrative
only and does not purport to be limiting in any way. Other aspects,
inventive features, and advantages of the devices and/or processes
described herein, as defined solely by the claims, will become apparent
in the non-limiting detailed description set forth herein.

BRIEF DESCRIPTION OF THE DRAWINGS

[0024]FIG. 1 (Prior Art) is a simplified diagram of one type of a
conventional Phase-Locked Loop (PLL).

[0025]FIG. 2 (Prior Art) is a simplified diagram of a Current Controlled
Oscillator (ICO) within the PLL of FIG. 1.

[0026]FIG. 3 (Prior Art) is a more detailed diagram of the ICO of FIG. 2.

[0027]FIG. 4 (Prior Art) is a simplified diagram that illustrates the bias
control circuit and one of the delay cells of FIG. 3 in further detail

[0028]FIGS. 5 and 6 (Prior Art) are diagrams that illustrate operation of
a delay cell of FIG. 3 during first and second portions of a switching
cycle of the switching transistors of the delay cell.

[0029]FIG. 7 (Prior Art) is a simplified waveform diagram that illustrates
oscillation of the output signal from a delay cell as it cycles.

[0030]FIG. 8 (Prior Art) is a chart that illustrates how the voltage swing
of the output signal of a delay cell in the ICOs of FIG. 2 increase as a
function of input control current (ICTL), and how the frequency of
oscillation of the ICO changes in a non-linear fashion as a function of
ICTL.

[0031]FIG. 9 is a simplified diagram of one example of a novel
Current-Controlled Oscillator (ICO) in accordance with one novel aspect.

[0032]FIG. 10 is a more detailed diagram of one of the delay cells of the
novel ICO of FIG. 9.

[0033]FIG. 11 is a circuit diagram of the novel bias control circuit of
the novel ICO of FIG. 9.

[0034]FIG. 12 is a simplified diagram of a conventional symmetric load of
a delay cell.

[0035]FIG. 13 is a diagram that illustrates the current-to-voltage
relationship (referred to as the I-V curve) for both the diode-connected
device and the current source-connected device of the conventional
symmetric load of FIG. 12.

[0036]FIG. 14 is a diagram that illustrates the composite I-V curves for
the conventional symmetric load of FIG. 12.

[0037]FIG. 15 is a simplified diagram of a novel symmetric load in the
novel delay cell of FIG. 10.

[0038]FIG. 16 is a chart that illustrates the current-to-voltage
relationship (referred to as the I-V curve) for both the diode-connected
device and the current source-connected device of the novel symmetric
load of FIG. 15.

[0039]FIG. 17 is a chart that illustrates the composite I-V curves for the
novel symmetric load of FIG. 15.

[0040]FIG. 18 is a more detailed diagram of one way the novel symmetric
load of FIG. 15 can be realized.

[0041]FIG. 19 is an equation that describes the effective resistance REFF
of a conventional symmetric load.

[0042]FIG. 20 is a diagram that illustrates the relationship of input
control current (ICTL) signal to the frequency of oscillation (FOSC) of a
conventional ICO that employs conventional symmetric loads.

[0043]FIG. 21 is an equation that defines the effective resistance REFF of
the novel symmetric load of FIGS. 15 and 18.

[0044]FIG. 22 is a diagram that illustrates the relationship of the input
control current ICTL to oscillation frequency FOSC in the novel Current
Controlled Oscillator ICO of FIG. 9 that includes the novel symmetric
load and novel bias control circuit.

[0045]FIG. 23 is a simplified flowchart of a method 200 in accordance with
one novel aspect.

[0046]FIG. 24 is a simplified flowchart of a method 300 in accordance with
one novel aspect.

DETAILED DESCRIPTION

[0047]FIG. 9 is a simplified diagram of one example of a
Current-Controlled Oscillator (ICO) 50 in accordance with one novel
aspect. ICO 50 may, for example, see use in a Phase-Locked Loop (PLL)
within a local oscillator of a radio receiver or a radio transmitter. ICO
50 includes a bias control circuit 51, five delay cell stages 52-56
coupled together in a ring, and a VCO buffer 57. Increasing an input
control current ICTL received on input lead 58 results in a corresponding
increase in the frequency of oscillation (FOSC) of an output signal LO on
output lead 59. Unlike the conventional circuit of FIG. 3, the input
control current to output frequency relationship is substantially linear
over a relatively wide frequency tuning range.

[0048]FIG. 10 is a more detailed diagram of delay cell 52 of FIG. 9. All
delay cells 52-56 are of identical construction. Delay cell 52 includes a
first symmetric load 60, a second symmetric load 61, a tail current
source transistor 62, a first switching transistor 63, a second switching
transistor 64, an output capacitor 65, a novel first level shift circuit
66, and a novel second level shift circuit 67. Each symmetric load
includes a diode-connected transistor and a current source-connected
transistor. As it is used in this application, the term diode-connected
transistor includes both a transistor with its gate and drain connected
together and a transistor with a level shifter connected between its gate
and drain. The two transistors are coupled together in parallel between a
supply voltage (VDD) node and another node so that the symmetric load
provides an effective resistance REFF between the two nodes. First
symmetric load 60 includes diode-connected transistor 68, current
source-connected transistor 69, and a novel level shift circuit 66.
Second symmetric load includes diode-connected transistor 70, current
source-connected transistor 71, and novel level shift circuit 67. Tail
current source transistor 62 is biased by bias voltage NBIAS so that tail
current source transistor 62 pulls a control current ICTL from node 72.
First and second switching transistors 63 and 64 are controlled to either
steer the control current ICTL to flow from the supply voltage VDD node,
through first symmetric load 60, through first switching transistor 63
and to node 72, or to flow from the supply voltage VDD node, through
second symmetric load 61, through second switching transistor 64 and to
node 72. As the delay cell switches, first switching transistor 63 turns
on as second switching transistor 64 turns off, and then second switching
transistor 64 turn on as first switching transistor 63 turns off, and so
forth. The switching of the switching transistors 63 and 64 is controlled
by the differential signal (VIP-VIN) on input nodes 73 and 74. The output
signal from the delay cell is present between nodes 76 and 75 across
capacitor 65. The output signal VOP-VON is present between output leads
78 and 77.

[0049]As in the conventional delay cell of FIG. 4, signal delay through
delay cell 52 is primarily controlled by the effective resistance REFF of
the symmetric loads 60 and 61 and the magnitude of the control current
ICTL flowing through tail current source transistor 62. To reduce the
signal delay through delay cell 52, the bias voltage PBIAS decreases as
control current ICTL is increased. Decreasing PBIAS increases the
magnitude of the gate-to-source voltage (VGS) on transistors 69 and 71,
thereby decreasing the effective resistances of the symmetric loads 60
and 61, thereby decreasing an RC time constant of the delay cell, and
thereby decreasing signal propagation delay through the delay cell.

[0050]Novel level shift circuit 66 includes a level shift transistor 79
and a level shift current source transistor 80. Transistor 80 is a source
follower. Similarly, novel level shift circuit 67 includes a level shift
transistor 81 and a level shift current source transistor 82. Transistor
82 is a source follower. Operation of these two level shift circuits 66
and 67 is explained in further detail below.

[0051]FIG. 11 is a circuit diagram of bias control circuit 51 of FIG. 9.
Bias control circuit 51 receives the input control current ICTL on input
lead 58 and outputs the PBIAS control signal on conductor 83, outputs the
NBIAS control signal on conductor 84, and outputs a novel Lower Swing
Limit Control Signal (LSLCS) on conductor 85. Current controlled current
source 86, the current mirror of N-channel transistors 87 and 88, and
N-channel transistor 89 and P-channel transistor 90 correspond to the
devices illustrated in bias control circuit 14 of the conventional
circuit of FIG. 4. The novel bias control circuit 51 of FIG. 9, however,
includes a replica circuit 91. Replica circuit 91 includes a replica of a
symmetric load 92 and 93 of a delay cell, a replica of a switching
transistor 94 of a delay cell, a replica of a tail current source
transistor 95 of the delay cell, and a replica of the transistors 96 and
97 of a level shift circuit of a delay cell. In addition, the novel bias
control circuit 51 includes an operational amplifier 98 operatively
coupled as illustrated.

[0052]Operational amplifier 98 in this case is a single-stage
current-mirror operational amplifier having N-channel inputs. The novel
bias control circuit 51 of FIG. 11 generates the PBIAS and NBIAS control
voltage signals on conductors 83 and 84 in similar fashion to the way
that the conventional bias control circuit 14 of FIG. 4 generates the
PBIAS and NBIAS control voltage signals.

[0053]Operational amplifier 98 is connected in a feedback loop with the
replica circuit 91.

[0054]The replica switching transistor 94 of the replica circuit 91 is
always on, so the current from the current mirror of N-channel
transistors 87 and 95 is always being pulled through the replica
symmetric load. This means that the voltage at the drains of transistors
92 and 93 is equal to the lower swing voltage on the output signal of the
delay cell 52 when the oscillator is oscillating. This voltage is present
on the inverting input lead of the operational amplifier 98. The
operational amplifier 98 receives a reference voltage (VREF) on its
non-inverting input lead and compares the voltage on the inverting input
lead to VREF. If VREF differs from the voltage on the inverting input
lead of the operational amplifier 98, operational amplifier 98 adjusts
the current through the replica symmetric load via the feedback loop
until the voltage at the replica symmetric load (on the drains of
transistors 92 and 93) is equal to VREF. Lead 85 provides this current
adjustment signal as the "Lower Swing Limit Control Signal" to each delay
cell of the oscillator, including to delay cell 52 as illustrated in FIG.
10.

[0055]FIG. 12 is a simplified diagram of a conventional symmetric load.
The conventional symmetric load includes a first transistor that is
connected in a configuration referred to here as a diode-connected
configuration, and a second transistor that is connected in a
configuration referred to here as a current source-connected
configuration.

[0056]FIG. 13 is a diagram that illustrates the current-to-voltage
relationship (referred to as the I-V curve) for the both the current
source-connected device and the diode-connected device of the
conventional symmetric load of FIG. 12. Each of the lines corresponds to
transistor operation at a different amount of drain current. For example,
line 99 describes transistor operation for a drain current of 50 μa;
line 100 describes transistor operation for a drain current of 100 μa;
line 101 describes transistor operation for a drain current of 150 μa;
and line 102 describes transistor operation for a drain current of 200
μa. FIG. 13 also includes a line 103 that describes operation of the
diode-connected transistor. As the gate-to-source voltage magnitude
increases and reaches a threshold voltage, the transistor turns on
rapidly resulting in an I-V curve similar to the I-V curve of a diode.
Because the two transistors are coupled together in parallel, the voltage
between the source and drain of one transistor is equal to the voltage
between the source and drain of the other transistor. If ICTL is divided
evenly between the current source-connected device and the
diode-connected device, the lower swing limit voltage corresponds to the
intersection of the I-V curves. Four such intersection points 104, 105,
106, and 107 are identified in the chart of FIG. 13.

[0057]FIG. 14 is a diagram that illustrates the composite I-V curves for
the conventional symmetric load of FIG. 13. The I-V curves for both
transistors for a given ICTL are combined such that there are four
composite curves 108, 109, 110 and 111 in FIG. 14. Arrow 112 represents
the voltage drop across the symmetric load, with AVCTL representing the
change in the control voltage VCTL when ICTL changes. The ratio of supply
voltage (VDD) minus lower swing limit voltage to ICTL is considered an
indication of the effective resistance of the symmetric load. Note that
in FIGS. 13 and 14 the voltage drop across the symmetric load (between
the source and drain of the two transistors) increases as the amount of
current flowing through the symmetric load increases. In the horizontal
dimension of FIG. 14, for example, the intersection points 113, 114, 115
and 116 occur at ever decreasing voltages. Because the sources of the
transistors of the symmetric load in the delay cell are coupled to a
supply voltage (VDD) node, the increasing voltage drop across the
symmetric loads with symmetric load current results in the lower swing
limit voltages on nodes 33 and 34 of FIG. 4 decreasing for increasing
amounts of ICTL. When all the ICTL current is steered through one of the
two symmetric loads and the voltage on the drains of the transistors of
the symmetric load has reached its minimum, this low voltage is the lower
swing limit voltage. There is a practical limit to how low the lower
swing limit voltage can go before delay cell operation is affected. How
low the lower swing limit voltage can go determines the tuning range
limit of the delay cell. It is desired that the delay cell has a wider
tuning range.

[0058]FIG. 15 is a diagram of a novel symmetric load 117 in accordance
with one novel aspect. Rather than just having one control input lead and
signal such as the conventional symmetric load of FIG. 12, the novel
symmetric load 117 of FIG. 15 has two control input leads 118 and 119 and
two control input signals 120 and 121. A level shift circuit 123 adjusts
and controls the gate voltage of the diode-connected transistor so that
the gate-to-source voltage magnitude of the diode-connected transistor is
increased by the change in VCTL (or equivalently ΔVDSAT) that the
control voltage VCTL decreased from its maximum setting as ICTL increases
as indicated in the I-V curves of FIG. 14.

[0059]FIG. 16 is a chart that illustrates operation of the novel symmetric
load 117 of FIG. 15. Now, for each I-V curve of the current
source-connected transistor, there is a different I-V curve for the
diode-connected transistor. Increasing ICTL effectively shifted the I-V
curve for the diode-connected device to the left. Note that in FIG. 16
there are now four I-V curves for the diode-connected transistor 124,
125, 126 and 127. Due to operation of the level shift circuit, the
voltage drop across the novel symmetric load remains at a constant
voltage despite increasing and decreasing amounts of control current
ICTL. The constant voltage drop is represented by the vertical dashed
line 128.

[0060]FIG. 17 is a chart that illustrates the composite I-V curves for the
novel symmetric load of FIG. 15.

[0061]FIG. 18 is a more detailed diagram of the novel symmetric load 117
of FIG. 15. Level shift circuit 123 includes a first zero threshold
voltage transistor (ZVT) 129 and a second level shift current source
transistor 130. How much the level shift circuit 123 adjusts the gate
voltage of diode-connected transistor 131 is controlled by the Lower
Swing Limit Control Signal (LSLCS). The symmetric loads 60 and 61 of the
delay cell of FIG. 10 and the replica symmetric load of the control
circuit of FIG. 11 are of the same construction as the novel symmetric
load illustrated in FIG. 18.

[0062]FIG. 19 is an equation that defines the effective resistance REFF of
a transistor of the prior art symmetric loads of FIG. 12. Effective
resistance REFF is a function of input control current ICTL. VT is the
inversion threshold voltage of either of the symmetric load transistors.
VDSAT is the saturation voltage of the current source symmetric load
transistor at a given value of ICTL. VDSAT plus VT defines the voltage
range over which the load I-V curve is symmetric as shown in FIG. 13. For
a given ICTL, the current swing for the composite symmetric load is ICTL
and the voltage swing is VDSAT plus VT.

[0063]It is seen that effective resistance REFF does not decrease
proportionally with increases in control current ICTL. Instead, effective
resistance REFF is approximately proportional to the inverse of the
square root of ICTL. In order to achieve increases in oscillation
frequency FOSC of a delay cell, increasingly large values of ICTL are
required to lower effective resistance REFF. Further, voltage swing
increases with control current ICTL, bringing the delay cell closer to
its tuning range limit.

[0064]FIG. 20 is a diagram that illustrates the relationship of control
current ICTL to frequency of oscillation FOSC as defined by the equation
of FIG. 19. Line 132 describing oscillation frequency FOSC as a function
of control current ICTL in the prior art delay cell becomes increasingly
flat for increased values of control current ICTL. At ICTLMAX, the
corresponding lower swing limit voltage reaches its practical limit,
limiting the oscillation frequency FOSC that can be achieved by the delay
cell.

[0065]FIG. 21 is an equation that describes the effective resistance REFF
of the novel symmetric load. Because the level shift circuit 123 of FIG.
15 holds the lower swing limit voltage constant, voltage swing remains
constant for all values of ICTL. Thus, the voltage swing variable defined
by VDSAT plus VT in FIG. 19 is replaced by a constant voltage swing in
the equation of FIG. 21. The effective resistance REFF of the novel
symmetric load is therefore approximately inversely proportional to ICTL.

[0066]FIG. 22 is a diagram that illustrates the relationship of control
current ICTL to oscillation frequency FOSC in a Current Controlled
Oscillator ICO using the novel symmetric load. As shown in FIG. 21,
effective resistance REFF of the novel symmetric load decreases
proportionally with increases in control current. Thus, oscillation
frequency FOSC increases linearly and proportionally with respect to
control current ICTL, as shown by line 133. Voltage swing, shown by line
134, remains substantially constant. The tuning range of the novel
Current Controlled Oscillator (ICO) 50 of FIG. 9 is therefore not limited
by voltage swing. In one example: 1) the lower swing limit voltage
changes by less than five percent over an FOSC range of more than three
hundred percent, and 2) FOSC varies less than ten percent from a
perfectly linear oscillating frequency to frequency control input signal
(ICTL) relationship over an FOSC range of more than three hundred
percent.

[0067]FIG. 23 is a simplified flowchart of a method 200 in accordance with
one novel aspect. In method 200, a gate-to-source voltage of a
diode-connected transistor of a symmetric load is controlled (step 201)
such that a swing limit voltage of an oscillating signal on a node of the
symmetric load is controlled to have a substantially constant value. In
one example of method 200, the swing limit voltage is a lower swing limit
voltage of an oscillating signal present on node 75 of FIG. 10.

[0068]FIG. 24 is a simplified flowchart of a method 300 in accordance with
one novel aspect. In method 300, a gate-to-source voltage of a
diode-connected transistor of a symmetric load is controlled (step 301)
such that an oscillating frequency (FOSC) of an oscillating signal on a
node of the symmetric load has a substantially linear relationship with
respect to an oscillator input frequency control signal. In one example
of method 300, the oscillating signal is present on node 75 of FIG. 10
whereas the oscillator frequency control signal is input signal ICTL that
is present on input lead 58 of ICO 50 of FIG. 9. The substantially linear
relationship between ICTL and FOSC is illustrated in FIG. 22.

[0069]The techniques described herein may be implemented by various means.
For example, the above-disclosed control techniques for controlling a
gate voltage of a diode-connected transistor of a symmetric load may be
implemented in hardware, firmware, software, or a combination thereof.
For a hardware implementation, the processing units used to perform the
techniques may be implemented within one or more dedicated circuits,
Application Specific Integrated Circuits (ASICs), Digital Signal
Processors (DSPs), Digital Signal Processing Devices (DSPDs),
Programmable Logic Devices (PLDs), Field Programmable Gate Arrays
(FPGAs), processors, controllers, micro-controllers, microprocessors,
electronic devices, other electronic units designed to perform the
functions described herein, a computer, or a combination thereof. For a
firmware and/or software implementation, the techniques may be
implemented with code (e.g., programs, routines, procedures, modules,
functions, instructions, etc.) that performs the functions described
herein. In general, any computer/processor-readable medium tangibly
embodying firmware and/or software code may be used in implementing the
techniques described herein. For example, the firmware and/or software
code may be stored in a memory and executed by a processor. The memory
may be implemented within the processor or may be external to the
processor. The firmware and/or software code may also be stored in a
computer/processor-readable medium such as Random Access Memory (RAM),
Read-Only Memory (ROM), Non-Volatile Random Access Memory (NVRAM),
Programmable Read-Only Memory (PROM), Electrically Erasable PROM
(EEPROM), FLASH memory, floppy disk, Compact Disc (CD), Digital Versatile
Disc (DVD), magnetic or optical data storage device, etc. The code may be
executable by one or more computers/processors and may cause the
computer/processor(s) to perform certain aspects of the functionality
described herein.

[0070]Although certain specific embodiments are described above for
instructional purposes, the teachings of this patent document have
general applicability and are not limited to the specific embodiments
described above. The delay cell of FIG. 10 as controlled by the control
circuit of FIG. 11, for example, has general utility as a delay cell and
sees use in applications other than in oscillators. Also, various ways of
controlling and adjusting the gate-to-source voltage of a diode-connected
transistor of a symmetric load other than the specific embodiment
illustrated in FIGS. 10 and 11 are possible. For example, the size of the
ZVT (zero threshold voltage) transistor may be sized such that its
gate-to-source voltage changes by the same amount as the load P-channel
transistor when the control current ICTL changes. Ways of implementing a
controllable level shifting device other than the specific circuit
illustrated above in FIGS. 10 and 18 are possible. The particular level
shifting circuit of FIGS. 10 and 18 is set forth above as one example of
a suitable circuit. Accordingly, various modifications, adaptations, and
combinations of the various features of the described specific
embodiments can be practiced without departing from the scope of the
claims that are set forth below.