Note: As of 07 June 2009, the Z10040A is replaced by the
Z10040B. The "B" version has the same performance specifications as the "A" but
has several improvements. There is no price change in the "B" version.

Note: The Z10040B will continue to be available. However, a new surface mount
Norton amplifier is now available, the Z10042A. Click here for the
Z10042A page.

The
main changes to the "B" version are:

Balanced or unbalanced input based on connector pins used

All resistors are now 1% metal film instead of 5% carbon film

Two color magnet wire (red and green) are supplied for the bifilar
windings of T1 and T3

Small changes to silk screening to improve legibility

Improved the pad spacing for shielded inductors L1-L4

The Z10040B is a kit (also available assembled and tested)
implementing the Norton noiseless feedback amplifier, as originally developed by
Dr. David Norton, and described at D. Norton and A. Podell, “Transistor
Amplifier with Impedance Matching Transformer,” U.S. Patent 3,891,934, June
1975. (This patent is expired.)

I should mention the Norton amplifier covered in this page is not the
"current difference" Norton amplifier such as in National Semiconductor's
LM3900. Rather, this Norton amplifier is intended for RF amplification, and is a
grounded base design with transformer feedback. The result is a very low noise,
broadband amplifier

The photo below shows an assembled Z10040B Norton amplifier.

The complete Z10040B Assembly and Operating manual may be
downloaded (PDF) by clicking here
or from my Documents page. [If you wish to view the Z10040A manual instead, click
here or go to my
Documents page. Also, the Z10040B manual for printed
circuit board revisions 1 and 2 may be downloaded from the
Documents page.]

The Z10040B modifies Dr. Lankford’s design in several
useful respects:

Automatic input disconnect upon DC
power removal along with over-voltage gas trap protection.

Clifton Laboratories also offers the
Z1202A DC power injector usable with the Z10040B to provide DC power over the
coaxial cable.

These features increase the Z10040B's
survivability and utility compared with an amplifier without these features.
Duplex power avoids the need to run a separate DC power line to a remote
amplifier.

Bypass
Relay Option
The Z10040B, when power is removed, reverts to "safe" mode with the input
grounded. Some purchasers have asked for a different behavior, i.e., when power
is removed, the Z10040B will be bypassed, with the input connected to the
output. The Z10080A bypass relay kit can
be added to a Z10040B to provide bypass on power removed operation.

Specifications Summary
The following specifications are extracted from the Assembly and Operating
manual. I'm still working on measuring a few of the parameters and in some
cases, the Norton amplifier is so good that the normal techniques and equipment
I use are not adequate to measure the performance. The performance values are
for an amplifier with bias adjusted for maximum IP2 performance.

When terminated with 50
ohm load, input VSWR is less than 2:1 over the range 300 KHz – 30 MHz,
and is typically below 1.5:1 over this range.

Common Mode Rejection in balanced input mode

Typically 50 dB at 1 MHz, decreases with increasing frequency.

What do all these numbers mean? Not everyone may be familiar with amplifier
specifications. I've added a (I hope) simple guide to reading an amplifier
specification sheet. The guide is later on this page—click
here to read it.

Because the Norton amplifier's IP2 performance is of great
interest, I've added a page describing how I measure IP2. I measured one Z10040A
at IP2 = 98.5 dBm and a second Z10040A within a dB or two of this figure. Click
Measuring IP2 for my IP2 test procedure.

I've prepared a similar page showing how I measure IP3 for
the Z10040 Norton amplifier. Click Measuring IP3
for the details.

Both kits and assembled amplifiers use matched 2N5109
transistors. I measure the DC current gain of all 2N5109 transistors in my stock
and match pairs within ±5%. I've added a page showing how DC matching
affects the transistor's parameters as seen on a curve tracer.
2N5109 Matching I've intentionally built a
prototype with large mis-matched pairs, and the effect upon the amplifier's
measured performance parameters are not all that significant. Still, using
matched 2N5109's provides an additional measure of assurance that the amplifier
will work as expected when built.

The Z10040B's standard configuration is for 11 dB (nominal) gain, with a
transformer winding ratio of 1:11:4. It's possible to increase the gain to
15.6 dB (nominal, 14.6 dB typical measured) if the transformers are built with
1:29:6 ratios. I've added a new Appendix F to the Z10040B manual with
measured performance data for a 1:29:6 version. Click
here to read the manual—it's a
4.5 Mb PDF file—and go to Appendix F.

As of 21 January 2010, I've added
Service Bulletin
No. 1 for the Z10040B covering potential instability where 2N5109s with
unusually high UHF gain are used. This modification should not be necessary for
any Z10040 amplifiers built or received as it is something I identified while
building an amplifier with a new lot of 2N5109 transistors of exceptionally high
UHF gain and all kits or assembled amplifiers using 2N5109s from this batch have
the modification. However, it is possible that a Z10040 may require replacement
transistors due to component failure in the future, so I have documented the fix
in this service bulletin. All new kits and assembled units will include the
modification as a preventative measure. Any Z10040A or B owners wish to make the
modification should send an E-mail to me and I will provide the two 10pF disc
ceramic capacitors. I will also update the Z10040B's Assembly Manual to reflect
this change within the next few days.

Bias adjusted for
minimum IP2/IP3 (some interaction between the two is usually seen and it
is not generally possible to minimize both IP2 and IP3 at the same bias
setting.) IP2 and IP3 performance data and frequency response data
are provided to the purchaser. This service is available for either
customer-built kits or as an option for assembled and tested units.
Customer is also responsible for return shipping charges.

Indoor enclosure kit

$27.50 (BNC)
$30.00 (UHF)
$35.00 (N)

Die cast
enclosure with BNC, UHF or Type N connectors. Kit price includes the die
cast enclosure, with all holes drilled, your choice of BNC, UHF or N
connectors, power connector, Teflon coaxial cable for jumpers, four 4-40
x 0.5 inch standoffs and associated hardware and a DC power cord.

Indoor enclosure assembled

$32.50 (BNC)
$35.00 (UHF)
$40.00 (N)

Assembled enclosure
(connectors and stand-offs are mounted and pre-wired) with BNC, UHF or N
connectors. Does not include assembly of Z10040B amplifier.

Payment may be made with a check or money
order payable to Clifton Laboratories at the address on the top of this page, or
via PayPal to
orders@cliftonlaboratories.com.

Please do not forget to include shipping and, for
delivery within the Commonwealth of Virginia, sales tax.

Orders are normally acknowledged within 24 hours of
receipt. Please contact me if you fail to receive an acknowledgement.

Is the Z10040B a difficult kit to assemble? I don't
think so, but you should read the
assembly manual and decide whether you can build it. Based on building
several Z10040A boards, you should allow about 1.5 to 2 hours to assemble the
kit up to the point of winding the transformers and completing the kit. Another
1.0 to 1.25 hours should be allowed to wind the four transformers. Thus,
the total time is in the range of 2.5 to 3.25 hours. The most difficult part of
the build is winding the four transformers. I've done my best to explain how to
wind these transformers in the assembly manual.

As
an independent view of how long it takes to assemble a Z10040 and how difficult
it is, a Z10040A purchaser recently noted:

It took me about 5 and half hours, but I wasn't rushing and like to
measure every part before I add it to the board. It's a very nice kit and
works well with my 2N5109 antenna described in the yahoo group. If someone
has experience building a kit with coils, it's just beyond beginner, but
certainly easier than average. I'm building a TAPR TADD-2 this weekend, so
the Z10040 seems really easy by comparison!

There's no significant difference in assembly time or
difficulty between the "A" and "B" versions.

Another builder reports:

I just finished the 2 preamps. I did not
hurry, so it took 2 hours for the first, and 1.5 for the second. The
documentation and the PCB quality are excellent! Congratulations for that,
you did a superb job. I don't think you get any problems, even when you are
not experienced. The transformers need a while, but it is not really
difficult. Using magnetic wire that can be soldered (isolation burns off)
would make things a bit easier/quicker.

After this builder's kit was shipped, I switched to a
different wire source. All kits are now shipped with solderable magnet wire.

Of
course, the Z10040B is also available assembled and with optional performance
measurement data. (A sample performance report can be view by clicking
here.)

With apologies to Bob Pease of National Semiconductor fame, I've borrowed his
"what's this ... stuff anyway" title phrase for this section.

When considering purchasing a preamplifier, several performance parameters
are useful in assessing differences amongst prospective amplifier candidates and
in deciding which best meets your requirements.

I've tried to keep the discussion at a general, introductory level and I hope
those readers familiar with this topic will excuse the simplified treatment. The
discussion also assumes operation at 30 MHz or below, where atmospheric noise is
almost always the limiting factor in usable sensitivity.

Gain

Gain is pretty simple, isn't it? The more gain, the better
as it will boost weak signals more than a lower gain amplifier. There's some
truth in this statement, but not much. In fact, too much gain can create worse
problems than too little gain. And, gain can't be considered in isolation; your
antenna system, receiver and "radio environment" must also be considered.

First, all amplifiers and all receivers, are imperfect devices and hence
create noise and "phantom" signals, to some degree or other. Of course, some
equipment is more susceptible to these problems than others, but none are
completely free from phantom signals under all conditions.

I'll use the term "phantom signal" to mean a signal that is created in the
receiver or amplifier or elsewhere in your radio gear. (We'll look at these
phantom signals in more detail when consider IP2 and IP3.) A phantom signal may
seem real in that you can hear it, and tune through it just like a real signal,
but in fact it's not really being radiated on the frequency your receiver is
tuned to. (In fact phantom signals can be created in nearby objects and radiated
into your receiving system, but since this discussion concerns receivers and
preamplifiers, we'll leave this specialized class of "radiated phantom signals"
for another time and place.)

A very important determinant of the degree to which your system produces
phantom signals is the overall signal input level. All else being equal, the
stronger the input signal (whether to a preamplifier such as the Z10040B) or to
your receiver, the greater the number and strength of phantom signals. Your
receiver may have very good frequency selective filtering to reduce phantom
signals, but if the preamplifier you add itself generates the problem, all the
filtering in the world between the preamplifier and your receiver won't address
the problem. Or, if the preamplifier is relatively flawless, it may well be
that the increased signal level your receiver sees causes it to generate phantom
signals.

All these words are a roundabout way of saying that if your receiver needs
additional gain provided by a preamplifier, the added gain should be kept
to the minimum necessary level. This is absolutely a case where "more is not
always better."

Unfortunately, I can't provide an easy way to determine how much gain is
sufficient and how much is excessive in every particular situation. But, in
general, a preamplifier is used for three reasons:

To overcome coaxial cable or other transmission and ancillary device
(such as filter) loss. In this case, the preamplifier gain should be
sufficient to cancel the loss and perhaps provide a dB or two reserve margin
above the loss. Typically the total gain one needs for this purpose will be
a few dB total, perhaps 6 dB or so at the most under usual conditions.

To help a receiver with inadequate overall gain work with a normal
antenna system. First, it's not a good idea to use a preamplifier to "fix" a
defective receiver. Rather, first repair the receiver and see if a
preamplifier is really necessary. If it is, then the preamplifier should
provide sufficient gain so that when the antenna is connected atmospheric
can be heard in the absence of a signal. A preamplifier with gain much
beyond the level necessary to detect atmospheric noise is not useful.

To make a correctly functioning, sensitive, receiver work with a highly
inefficient antenna. First, one should consider whether a more efficient
antenna system can be constructed. Assuming this is not the case, such
as for certain specialized antenna systems, such as the Beverage, or Dr.
Lankford's various anti-noise antennas, then sufficient preamplifier gain
will be necessary to detect atmospheric noise. Dr. Lankford's
15 foot anti-noise antenna, for example, is commonly used with a Norton
amplifier configured for 12 dB nominal gain, although other amplifiers with
similar gain may be used, of course. Beverage antennas may require slightly
more gain, 15 dB or so, that might be obtained from a Norton amplifier such
as the Z10040B with a 1:19:5 winding ratio. (See the
Z10040B manual for the
relationship between gain and transformer turns.) In some extreme cases, 20
or 30 dB gain may be required, such as a very short antenna. (Short in terms
of wavelength, of course.)

Before adding high gain amplifiers (or even worse,
cascaded high gain amplifiers), however, one should consider
whether a more efficient approach is possible. For example, extracting a
usable signal from an electrically short antenna may be better
accomplished with a high impedance source follower circuit than a 30 dB
gain 50 ohm input impedance amplifier. This is, after all, how a voltage
probe active antenna works. Or, a passive matching network may be in
order.

Amplifier gain is normally stated as the "-3 dB" point. 3
dB is half-power or 70.7% of voltage. An amplifier with 12 dB gain in mid-band,
will still have 9 dB gain at the -3 db point. Depending on the amplifier design,
for frequencies above and below the 3 dB points, gain may drop quickly with
frequency or slowly with frequency. A good amplifier supplier will provide
typical gain versus frequency data. You can then use the gain versus frequency
data to judge whether the amplifier provides useful gain at the frequency of
interest, with "useful" gain being based on the considerations discussed under
"gain" above. If the amplifier is purchased as a kit, such as the Z10040B, it
may be possible to make small component changes to shift the frequency response
up or down. For special low frequency extension (below 100 KHz) of the Z10040B,
please contact Clifton Laboratories.

1 dB Compression Point

An amplifier's gain is a function
of the input signal level. At some input level, an amplifier becomes saturated
and can no longer increase its output power in response to an increase in input
power.

The normal parameter used to quantify this effect is the "1 dB compression
point." The 1 dB compression point is determined by calculating the
amplifier's gain (output power divided by input power, or, in terms of dB,
output power (dBm) minus input power (dBm)) as the amplifier's input signal
level increases. At some input signal level, the amplifier can no longer
perfectly increase its output power for a change in input and the amplifier gain
decreases. At the 1 dB compression point, the amplifier's gain drops 1 dB below
the gain when the input signal level is safely below saturation.

The figure below, taken for the first "production" Z10040A amplifier, shows
the data necessary to compute the 1 dB compression point. The vertical axis is
amplifier gain, with each division equal to 0.2 dB. The top of the graticule
line is 11.13 dB. (I picked this odd value to make the peak gain align with the
2nd graticule line.) The "B" amplifier's performance is identical.

The horizontal scale is amplifier drive in dBm. The left edge is +15 dBm and
the right edge is +20 dBm, with each horizontal division representing 0.5
dBm.

Looking at the plot, we see that for input signal levels
between +15 and +17.5 dBm, the gain is essentially constant. At the 5th graticule line
(corresponding to +17.5 dBm input to the Z10040A), however, we see the gain
begin to drop. The blue arrow shows when the
Z10040A's gain drops 1.0 dB (5 divisions) from the peak linear value. The
corresponding input signal level is +19.45 dBm. The gain at this point is 9.98
dB, so the Z10040A's output power at the 1 dB compression point is +19.45 dBm
+9.98 dB = +29.43 dBm. This is not far below 1 watt output power.

Why is the 1 dB
compression point of interest and what does this value mean? As an amplifier
becomes non-linear, its distortion increases and thus its propensity to produce
phantom signals. All else being equal, the higher the 1 dB gain reduction value,
the less likely the amplifier is to produce phantom signals of significance.

The 1 dB compression point performance of a particular
amplifier should not be considered in isolation. For
example, if your antenna is located in an area of very strong AM broadcast
signals an amplifier with a high 1 dB compression value will be of more value
than if you are far from strong signal sources.

A word on dBm may also be useful. Most of us are familiar with the term
decibel, or dB, as a relationship between two power levels P1 and P2: dB =
10*Log10(P1/P2). The inverse relationship, given dB with the desired
result being the power ratio (PR) is PR= 10(dB/10).

The term dBm is an absolute power level, with the reference
power being 1 milliwatt, or 0.001 watts. Hence, a signal of 0 dBm has 1
milliwatt power, a signal of 29.43 dBm has 10(29.43/10) power,
or 877 milliwatts or 0.877 watts.

One more thing to watch for. Normally the 1 dB compression figure is quoted
with respect to the input signal level. If quoted with respect to the output
signal level, it will be inflated by the amplifier's gain.

Noise Figure

All amplifiers create
noise, some more than
others. Consider an noisy signal consisting of -100 dBm noise and a -90
dBm signal. The input signal to noise ratio is 10 dB, i.e., the signal is
10 dB stronger than the noise.

Suppose a perfect, noiseless amplifier has 20 dB gain. It
will increase the signal by 20 dB to -70 dBm and also amplify the input
noise by 20 dB to -80 dBm. At the amplifier output, therefore, the S/N ratio
remains 10 dB as the signal and the noise are identically increased.

A practical amplifier, however, generates internal noise.
This internally generated noise adds to the amplified input noise and degrades
the output S/N ratio. Noise figure is a method of quantifying the amount of
output S/N degradation caused by the amplifier. (Noise figure applies to things
like attenuators and mixers as well as amplifiers, of course.)

The following discussion is abridged from Hayward, et al.,
Experimental Methods in RF Design. If not on your library shelf, this book
should be.

The noise factor is defined as:

where

F is the noise factor
Nout is the output noise power delivered to the load
Nin is the noise power available from the input resistance
G in the power gain

This definition is in algebraic, not decibel form. For
example, G=100 for an amplifier with 20 dB gain, and the noise power values are
in watts (or microwatts or nanowatts, as you might prefer.)

Nin is the noise power available from the source
resistance at room temperature, considered to be 290 K (or 17 °C or 62.3 °F). If
the amplifier is perfect and contributes no noise of its own, then Nout
= G*Nin and F=1 or 0 dB.

The noise
factor equation can be recast in terms of the input and output signal to noise
ratios:

where
Sin/Nin is the signal-to-noise ratio at the amplifier
input
Sout/Nout is the signal-to-noise ratio at the amplifier
output

As with the earlier equation, signal and noise values are
algebraic, not decibel.

The noise figure is
10log(noise factor) and is in dB.

What noise figure
is needed? A simplistic answer is as low as possible, but in fact the lowest
noise figure amplifier may not be the best in terms of intermodulation
performance or other key factors that are more important for low, medium and
high frequency reception. The Radio Society of Great Britain's Radio
Communications Handbook (8th ed.) notes the following:

Because of galactic, atmospheric and man-made noise
always present on HF, there is little need for a receiver noise figure of
less than 15-17 dB on bands up to about 18-20 MHz, or less than about 10 dB
on 30 MHz, even in quiet sites. It may, however, be an advantage if
the first stages (preamplifier or mixer or post-mixer) have a lower noise
figure since this will permit good reception with an electrically short
antenna or allow the use of a narrow-band filter, which attenuates the
signal power...

As a practical matter, therefore, in most circumstances,
it will be difficult to distinguish between a preamplifier with a noise figure
of 2 dB versus one with a noise figure of 3 or 4 dB when used below 30 MHz.

IP3 and IP2

The
major source of phantom signals created in the amplifier are from
"intermodulation distortion." Intermodulation distortion creates new, unwanted,
phantom signals from combinations of input signals.

If amplifiers only had to handle one simple signal, life
would be easy. Our concerns would be limited to harmonics. However, a typical
preamplifier is faced with an input of hundreds or thousands of individual
signals, some of which may be very strong and others just above the noise level.

The two simplest, yet useful, test protocols quantifying
how an amplifier deals with multiple signals are the 2nd and 3rd order
intermodulation tests. The output of these tests is reduced to two single
numbers, the "second order intercept" or IP2 and the "third order intercept" or
IP3.

The mathematical relationship between the two input
signals on frequencies f1 and f2 and the frequencies of the resultant
intermodulation products is:

where n+m equals the "intermodulation order."

Thus, for second order intermodulation products, possible
values of n and m are:

n=0, m=2
n=1,m=1
n=2,m=0

The corresponding intermodulation product frequencies are:

2f1
f1+f2
f1-f2
2f2

This relationship is shown below, in the illustration from
Anritsu's Application Note Intermodulation Distortion (IMD) Measurements
Using the 37300 Series Vector Network Analyzer.

With third order intermodulation, n+m=3, so we have the
following possible frequency combinations (valid n and m values are 1,2 and
2,1):

2f1+f2
2f1-f2
f1+2f2
f1-2f2

Some of these combinations may yield "negative"
frequencies. It's the absolute value of the frequency difference that counts.

The figure below, also from Anritsu's Application Note
shows the second and third order intermodulation products from a pair of
signals.

To put this mathematical argument in prospective, consider
what happens if f1 and f2 are 980 KHz (f1) and 1400 KHz (f2), typical AM
broadcast band channels.

These two signals, when passed through any practical
amplifier, will combine and create new phantom signals. Specifically:

If you are an amateur radio operator, the phantom signals
at 1960 KHz and 3780 KHz will be objectionable. Or if you are a medium wave
DX'er, you won't care for the 560 KHz IMD signal either. Worse yet, additional,
higher order, IMD products are also created although their amplitude diminishes
with increasing IMD order.

The relationship the input and IMD signals is that for a 1
dB change in the f1 and f2 amplitude levels, the nth order
intermodulation product changes by n dB. Thus, if the two input signals increase
by 10 dB, the 2nd order intermodulation signals increase by 20 dB.
Likewise, the 3rd order intermodulation product signals increase by
30 dB. (The same rule applies to decreases, of course.)

If we test an amplifier by inputting two equal level
signals and vary their levels while measuring the second and third order IMD
products, and then graph the results, the plot will resemble the one
below, extracted from the ARRL's 2006 Radio Amateur's Handbook.

The plot show the effects of gain compression, previously
discussed. It also shows the second and third order IMD levels and how the
increase at different levels. (When speaking of the IMD level, we mean the
level of any one of the products, as in theory all nth order products
are of equal level. In practice, there can be differences in level among the
various individual IMD products comprising the set of nth order
products. In this case one normally takes the strongest IMD product as the
measured value.)

If we project the desired output, and the 2nd and 3rd
order IMD product levels, we see that the projected lines intersect. The
intersection of the desired output and the 3rd order IMD product is
the "third order IMD intercept point" or IP3, and the corresponding 2nd
order intercept point is IP2.

As should be apparent from the plot, it is normally
physically impossible to operate a practical amplifier such that the desired
output has the same level as the third order intermodulation product or the
second order intermodulation product. Practical amplifiers will saturate long
before this occurs. Hence, it is necessary to develop IP2 and IP3 specifications
by projecting or extrapolating measurements taken at lower signal levels.
To this degree, therefore, IP2 and IP3 are fictitious values, although they are
firmly grounded in real data.

I should also mention that not all amplifiers are well behaved and follow the
nice 1:n amplitude relationship. This can cause some difficulty in deciding how
to extrapolate the lower level data. If, for example, the third order IMD
product increases at 2.5 dB for every 1 dB increase in input level over some
range of input levels and at 3.5 dB for other input levels, how then should the
third order intercept be determined?

All this theoretical discussion is well and good, but what
does it mean when it comes time to evaluate a preamplifier? As with the other
specifications, all else being equal, it's better to have an amplifier with
higher IP2 and IP3 performance than lower. Of course, seldom are all things
equal.

We can, from IP2 and IP3 intercept data,
estimate the level of interference expected from a pair of input signals. (And,
of course, all this discussion is greatly simplified because seldom is it
the case that only two signals are the cause of IMD and that even rarer
will it be the case that the two signals are of equal amplitude. It is possible
to extend these prediction techniques to multiple signals of arbitrary levels,
but that's way beyond the purpose of this discussion.)

Suppose we are considering two amplifiers, A1 and
A2, both with 10 dB gain. A1 has an IP3 of +15 dBm and A2 has an IP3 of +25 dBm.
The two interfering signals are local AM broadcast stations at 980 KHz and 1400
KHz and the measured signal at the amplifier input of both of these stations is
-20 dBm, very strong signals, 53 dB over S9 assuming S9 is 50 microvolts an a 50
ohm system. After passing through either A1 or A2, the signal level is -10 dbm.

Consider amplifier A1, with an IP3 of +15 dBm. At the IP3
point, both 980 and 1400 KHz fundamental signals and the 3rd order
IMD product would be at +15 dBm. However, in fact the f1 and f2 output signals
are at -10 dBm, or 25 dB below A1's IP3 point. Since 3rd order
intermodulation interference products drop 3 db for every 1 db reduction in
input level, the 3rd order products will be at 15 dBm - 3 * 25 dB =
-60 dBm.

Performing the same calculation for A2, we note the f1 and
f2 signals are 35 dB below A2's IP3 point and hence the resulting 3rd
order IMD products will be 3 * 35, or -105 dB from the IP3 point, corresponding
to a 3rd order interference level of -80 dBm.

A shortcut to this comparison is to simply note that the nth
order IMD product level difference between two amplifiers with identical gains
is (n-1) times the difference in IP3 levels in dB. In this example, n=3 and the
difference in IP3 levels is 10 dB. Hence the amplifier with +25 dBm IP3 will
have 20 dB lower 3rd order IMD products than one with an IP3 of +15 dBm.

To help put this in prospective, the strongest medium wave
AM broadcast signal I see using an 80 meter inverted vee antenna in my suburban
Washington DC location is -18 dBm, not far from our sample calculation.

The spectrum analyzer plot below shows the medium wave
band in mid-morning on January 2009.

The spectrum analyzer image below, in contrast is from Ron,
K8AQC, who lives in suburban Detroit, a couple miles north of a 50 KW AM
directional station at 1500 KHz. That signal is -0.4 dBm into a short whip
receiving antenna. In addition there are two stations at -10 dbm and a total of
7 stations at -20 dBm or stronger. That's a much denser RF environment than I
experience.

The data below is from the production Z10040A printed circuit
board and should be fully representative of assembled kits or wired and tested
amplifiers. These are not warranted performance values, however. The Z10040B has
identical performance.

Gain versus Frequency, direct DC power feed.

Gain versus Frequency, duplex DC power feed with Z1202A
Power Coupler

Input VSWR versus frequency, 50 ohm system

1 dB Compression Point

IP3 Typical

IP3 intercept computed from this
measurement is +46 dBm. This number is subject to revision when I complete a
better test setup.

Noise Figure

Measured with
an HP8970A noise figure meter and AIL 7615 noise source. Noise figure
performance varies significantly between units and with supply voltage and the figure below is
representative of only a limited sample set.