Abstract:

In an OFDM system a plurality of subcarriers interferes with a considered
subcarrier in case of carrier frequency offsets. A method and a
corresponding receiver topology are disclosed for reducing the
interference caused by frequency offsets of the subcarriers, wherein in a
decision feedback equalizer the resources for computing the interference
of subcarriers are adaptively allocated, such that only the most
interfering subcarriers are considered when subtracting the interfering
symbols from considered subcarrier symbols.

Claims:

1. A method for reducing symbol detection error in an OFDM communication
system comprising the steps ofreceiving a plurality of OFDM subcarrier
signals in the receiver,estimating symbols of the subcarriers,estimating
an interference on one subcarrier caused by received
subcarriers,selecting from the plurality of received subcarriers at least
the most interfering subcarrier,subtracting the estimated interference of
the at least selected subcarrier from the received subcarrier signals
thus producing subcarrier signals of reduced interference, andestimating
symbols of the subcarrier signals of reduced interference.

2. The method of claim 1, further comprising the step of estimating the
channel characteristics for each received subcarrier.

3. The method of claim 2, wherein the step of estimating an interference
on the one subcarrier is based on the estimated channel characteristics.

4. The method of claim 1, wherein the step of estimating symbols of
subcarriers comprises spatial detection of subcarriers.

5. The method of claim 1, wherein the step of selecting from the plurality
of received subcarriers at least the most interfering subcarrier
comprises the step of calculating the accumulated interference power of
subcarriers, calculating the ratio of interference power of single
subcarriers to the accumulated interference power, and selecting
subcarriers exhibiting the highest ratio.

6. The method of claim 1, further comprising the step of dynamically
allocating processing resources corresponding to the number of selected
subcarriers.

7. The method of claim 1, wherein the OFDM system is a cell phone system.

8. A receiver in an OFDM communication system comprising a decision
feedback equalizer (DFE) for reducing the interference on one of a
plurality of received subcarriers caused by subcarriers, wherein the DFE
forward path comprises means for estimating symbols of a plurality of
received subcarriers, and wherein the DFE feedback path comprises means
for estimating the interference of subcarriers on a single subcarrier and
for selecting at least the most interfering subcarrier, and comprising
means for subtracting the at least most interfering estimates from a
received subcarrier.

9. The receiver of claim 8, wherein the feedback path comprises a feedback
filter for estimating the interference of subcarriers on a single
subcarrier and a subcarrier selection unit communicatively coupled to the
feedback filter, wherein the subcarrier selection unit is communicatively
coupled to the output of the forward path.

10. The receiver of claim 8, further comprising means for estimating the
frequency offset of a subcarrier.

11. The receiver of claim 8, further comprising a channel estimation block
for estimating an effective channel of a subcarrier, and wherein the
channel estimation block is communicatively coupled to the feedback
filter in the feedback path.

Description:

BACKGROUND OF THE INVENTION

[0001]The invention relates to a method and a corresponding receiver for
compensating carrier frequency offsets in OFDM systems.

[0002]Since the advent of the orthogonal frequency division modulation
(OFDM) scheme, it has been used in various transmission systems for
dealing with so-called frequency selective fading and for transmitting
large amounts of digital data. In an OFDM scheme an available frequency
band of limited bandwidth is split up into a plurality of l smaller
frequency bands, wherein each of the smaller frequency bands forms a
so-called subchannel. The center frequency of each subchannel is used as
carrier, a so-called subcarrier, wherein the modulation is chosen such
that the modulated subcarrier wave does not exceed the bandwidth of the
subchannel.

[0003]In this way each of the l subchannels can be used for transmitting
data by modulating the subcarriers using a conventional modulation method
at a comparatively low symbol rate, such that the modulated subcarrier
does not exceed the bandwidth of the subchannel band. As a consequence
the duration for transmitting one symbol over a subchannel is extended by
factor l in comparison to a single carrier transmission system using the
entire available bandwidth. Due to the longer duration for transmitting
one symbol, each subcarrier's signal is likely to remain unaffected by
multipath propagation.

[0004]Each of the l subchannels can be used independently. Hence, the data
to be transmitted via the subchannels may originate from a single source
or from a plurality of independent sources. In case the data originate
from a single source, the single data stream is passed through a
serial-to-parallel converter for splitting the single input stream into a
plurality of l parallel output streams for transmitting these streams at
the same time through a corresponding plurality of l parallel
subchannels.

[0005]Alternatively the data to be transmitted may originate from a
plurality of independent sources, such that each of the subchannels can
be used for transmitting unrelated data. Hence, each subcarrier may be
used by a different transmitter for transmitting arbitrary data.

[0006]In still another embodiment the OFDM system can be a cell phone
system comprising a plurality of cells, each provided by a receiver
station, i.e. a so-called base station or node-B. The available frequency
band in the cell is divided into a plurality of subchannels forming an
OFDM transmission system, wherein the center frequency of each subchannel
may be used as carrier frequency, i.e. a subcarrier. A plurality of
independent transmitters, which may be cell phones or PDAs or other
computers, are assigned to the receiver station for data exchange. The
plurality of independently operating transmitter stations use the
plurality of subcarriers, wherein more than one transmitter station may
use one subcarrier simultaneously, i.e. one subcarrier may be used by
more than one transmitter station. At the receiver station, transmitter
stations, i.e. the signals of the transmitters using the same subcarrier
and the same symbol set can be distinguished using spatial layer
detection. Spatial layer detection schemes make use of multipath
propagation for distinguishing the transmitters.

[0007]In any case, i.e. regardless of the data origin, the plurality of
subcarrier frequencies--in an ideal system--is modulated such that the
subcarriers do not interfere with each other, i.e. any inter channel or
intercarrier interference (ICI) is avoided.

[0008]At the receiver side a superposition of complex OFDM signals is
received, wherein the received signal is a superposition of the
subcarrier signals superimposed by noise and other interfering signals.
In the receiver the received OFDM signal is demultiplexed into the
plurality of subcarrier frequency bands. Each modulated subcarrier is
then demodulated to generate a data stream from each of the subcarriers.
In case the transmitted data originate from a single data source, the
data streams related to the subcarriers may be recombined to from a data
stream corresponding to the original data source. In case the subcarriers
originate from independent sources, the demodulated independent data
streams may be further processed independently.

[0009]This scheme of splitting an available frequency bandwidth into a
plurality of subchannels and using each subchannel for transmitting data
has many advantages and has been improved during the last decades. For
example the implementation complexity has been reduced by deploying the
fast Fourier transformation (FFT) and the corresponding inverse fast
Fourier trans (IFFT), which allows to modulate a plurality of subchannels
onto the subcarriers in a single processing step.

[0010]The subcarriers are called orthogonal if--under ideal
conditions--they do not interfere with each other, i.e. if there is no
inter channel interference (ICI). The frequencies of the subchannels and
their bandwidth are chosen non-overlapping. As a consequence in the
frequency domain all spectra of the subcarriers exhibit zero-crossings at
all of the neighboring subcarrier frequencies. Consequently all of these
subcarriers are orthogonal to each other, i.e. they do not interfere with
each other.

[0011]The property of ideal orthogonality requires that their frequency
bands do not overlap. However, this property exists in ideal examples
only. In real world applications there may be a frequency offset between
the subcarrier frequencies. As a consequence the subcarriers are not
perfectly orthogonal to each other, such that the subcarriers interfere
with each other.

[0012]Offsets between orthogonal subcarriers, such that these are not
perfectly orthogonal but interfere with each other, may have different
origins. In one embodiment, which particularly is related to wireless
communications and wherein the communication system implements an OFDM
system, each mobile terminal uses at least one subcarrier and employs its
own local oscillator. The output frequencies of these oscillators may
deviate from an ideal frequency for example due to production variances.
Another source for carrier frequency offsets is the Doppler effect caused
by moving mobile stations.

[0013]These offsets for example can be prevented by using synchronization
means. In one example each oscillator can be synchronized to a precise
and available clock signal, for example that of the global positioning
system (GPS). Alternatively the IEEE 1588 time protocol could be used to
synchronize an oscillator. In any case, employing any one of these
additional means requires additional effort and costs.

[0014]Accordingly there is a need for a reliable and cost effective
solution to handle the impact of carrier frequency offsets.

BRIEF DESCRIPTION OF THE FIGURES

[0015]FIG. 1 depicts a topology for implementing the proposed method

DETAILED DESCRIPTION OF THE INVENTION

[0016]The present invention will now be described in detail with reference
to a few preferred embodiments thereof as illustrated in the accompanying
drawing. In the following description, numerous specific details are set
forth in order to provide a thorough understanding of the present
invention. It will be apparent, however, to one skilled in the art, that
the present invention may be practiced without some or all of these
specific details. In other instances, well known processes and steps have
not been described in detail in order not to unnecessarily obscure the
present invention.

[0017]FIG. 1 depicts a first embodiment illustrating a topology 100 for
wireless transmission of data, comprising a plurality of a number of K
transmitter stations 110 to 111, wherein one station is indexed by k. In
one embodiment the transmitters 110, 111 may be mobile stations or fixed
transmitter stations, i.e. stations that do not move. Mobile stations may
be cell phones or PDAs or movable laptop computers or any other equipment
capable of communicating with the receiving station. The fixed stations
may be PCs, which usually do not move during operation, i.e. during
communication with the receiving station.

[0018]A plurality of M receiver stations 120, 121 receives the signals
transmitted by the K transmitter stations 110 to 111. A receiver station
or a receiving branch of the receiving entity is indexed by m. Each of
the receiver stations receives the signals transmitted by each of the K
transmitters. Accordingly each receiver station may receive K signals
simultaneously. Generally the receiver stations 120, 121 may form part of
any arbitrary OFDM system. In one embodiment the receiver stations may be
base stations of a cell phone system, in which a plurality of mobile
users, i.e. each with a cell phone or equivalent equipment, transmits
data to at least one base station.

[0019]Note that although the description describes the transmission of
information from stations 110, 111 to receiving stations 120, 121 a
person skilled in the art is well aware that communication may be
performed also in the opposite transmission direction, i.e. from a base
station to a particular transmitter station.

[0020]According to the general understanding of an OFDM system the
transmission of information from the stations 110, 110 to the receiver
stations 120, 121 is separated by frequency. That is, each transmitter
station 110, 111 employs one of a plurality of D orthogonal frequencies.
These frequencies are called subcarrier frequencies, or simply
subcarriers. In an ideal system the subcarriers do not interfere with
each other, i.e. they are perfectly orthogonal.

[0021]However, as mentioned above, transmitters may operate independently.
An individual transmitter may accordingly transmit its individual signal
at an arbitrary time. Consequently, in the plurality of K transmitters at
least two or more of them may transmit at the same time. In that case
each of the M receiver stations will receive a plurality of these signals
at the same time.

[0022]Note that--as mentioned above--more than one transmitter station may
use a subcarrier at the same time, i.e. the described communications
system may be a multiuser MIMO OFDM system. These transmitter signals are
separated at the receiving entity using conventional spatial layer
detection. In the following a subcarrier is denoted by l, wherein
l.di-elect cons.D,

[0023]Each of the transmitters 110, 111 comprises a local oscillator
providing the individual carrier frequency for said transmitter, which
for the k-th transmitter may be denoted as ejφk,T. Each
local oscillator may deviate from the tuned subcarrier frequency, for
example due to production tolerances. Similar to a transmitter, each
receiving station may have its local oscillator, which for the m-th
transmitter or m-th receiving branch of the M receivers is given as
ejφm,R. The provided carrier frequency and accordingly the
signal as received by the receiving stations 120, 121 may have a carrier
frequency offset, abbreviated as CFO. The resulting phase rotation in
time domain between the k-th transmitter T and the m-th receiver branch R
accordingly is ej(φk,T.sup.-φm,R.sup.).

[0024]Another source causing a carrier frequency offset at the receiving
station may be the motion of a mobile station, i.e. the Doppler effect
caused by the motion of the transmitter.

[0025]Each of the spatial links in the OFDM system accordingly may exhibit
an individual carrier frequency offset, such that each of the receiving
stations has to process a plurality of subcarrier frequencies each having
an individual carrier frequency offset. As a consequence of the plurality
of offsets, the subcarriers are not perfectly orthogonal, such that they
interfere with each other. That is, each subcarrier differs from its
ideal frequency thus distorting other subcarriers and at the same time is
distorted itself by other subcarriers, which also deviate from their
carrier frequency by a frequency offset.

[0026]The processing of the received subcarriers at the receiver station
120 aims at reproducing the transmitted data from the received
subcarriers as best as possible while taking into account, that the
subcarriers are not perfectly orthogonal to each other, the signals thus
interfering with each other.

[0027]In order to optimize the processing of the received subcarrier
frequencies the receiver station employs a method for eliminating or at
least reducing the interference on a subcarrier signal caused by the
frequency offset of at least one other subcarrier. In a first step the
interference of subcarrier frequencies caused on one particular
subcarrier frequency is calculated. Subsequently the carrier frequencies
exhibiting the strongest interferences are determined. These carrier
frequencies are then selected to be taken into account for further
processing. In particular the subcarrier frequencies exhibiting the
biggest carrier frequency offsets and accordingly strongest interference
on the one desired subcarrier are determined, and the interference of
these determined subcarriers is subtracted from a desired frequency for
improving the processing of the received desired signal.

[0028]In the following said method is described with reference to block
130, which depicts a block diagram of a central processing entity
comprising a carrier frequency offset compensation block 140 for
performing the method. In this topology a plurality of M receiver
stations 120, 121 is communicatively coupled to one central processing
entity 130, which processes the received signals of a plurality of
receiving stations. Generally the method as described hereinafter may be
performed also in other topologies, wherein for example one processing
entity 130 is coupled to only one receiving station 120.

[0029]Note that receiver stations 120, 121 do not individually demodulate
received signals. Instead each base station forwards its received signals
to a central processing entity 130. In one embodiment a receiver station
may be implemented by a so-called base station (BS), wherein a base
station may perform some basic processing of a received signal such as
baseband processing.

[0030]Each of the receiver stations 120, 121 passes synchronization data
150 to a synchronization unit 160 for further processing, wherein
synchronization unit 160 may comprise a processing block 161 for
estimating the carrier frequency offset, an FFT processing block 162 for
computing a Fast Fourier Transformation of the received signal and a
Channel Estimator block 163 for estimating channel coefficients.
Synchronization unit 160 accordingly passes synchronization data, i.e.
the carrier frequency offsets of received subcarriers and estimated
channel coefficients to block 140 in a frequency domain representation.
Synchronization data may comprise so-called protocol overhead data, i.e.
in one embodiment training data for training equalizers in the receivers.
The synchronization units estimate channel coefficients of all allocated
subcarriers and pass this information to block 140, i.e. to the Joint
Detection and Carrier Frequency Offset Compensation block 140, in
particular to Feedback filter 145 comprised in said block 140. Feedback
filter 145 accordingly receives information about channel characteristics
of all considered channels, wherein two or more channels may be related
to one subcarrier.

[0031]Frequency offset estimator block 161 estimates the frequency offset
of allocated subcarriers and passes the estimated frequency offset of the
received subcarriers to Fast Fourier Transformation block 162 and also to
Feedback Filter block 145.

[0035]The forward path processes the user data in frequency domain
representation to estimate the user data, that a user intentionally
transmitted, i.e. the payload data a user transmitted. Said user data in
one embodiment may be represented by the I/Q values of a symbol in a
signal constellation or may be represented by the bits, wherein the bits
in a conventional way can be mapped to symbols of a signal constellation.
Output 180 of the forward path of the DFE accordingly is the estimated
user data, which may be in frequency domain representation.

[0036]Said forward path of the DFE in one embodiment may comprise an adder
141, a spatial layer detection and common phase error compensation block
142 and an estimator block 143. The user data in frequency domain
representation are coupled to adder 141, which is furthermore
communicatively coupled to feedback filter 145, for subtracting output
from said filter from the user data. Adder 141 then forwards
the--manipulated--user data to spatial layer detection and common phase
error compensation block 142, which determines the user data symbols from
the received subcarriers.

[0037]As explained later on, adder 141 also receives input from feedback
filter 145 for subtracting that from the user data as received from FFT
block 131. Adder 141 passes its output to a spatial layer detection and
common phase error compensation block 142.

[0038]Spatial layer detection and common phase error compensation block
142 processes its received input to determine the user data, i.e. the
symbols as transmitted by each subcarrier. Block 142 furthermore receives
frequency offsets and estimated channel coefficients of subcarriers from
synchronization block 160, either via a direct coupling or--as
illustrated in the figure--via feedback filter 145. Based on the
information provided from filter 145 and on the received user data block
142 compensates a common phase error of received user data and equalizes
user data transmitted on subcarriers. In one embodiment, when the
communication system is a multiuser MIMO-OFDM system, block 142 also
performs a spatial layer detection on its received input to detect user
data transmitted on the same subcarrier.

[0039]The user data detected from the subcarriers, i.e. symbol estimates
of user data, is then passed to estimator block 143 for determining the
most probable user data. In one embodiment estimator 143 in a
conventional way calculates the smallest Euclidean distance between a
transmitted symbol and symbols of a used signal constellation. The user
data, i.e. the symbols as determined by estimator 143, then is forwarded
as indicated by the arrow 180 to arbitrary processing blocks.

[0040]In the following description it is assumed estimator block 143
outputs the user data as symbols, i.e. as in-phase I and quadrature
values Q of the symbols. As these I/Q values can be mapped easily to a
sequence of bits by applying a signal constellation to the I/Q values of
a symbol, a representation of the user data as a bit sequence is
considered to be equivalent.

[0041]Besides being output the estimated user data is forwarded to
subcarrier selection block 144 as input. Based on the carrier frequency
offset as provided by block 161 and based on the provided user data
symbols, subcarrier selection block 144 calculates the interference of
subcarriers related to a considered subcarrier, selects the most
interfering subcarrier symbols and forwards only the selected subset of
subcarrier symbols to feedback filter 145 for further processing. More
particularly block 144 calculates the interference power of all
subcarriers taken as input and also calculates the total interference
power of all subcarriers. Based on these calculations the subcarriers
exhibiting the highest relative interference power are selected as the
most interfering subcarriers. Subcarrier selection block 144 forwards
only the selected subcarriers to filter 145 for further processing.

[0042]In one embodiment the most adjacent subcarriers are considered for
calculating the most interfering subcarriers with regard to a considered
subcarrier. Accordingly the selection of subcarriers forwarded may be all
or a selection of the most adjacent subcarriers.

[0043]As indicated by the switches drawn between subcarrier selection
block 144 and feedback filter 145 only a subset of the subcarrier symbols
as output by estimator 143 is forwarded to feedback filter 145. Based on
these selected symbols and based on the carrier frequency offsets as
provided by block 161 and based on the channel coefficients as provided
by channel estimator block 163, feedback filter 145 calculates the
interfering signals in frequency domain representation. Since only a
subset of the subcarrier symbols is passed to filter 145, the processing
power required for the calculation of the interfering signals can be
dynamically allocated in order to optimize system performance of the
receiver.

[0044]Vice versa the number of considered subcarriers can be adjusted
according to an available and allocated processing power. In systems
having limited processing power, a predefined quota of processing power
may be allocated for calculating interfering signals in filter 145, and
selection block 144 accordingly selects a number of most interfering
subcarrier symbols, thus achieving an optimal equalization.

[0045]The signals calculated by filter 145 are passed to adder 141, which
subtracts these interfering signals from the signals as received from FFT
block 131. In this way the input of processing block 142 exhibits reduced
interference, wherein particularly interference caused by carrier
frequency offsets (CFOs) is reduced.

[0046]In one embodiment a plurality of operation cycles may be performed,
such that reducing the interference of the user data is reduced
iteratively.

[0047]The decision feedback loop comprising selection block 144 and filter
145 in this way receives a plurality of OFDM subcarrier signals,
estimates the symbols of user data of the subcarriers, estimates an
interference on one subcarrier caused by received subcarriers, selects
from the plurality of received subcarriers at least the most interfering
subcarrier and subtracts the estimated interference of the at least
selected subcarrier from the one received subcarrier signal.

[0048]By subtracting the calculated interfering signals from the user data
signals as received from FFT block 131, adder 141 provides manipulated
user data to block 142. Based on these manipulated user data the
processing blocks in the forward path of the DFE again determine symbols
of the user data, wherein these symbols exhibit less distortion caused by
carrier frequency offsets. Accordingly estimator block 143 is capable to
estimate the symbols more accurately, while at the same time the
processing resources available in central processing unit 130,
particularly in the joint detection and carrier frequency offset
compensation block 140.

[0049]In this way the described feedback loop describes a decision
feedback equalizer (DFE) for eliminating or at least reducing the
interchannel interference caused by carrier frequency offsets in
subcarriers of an OFDM communication system by performing the steps of
receiving a plurality of OFDM subcarrier signals in the receiver,
estimating symbols of the subcarriers, estimating an interference on one
subcarrier caused by received subcarriers, selecting from the plurality
of received subcarriers at least the most interfering subcarrier,
subtracting the estimated interference of the at least selected
subcarrier from the received subcarrier signals thus producing subcarrier
signals of reduced interference, and estimating symbols of the subcarrier
signals of reduced interference.

[0050]In the following the mathematical background for the described
method for eliminating or at least reducing the interference caused by
the carrier frequency offsets is described.

[0051]The described topology may be considered a general OFDM system
wherein a number of K active users, each denoted by an index k,
simultaneously transmit data on a subset of a number of D subcarriers,
then the time domain representation on the base band signal for a
particular user k can be denoted as an inverse discrete Fourier
transformation (IDFT)

[0052]wherein Xik[l]=0 .A-inverted. lD and N represents the DFT
size. The OFDM symbol index is given by i=0, . . . , NS. After the
transmission over the channel, wherein the channel can be described by
its impulse response vector h consisting of NCIR discrete channel
taps, the signal of the i-th symbol at the m-th receiver branch is

[0054]The phase rotation error between the up and down conversion process
on the link between the k-th user and the m-th receiver station is
Δφm,k=φk-φm. φ in general is
referred to as phase noise, whereas here we relate to the carrier
frequency offset (CFO) impact such that Δφim,k[n] can
be defined as a linear phase process for each link as

Δφim,k[n]=2π Δfm,k n
TS+φ0m,k (3)

[0055]For a general system model the carrier frequency offset CFO
Δfm,k is normalized to subcarrier spacing BSC which leads
with nTS=n/NBSC to

[0056]If OFDM symbols are transmitted consecutively φ0 is given
as φ0=Δφi-1m,k[N-1]. The channel impulse
response hm,k is modeled such that

σ h 2 = ε { h λ m , k 2 } ,
##EQU00004##

wherein σh2 is the mean power of the channel taps as
determined by channel estimator block 163, where the
h.sub.λm,k are independent and identically Rayleigh
distributed channel coefficients. In the following the OFDM symbol index
i is omitted. With l.di-elect cons.D the received signal at the l-th
subcarrier in the frequency domain is obtained by a discrete Fourier
transformation (DFT), that is

[0057]Subsequently vector matrix notation is used to simplify the
transmission model. With Fourier transform matrix F and the link CFO
matrix Φm,k=diag(ejΔφm,k.sup.[n]), where
Φm,k .di-elect cons..sup.×N, equation (5) can be written
as

[0058]In this way user data symbols in frequency domain representation
Ym can be calculated in the forward path of the decision feedback
equalizer, i.e. in blocks 142 and 143 according to equation (7) or (8),
wherein Em,k represents the estimated channel offset of the k-th
user at the m-th receiver and Hm,k represents the estimated channel
of the k-th user to the m-th receiver, Xk represents the symbol sent
by the k-th user over said channel and wm represents the Gaussian
noise of said channel at the m-th receiver.

[0059]With FFH=IN, the unitary similarity transformation of
Φm,k is used and the elements in Em,k are stated as

[0061]with κ=Δεm,k+p-l .A-inverted. l,p=1 . . .
N, wherein l represents the index of the subcarrier of interest and p
represents an adjacent subcarrier impacting the l-th subcarrier. Based on
this model, the frequency domain transmission of equation (6) can be
rewritten for the l-th subcarrier to

[0062]In equation (12) {tilde over (H)}m,k[l,p] represents the
effective transmission channel in the frequency domain. The first term of
equation (12) represents the inter channel interference (ICI) free
spatial multiplexing signal transmission with a user common phase error
(CPE) term Em,k[l,l] which causes an additional phase increment of
the channel of one OFDM symbol. The second term of equation (12), denoted
YICI, includes self and multiuser ICI, which destroys the
orthogonality of the subcarrier symbols Xk.left brkt-bot.l.right
brkt-bot. and leads to amplitude and phase errors of received symbols.

[0063]Considering the signal to interference and noise ratio (SINR), term
YICI of equation (12) represents the impact of the intercarrier
interference of all other carrier frequencies on the l-th received
subcarrier of receiver branch m. As mentioned above in equations (7) and
(8) in case of randomized transmit symbols the first term in equation
(12) is uncorrelated to YICI. Accordingly the SINR on the l-th
subcarrier at the m-th receiver branch is given as

[0064]In the numerator the common phase error (CPE) in equation (13) is
fully compensated by the receiver algorithms. For the ICI power
estimation of γlm,k results can be calculated using a
conventional method wherein the ICI power is a sum over the interference
power from adjacent subcarriers affecting subcarrier 1 with mean channel
power and mean transmitted signal power. γlm,k can be
expressed as ε{YICI YICIH} as

wherein Δεm,k is an approximately quadratic factor.
Equation (16) gives a convenient term to be evaluated in algorithms for
compensating carrier frequency offsets. In particular the calculation of
the interference power of the l-th subcarrier can be calculated according
to this equation in subcarrier selection block 144.

[0068]Further, from equation (13) it is derived that in a multi-user MIMO
system the interference power of each link superimposes the other links,
i.e. in case of a carrier frequency offset between the carrier
frequencies the interference power of each carrier frequency superimposes
on each other carrier frequency, and that the resulting approximated ICI
power at one receiver branch can be obtained by

[0070]Equation (13) furthermore reveals that the joint noise and
interference term at each subcarrier comprises
σ.sub.Y,m2=σICI,m2+σW2, that
is the joint noise and interference comprises the average white Gaussian
noise power and the interference power of the l-th subcarrier. Further,
equation (16) reveals that the interchannel interference noise
σICI,m2 can vary according to the distribution
Δεm,k So for each receiver branch m or receiver
station m there is another effective noise term σ.sub.Y,m2.

[0071]As the mentioned above the transmitted data must be detected in the
forward path of the DFE circuit, i.e. in particular the transmitted
symbols are estimated in estimator 143, and the carrier frequency offsets
(CFOs) must be detected for subsequently determining the interference
impact caused by the CFOs.

[0072]For detecting the symbols the spatial layers must be detected, which
requires the knowledge of the frequency errors at the receiving entity.
The estimation of the multi-user CFO can be handled conventionally.
Various conventional methods are known, also for a multi-user
asynchronous OFDMA scheme. According to equation (10), the CPE can be
seen and treated as mean phase rotation during one OFDM symbol. With
equations (4) and (10) the CPE Δ0 of the i-th OFDM symbol with
respect to the CP on each link can be denoted as

[0073]As a result for every link on different subcarriers an equal but
over more than one OFDM symbol a changing CPE occurs. Normally, this is
an additional phase offset on each OFDM symbol, which will be inherently
estimated by the channel estimation process and will also be taken into
account during the equalization process by estimating the spatial
decorrelation filter G. Alternatively the CPE impact on the channel can
be directly obtained from the multiuser CFO estimates and can be
developed conventionally into a time series.

[0074]In the following perfect knowledge of the CFO and CPE for each link
is assumed.

[0075]Based on equation (12) two possible user layer detection strategies
are possible as described in the following, wherein the following
description and mathematical derivations are always denoted for the l-th
subcarrier and for a number of K users and a number of M receiver
stations.

[0076]In one embodiment the spatial layer detection is performed with
common phase error (CPE) correction. As mentioned above intercarrier
interference at one particular receiving branch m can be interpreted and
treated as additional noise with σICI,m2. Apparently in
the case of low ICI the thermal noise floor is stronger than the portion
caused by the ICI. In cellular systems, such as in cell phone systems, we
suppose that the CFO is partially compensated at the terminal side, such
that the assumption of low ICI holds. However, the CPE phase de-rotation
cannot be neglected as the phase error leads to performance degradations.
With equation (18) a matrix containing all link CPEs in the i--the OFDM
symbol can be derived from the measured CFOs, which is

[0077]Δθi is inherently superimposed thus leading to the
effective channel {tilde over (H)}. Due to the nature of the phase
difference matrix, the time variant channel can be decomposed in a static
part and a dynamic phase rotation part. Then we can simplify the channel
inversion having the consequence that due to the static part a continuous
tracking is not necessary.

the effective channel can be inverted in a low complexity manner. Note
that mathematical operator ".", i.e. a dot, denotes the element wise
matrix product. Hence, multiuser detection filters can be applied with
restriction to the coupled channel model according to transmission
equation

where {tilde over (W)} is the effective noise vector and {circumflex over
(X)}i represents an estimation of a sent i-th symbol. It is known
that the maximum likelihood detector yields the optimum results by
minimizing the error of Y-H S. Therefore, the optimum detector can be
achieved using the equation

wherein σ.sub.Y,m2 represents an effective noise vector with
elements for each receiver station. With equation (23) after extraction
of the k-th row of G and the k-th column of θiH, it is
possible to equalize the transmit symbols of the k-th user {tilde over
(X)}ik using

{tilde over
(X)}ik=GT[k;:]ΔθiH[:;k]Yi
(26)

wherein GT.left brkt-bot.k;:.right brkt-bot. denotes all elements in
the k-th row of GT and θiH.left brkt-top.:;k.right
brkt-bot. denotes all elements of the k-th column of 0iH.

[0080]This equation directly allows successive interference cancellation
of the spatial layers wherein already detected spatial symbols
Xik are subtracted from the received signals iteratively as
described above in a plurality of iteration cycles, that is

wherein z denotes a desired user on subcarrier l and index k denotes other
users using the same subcarrier l, and wherein {tilde over
(X)}=Q({circumflex over (X)}) implements the symbol decision of the
current user data to avoid error propagation at the interference
reduction stage. This can be accomplished by decoding the information
bits and remodulate the estimated symbols. In this context robust
decoding techniques can be exploited to increase the system performance.
For example the GENIE approach is used here where always the perfect
known symbols are canceled. Furthermore, depending on the largest layer
SINR, the order of the detection process can be modified, which results
in an additional permutation of the user layers.

[0081]The detection of the layer can be performed including the correction
or at least the reduction of inter channel interference (ICI). Based on
the first signal detection step it is further possible to reduce the
remaining interference. Already detected symbols will be remodulated and
feed back in order to estimate the intercarrier interference YICI,
which is subtracted before the channel equalization is applied again.

[0082]In a system with limited processing resources in the feedback path
only adjacent, but strongest interfering, subcarriers are taken into
account for reducing the interchannel interference, wherein the selection
of considered subcarriers depends on

α [ m , k ] = γ l m , k σ ICI , m 2
α max ( 28 ) ##EQU00020##

wherein α is the number of considered interfering subcarriers and
αm,k=0,1, . . . , N/2-1, αmax.di-elect
cons.[0:1:(N/2-1)KM].

[0083]Considering now above given equations (12) and (23), the iterative
scalable decision feedback equalization (DFE) scheme can be written as

as the ICI estimate on the l-th subcarrier and t as iteration numbering.
For further simplification the processing entity, i.e. the processing
resource in the feedback path of the DFE, can evaluate
γlm,k and σICI,m2 fairly easily, if the
linear approximations derived as above are used, i.e. particularly when
under consideration of equation 16 as above. Note that the noise variance
in Gt changes to (σw2/σs2)IM in
the interference reduction steps.

[0084]In one embodiment above mentioned calculations may be performed
using conventional digital signal processors (DSP), wherein a DSP may be
implemented by a general purpose processor. Alternatively application
specific integrated circuits (ASICs) may be deployed for implementing the
method. However in one embodiment the described method may be implemented
using conventional hardware, wherein the method steps are implemented in
program code executable in a receiver of an OFDM system.