I was cleaning up my workshop when I discovered I had been given
an old BWD Electronics
141A audio oscillator which had many problems.I delayed my life for 2 weeks while I rebuilt
the darn thing.
It was made in Australia in the 1960s or 70s and designed to be
used with batteries if need be.
Therefore contained the bare minimum of current needing devices
and used the bare minimum
of other parts. Current and parts cost money, and bean-counters
and CEOs hate any expenses,
and BWD had no ambitions to seriously compete with other expensive
brands like HP at the time.
The 141A unit was "budget test gear" and since it offered nothing
special, it is not disrespectful
to dismiss ideas used by the original designers and to consider
the unit to be completely
expendable in pursuit of better operation, while keeping the box,
and the few parts inside worth
actually re-cycling.
I have never ever needed battery operated test gear.

The only way to improve the performance was to revise the circuit
inside the small box completely.
After measuring and making an intelligent appraisal of the
existing usability, and having to give the unit
a 1 out of 10 score, I removed all discrete bjt + R + C and tracks
from the circuit board, except 6 pairs
of plastic and styrene C and tracks around the interactive press
button F range switches.

The 15k wire wound reverse log dual gang Compton pot for varying
was intermittent and thus faulty
and junked. It is impossible to dismantle it and clean it, or
inspect its internals. Switched resistors
were used instead of a pot for F control.
I removed the entire PSU which was not original, but a bodgy
replacement by someone.
The new PT and R&C parts of PSU are in a box screwed to the
steel plate rear panel.
I put in a new amp on an additional board for the Wien bridge
oscillator.
Another board has the j-fet as variable NFB resistance needed to
control output level.
I built a new Schmitt Trigger square wave circuit where the old
one was.
Another small board with a complementary pair of darlington
connected bjts connected as a PP emitter
follower buffer preserves the HF content of all output signals and
reduces Rout to 55 ohms for all
levels of operation. I re-designed the output attenuator switch to
give 0dB, -10dB, -20dB -30dB
levels which is more usable to me than the original. I replaced
the variable output level pot.

My new circuits contain more parts than the overly simplistic
previous original had, but it work better,
and there is now no DC offset at the output, and peak voltage for
sine waves and square waves is 5.1V,
and the rise time for square waves excellent, 104V/uS. There still
a few trimpots to adjust, and 3 trimC but
that is inevitable with much analog test gear where best
performance depends on very small changes
to R and C values. There are 3 schematics :- Fig 1 shows the switched R
& C positive FB network and the simplified amp schematic with
output attenuator
and the negative FB network with j-fet to vary the amount of NFB.Fig 2 shows the full
details of the amp using discrete j-fets and bjts which gave wider
BW than most common op-amps.Fig 3 Gives basic
properties of the Wien Bridge network. Fig 4 shows the Schmitt
Trigger square wave schematic.

Fig 1.
The Fig 1 schematic took me many hours to design and construct and
test and adjust and revise before
I got flawless operation over the six F ranges covering 1Hz to
1MHz . The original BWD 141A Comptom pot
was replaced with a 1950 dual wafer 12 position break-before-make
( BBM ) switch taken from a defunct tube
tester I was given which was quite beyond repair. I found the
amplitude bounced badly at first with this
switch so I converted it to make-before-break ( MBB ). It required
careful metal work with 2mm x 3mm brass
plates and a steady hand with soldering iron and Dremel to clean
up where needed, with two pairs of reading glasses.
This reduced the trace bounce on the oscilloscope.

If you build this, RS
components sell suitable 40mm dia wafer type switches with MBB
with 12 positions and
1 pole per wafer. I arranged the resistance combinations "R comb"
hanging off the large lugs of the old rugged switch.

Although I show rotary switches for capacitors the original press
button spring loaded interactiveswitches have
been
retained and the action is so fast that little trace bounce occurs
from high transient amp voltages at amp Vo.

The much more detailed schematic of the oscillator amp, buffered output etc and PSU is
here :-Fig 2.The original BWD 141A used an
ITT P54 thermistor which is a low wattage type of 50k cold R and
minimum of maybe100r.
It was used where the Vo was 2.5Vrms, and thus 1.66Vrms was across
the thermistor and this slightly heated it to reduce its
resistance to approximately 2,000 ohms. It worked well except that
the Vo bounced crazily. It was used where the R12 2k7
above is used between oscillator amp Vo and NFB input port.
The principle with P54 is that If Vo rises, thermistor gets
hotter, reduces its R, and increases NFB thus reducing Vo
until an equilibrium is achieved after the yo-yo of Vo settles
down. THD was at least less than 0.1%.

I didn't want to rely on getting a replacement P54 in future. No
modern thermistors seem designed to work to give a
sufficient temperature rise to cause enough useful resistance
change. All mainly seem designed to change their resistance
usefully as a result of an external change in temperature. So
modern thermistors are useless as substitutes for P54.

In the case of the Wien bridge oscillator, we want the thermistor
( or lamp globe ) to operate at a near constant T
well above room T and to vary its R due to minute changes to power
and heat generated by Vac or Vdc across the
"element" or tiny glass coated thermistor or tungsten lamp
filament within a glass tube. But lamps are very prone to Vo
bounce when switching anything, because like thermistors, some
transient high current causes a delayed rise in or fall
or resistance because temp change is not instant. Therefore, with
temperature dependent R there is an unwanted time
constant that can cause LF amplitude modulation of the Vo signal.

So, I changed the NFB circuit to use a j-fet and rectifier circuit
to produce a negative gate bias of between about -0.8V
and -1.1V.
The negative bias voltage for j-fet gate begins to be generated
when Vo = exceeds 3.2V peak. The 8 x 1N914 do
not begin conducting until their forward voltage = 0.4V each, or
3.2V across all. The turn on voltage of a silicon
diode is not linear, and the forward voltage across an Si diode
does not reach a max of about 0.7V until the max
allowable rated I flows.

The series diode action was found to be a smoother than trying to
use a low voltage zener and series 1 x 1N914.
When gate bias = 0Vdc, the j-fet has minimum Rd = 180 ohms approx.
This then makes ß a lot less than 0.333 so that
much less FB is present and oscillations start and increase Vo
very quickly. But as Vo increase, gate bias becomes
more negative and j-fet Rd increases which increases ß so
the NFB increases thus preventing an increase in Vo
when the NFB applied to Q2 gate is very nearly the same amplitude
to PFB signal at Q1 gate.
Equilibrium is reached and oscillations and "magically", you get
the same Vo amplitude for all F between 1Hz and 1MHz.
The j-fet characteristic curves do not describe their properties
fully but for each value of negative gate bias the
"diode" resistance line with high line slope near vertical id axis
varies with gate bias voltage. What isn't shown is that
the Rd operates even if the applied voltage at drain goes negative
below the source voltage.
The amount of current change in Rd across the Rd "diode
resistance" has to be kept small because the diode
R line on curves is not straight, and rolls over to a virtually
flat high value R line because above about 2Vds,
the Rd maybe thousands of ohms. If we have Vo at 3.66Vac, then
there is about 1.22Vac at both NFB and PFB
ports of the amp. In fact, the amp has open loop gain at 1kHz =
40,000 approx, and PFB voltage is only 0.09mV
above the NFB voltage. The gate to gate voltage cannot easily be
measured.

The ideal region of operation for j-fet is where Rd = 300r approx.
If I have R12 = 2,700r, and ß = 0.33,
then the whole lower R of the FB network must be 1/2 x 2,700r =
1,350 ohms.
The total R of both R = 1.35k + 2k7 = 4.05k, and if Vo = 3.66V,
then Iac = 0.903mA.
Therefore if Rd = 300r, then Vd = 0.271Vac, not much you may say,
but little things have great and fundamental
importance. Therefore the VR1 1k pot plus R6 470r must equal
1,350r - 300r = 880r so pot is set for 880r - 470r
= 410r. Because the R&C values used in the Wien network cannot
be accurate enough to demand ß always
exactly equal 0.33, and because of other slight variables in the
circuit, using a pot for NFB adjustment is
imperative.

If Vo goes higher than 3.66Vac, ß would rise above 0.333,
and oscillations would stop. To gain stable equilibrium
and thus stable Vo there
must the pot VR1 1k0 to give some fine adjustment of NFB so that
oscillations do not dither
and bounce, or have too much THD. The VR1 pot controls Vo across a
small Vo range. If pot resistance is set high,
oscillations to stop and start and "dither about", but THD is low.
If pot R is too low then you get very stable Vo but
high THD, easily exceeding 3% mainly 2H and seen in the CRO as you
twiddle the trim pot with a screw driver.Between the two extremes
there is a sweet spot where action is good, and THD < 0.5%, and
the production level of
gate bias with diodes must be
just right to get the sweet spot.

To further reduce bouncing of the Vo level when switches are used,
the Vo has a voltage clamp using 5.1 zeners and
1N4007 seen in Fig 2 below. During switching, the oscillations try
to change momentarily to higher voltage transient
bursts of other F and as square waves. The zener+diode clamp stops
Vo exceeding about 5.7V which is enough
to prevent most Vo bounce. The amp I have uses +/-20V rails and
transient Vo peak V can be up to +/- 12V.
The circuit must not generate excessive gate bias quickly which
then subsides much too slowly to allow
oscillations to continue. All operations MUST be free of delays
due to troubles with signal bounce and intermittent
oscillations, or delays between stopped and re-starting.

The major troubles you never
see mentioned or quantified in 99% of online information about WB
oscillators are
the issues of stability when changing F, and the distortion
generated by the darn little j-fet.
Now if you look closely to my above Fig 2 you will see I have a
local shunt NFB network used between drain and gate
using R8, C8, R9. At above 1Hz, the ß = 0.5, and so 1/2 of
whatever Vd exists is applied to the gate, so any THD
generated by the j-fet is half applied to the gate, thus it tends
to be amplified to reduce itself. Now the circuit works
without R8, C8, R9, and typical THD will be 2%, mainly 2H, because
the naked Rd without NFB is not a linear
resistance but one which generates up to 10% 2H right at the
drain. Part of this feeds into the NFB port and makes
the maybe 2% of 2H at Vo.

Now you may think the use of R8, C8, R9 changes the ß but
they make negligible difference to the range of values
wanted for Rd because that range is between about 200r and 1k0,
and the loading effect is negligible.
The R8 and R9 are 10k each, and at very low F where C11, C12 are
high Z, the value of R17 needs to be about10k
to prevent excessive FB at LF which makes LF stability very bad
because you have a high open loop gain amp
also involved and, where possible, shit happens that you cannot
forecast. I had R12 = 100k at first and LF stability
was terrible, but all calmed down when I made R12 = 10k. But then
you need to have sufficiently long time constants
for oscillation down to 1Hz so C11 had to be 220uF. C8, also
needed to be 220uF.
The other needed thing about the NFB loop is that it controls the
amp Vdc output offset. Ideally, Vdc at Vo should
= 0.0Vdc, and pigs will fly before you get that. BUT, I have used
fets for the amp inputs which have high Z in and
thus I get away with using the C4 2u2 + R7 1M0 network to allow Vo
DC to reach Q2g, without being mauled by the
rest of the NFB network. The loading effect of 1M in parallel with
R12 2k7 is negligible.
With R8, C8, R9, the THD drops to less than 0.03%. But other
factors affect THD. The limiting voltage clamp
diodes begin to conduct on both + and - wave crests at Vo when Vo
reaches +/- 4.9Vpk.
Such wave crest suppression causes some slight 3H++ to be
generated. But in my unit I ended up getting
THD = 0.09% at 1kHz, mainly 3H. This is much better than having 2%
2H, and is much better than many other
signal generators and function generators using chips to generate
a fabricated sine wave.

I also tried 6 x series 1N914 in 2 directions ( 12 diodes total )
to act as a voltage clamp plus VR pot across R12 2k7
but the sharp threshold action of diodes causes high 3H at maybe
5%, so that idea is hopeless.

I tried using a few 12V x 50mA lamps in series and a variable DC
current bias with pnp bjt from +20Vdc.
This made what I thought could be a good variable R for NFB
control. Two more bjts and a rectifier were
needed and it looked good in theory but amplitude stability was
appallingly bad, so I quit that idea. THD
was about the same as the j-fet with local FB though.

Now do you see how much
everybody else online in the world does not say about their
published electronic junk?
There is always more than ONE simple thing to consider about any
single R, C, or any point of connection anywhere
around any analog circuit. But at least we can get an idea,
something you cannot do with a multi-layered PCB
with zillions of multi-pin ICs, all operating digitally.The Fig 2 shows the whole
voltage amp and it looks like a good quality solid state audio
power amp. Indeed it could
be a power amp but the load to be driven is a much higher ohm
value so small signal bjts can be used throughout and with
no heat sinks.There are 4 loads on the amp, in
parallel.1. There is the NFB R
network of R10, VR1, R6, and Rd of j-fet. 2. There is the output
attenuator, minimum 772 ohms.3. There is the PFB
network of switched R&C.4. There is the series
resistance R1 5k6 Fig 4 from amp to Schmitt trigger square wave
circuit.

1. The NFB
resistance network is always 1.5 x R12 = 1.5 x 2k7 = 4,050r. Why?
Because when Vo is stable and 3.66Vac,
Q2 gate has 1/3 of Vo present, or 1.22Vrms. So there is 2.44V
across 2k7, and I = 0.904mA so load = Vo / I = 3.66 / 0.904
= 4.05k ohms.
The PFB signal at Q1 gate may
be 1.221Vrms and is only
just above Q2 gate. The gate to gate voltage is extremely low
where open loop amp gain may be say 40,000 which would make Vg-g =
0.09mV, and quite difficult to measure.

2. The output
attenuator load is VR2, 772 ohms and it came from my junk box, and
seemed to have no serious
faults so common in old pots when dust, dirt, pollution,
corrosion, and loose terminals make many useless.
I always clean out the pot with white spirits and plenty of
turning actions then blow out with compressed air
and then compress rivets holding lugs to tracks using long nose
pliers in a vice. I seal up the hole near the lugs
with cardboard and tape and THEN such pots give many more years of
trouble free operation.
Most cheap log pots you buy won't be exactly 1k0, and somewhere
between 700r and 1k2, and mine is 772 ohms.
If I ever change to another "1k0" replacement LOG pot, a parallel
trimming R will be needed to ensure the total
nominal R value = 772 ohms. If the R is not 772r, the switch S2
attenuator with its set resistances cannot give
correct -10dB reductions of output signal amplitude because the
attenuator R values are calculated for only 772r.
-10dB is a nicely convenient reduction of output level. It means
Vo is reduced by factor 1 / square root 10,
or x 0.3162. This is near 1/3. Two such reductions of x
0.3162 give a reduction of x 0.1, and you will eventually
find this engineers' way of reducing outputs and inputs is a
pleasure to use.

3. The positive
FB circuit is the Wien bridge RC network. For where the WB network
has equal two value C and R,
the input load impedance becomes 2.121 x resistance value, and
because R is varied between R for highest F
and 10xR for lowest F, the load we need to consider is the lowest
because it needs the highest current and
in this case R = 1,590r, and load = 2.121 x 1,590 = becomes
3,372r. So where did the constant 2.121 come from?

The factor 2.121 is the factor calculated from 1.414 x 1.5. Huh?
Why?
In all Wien RC bridges, the output signal response is that of a
low Q tuned circuit and Vo can only ever rise to 0.333 x Vin
which occurs at only one frequency, that of "resonance" Fo, where
phase shift between Vin and Vo = zero degrees.
See Fig 3 below.
Fig 3.
From the above basic Wien bridge when operating at Fo we see Vin =
10Vrms and V0 = 3.3V.
So that 6.67V exists across R1 in series with C1. And we need to
know that phase shift between
Vin and Vo = 0 degrees, lest all our statements about measured
voltages are wrong.

At 1kHz Fo, reactance XC = R ohms, and impedance, Z for R and C in
series
= 1.414 x R = 2,248 ohms. Therefore current flow = 6.67V / 2,484r
= 2.966mA.
Therefore impedance looking into the input = 10V / 2.966mA = 3,370
ohms.
The result validity depends on the the 0 degrees of phase shift
between Vi and Vo.
The same 2.966mA flows through the Z of R2 and C2 in parallel, and
may be calculated as 0.707 x R
= 1,124r, or we may calculate from observations, Z = 3.3V /
2.966mA = 1,123 ohms.
So total Z between Input and 0V = 2,248 + 1,123 = 3,372 ohms.
The simplest way to calculate Z in for Wien bridge network at Fo
is Z in = 2.121 x R.
For decreasing frequencies below Fo, the Z in rises towards
infinity at 0.0Hz
because the C become open circuits.
For F increasing above Fo, Z in reduces to a minimum of R,
because the C become short circuits at HF.

4.There is a
feed resistance from oscillator amp and square wave circuit, Fig
4, R1, 5k6.
The voltage at the Q9 base end of R1 is less than 1/5 of
Oscillator output so in fact effective
loading is about 6k2 from this resistance of 5k6. ( See how there
is always more than ONE thing
to consider
about any single part in an electronic circuit! ).

The amp must be able to power the load with little THD caused by
loading.
I like my signal generators to make +/-5V peak at least, and in
this case 3.66Vrms.
So load current at 3.66V = 3.66 / 501 = 7.3mA rms. It could easily
be possible to have idle DC current
in Q6 and Q7 to give pure class A operation if Ic =11Adc.
But this is not necessary, and the Q6&7 may operate
in class AB at low enough THD because the NFB acts to reduce
distortion in the amp which has very wide
open loop bandwidth at 3.66Vac thus allowing 1MHz without any
attenuation. But anyone is free to have
Ic = say 20 mA, and perhaps gain a slight amount of better HF
stability. I found that capacitive loads on the
amp may cause oscillations cease but very little HF gain reduction
was needed with Zobel R&C between
Q4 base and collector, see R9 and VC1, Fig2.

The Vo from the sine wave
oscillator in Fig 2 is prone to poor HF performance especially if
there is even a small amount
of C, say 100pF between the output side of R16, 33r, and 0V. This
is enough to destabilize HF oscillations above 300kHz.
But the shown switched attenuator S2 and pot VR2 cause negligible
C shunting themselves. To protect the oscillator amp
from effects of C shunting of signal due to cables or C in of gear
and from low Rin of gear connected, I decided a
buffer output stage was needed.

To make it rugged, simple, and wide band, I used PN100+BD139 and
PN200+BD140
connected as a complementary emitter follower pair which gives
gain close to 1.0. Its Rout is set by 22r emitter resistors
and the 47r output resistance. A zener diode + diode 10V clamp
prevents high V from something external to the unit
providing a high voltage signal back into the unit output which
would instantly fuse the bjts. The 47r is 1/8 W rated,
and fuses open if high voltage enters the unit. Diode clamps are
also placed from 22r junction to rails for further
limiting of stray unwanted back flows of current, like when you
accidentally touch an RCA lead on an anode at +400V.

The buffer works in class A with all loads with Z above about
400r. Now that allows 400pF as a load at 1MHz.
Class AB action occurs with lower loads. A shorted output with
maximum Vo of 3.6Vac being generated
causes a peak current of 74mA, and heats the 47r, but won't blow
it, and causes about 0.5W of heat in
each of BD139, BD140 which won't kill them. Notice the R36,
R37, both 680r emitter resistors for drivers
Q9 and Q10. These R make the idle current in Q9 and Q10 larger
than if it was only the base current
of Q11 and Q12, and give a path for discharging turn off current from bjts
at HF, something bjts don't do so
well.

With the attenuator S2 and VR2 placed before the buffer, there is
no attenuation of HF content from the
sine wave oscillator and the HF content of the square wave appears
well preserved. I found the square wave
at 1MHz to look well compared to many other units I have used, and
its highest F appears to be 10mhz, and
the rise time was calculated to be 104V/uS.

However, most good audio amps are not designed to operate above
65kHz, -3dB pole for the amp, and the
response may be measured with test sine waves of 1Hz to 200kHz.
However, to examine reactive behaviors
and transformer resonant behaviour, it is often useful to have
sine wave signals up to 1MHz, or beyond,
especially where mosfets are used in an SS amp which may possibly
produce 5MHz, intentionally, or not
intended, when they oscillate, or are provoked into oscillations
with a fast enough square wave.

The Schmitt Trigger square wave circuit in the original BWD 141
had 2 x BC557 pnp transistors with feeble Idc flow.
The good "squareness" of square waves disappeared above 100kHz,
and at LF.
The original 141 had a very low square wave output level and
squareness above 100kHz was poor, ie, there
was much HF content removed by the attenuator network and
following cable capacitance.

I searched the Internet, and found
many examples for Schmitt Trigger sine to square wave converters.
None of the pages had any mention about the troubles ppl face with
distortions and odd behaviors
before they get a good square wave. I tried a few online circuits
with npn bjts, and then with j-fets,
and mosfets, but the npn bjts were just fine using BC546 or PN100
or 2N2222. During my trials of the devices
I played around with variations on R&C values until I
eventually found further experimentation pointless,
unless I was to try something one never sees, such as a pair of
Schmitt Trigger circuits with two pairs of N bjts
and two pairs of P bjts wired in parallel with cap coupling. But I
did get good enough performance with the
circuit below......
Fig 4.
The input to the Q13 base network via the above R1 5k6 is
permanently connected to the sine wave
amp output in Fig 2, ie, output side of R17, 33r. The square wave
converter works continuously
while ever the whole unit is turned on, so that when switching
from sine wave to square wave
there is minimum delay while Vdc rails and working voltages
establish themselves because
long time constants are involved.

When switching S1 Fig 2 to sine wave, there is no sign of any
leakage of square wave gate crashing the
sine wave party.

As sine wave F rises above 500kHz, there are some difficulty
getting enough energy to trigger the base input
at Q13 and Q14 to maintain square wave symmetry.
To ensure enough triggering voltage at HF appears at Q9 base to
give symmetry, R1a 150r and C1a 100pF
have been added as a Zobel network across R1 5k6.
This was found to have a negligible extra loading on the sine wave
amp at HF.

The Q15 and Q16 complementary pair emitter follower buffer
overcomes the problem of attenuation of HF
on positive going V rise at R9, 1k1.
The 1k1 is not a low enough resistance to overcome any stray C in
following attenuator network, and the
attenuator network has low minimum R in of 770r, so the buffer IS
NEEDED.

The Fig 4 schematic has 470uF bypassing R6 15k. In most other
schematics online, R6 is not bypassed,
or bypassed with a small value of C, perhaps 100pF. But after
trialing several values, I concluded 470uF was fine.
With no bypass C or low values of C, the delivery of positive FB
from Q9 collector to Q10 base is rather too slow.
And I found the square wave's lower horizontal had a large kink,
so the wave looked wrong, and unacceptable.
The RC coupling as I have it works best.

VR3 and R7 must be adjusted carefully to get the wanted level of
+/- 5.17Vpk AND good wave symmetry
where both top and bottom horizontals of the square wave have
equal length, and time between verticals
is equal. The range of VR3 pot movement needed to get square waves
to occur at all is rather small.
The R7 value was adjusted with a temporary pot to get the wanted
peak Vo while VR3 was adjusted
for best symmetry as R7 is adjusted.

Nobody else tells you that you MUST adjust these two resistors
together because they are somewhat
interactive, and beginners will be frustrated when they build
something they saw online and it fails to
work and they stare at it for 10 years without knowing what
questions to ask or what to try to get it
working. When using 10kHz sine wave input you should get a
splendid
looking square wave and you should find all other F give the same
peak Vo and good symmetry.

I have no provision for applying DC offsets at the output, so that
in effect a positive or negative bias
voltage cannot be included in the total Vo signal. Nor can you
vary the asymmetry for a weird looking
square waves.
I have never needed Vdc offsets, and have always found that having
Vdc content in signal gene
outputs is a royal pain in the arse when testing something where
you WANT ONLY 0Vdc at the
input to something you are testing, and if you do want some DC,
then make up a suitable C&R coupling
circuit and add the Vdc bias manually. If there is to be any
biasing of amps under test, it will be
done within the amp, which should not have any Vdc present at its
input, say from a grid which
is drawing some slight grid current at idle, and every time you
switch sources you get a loud click.

I do not propose to add more boards for a "triangle wave" or saw
tooth wave, or for AM or FM modulation.
To test audio amps, there is no need for more than what this unit
now does, ie, give sine and
square waves up to 5.15Vpk level, 1Hz to 1MHz, and with no Vdc
content.

For the extra functions an additional box could be made with a
suitable circuit you will have to research
and develop after perhaps beginning with something you'll find
online but which probably will not be fully
explained properly, or which just won't work very well. A good
triangle wave with straight lines can indicate
very basic linearity of an amp because you can see when the
straight lines get a bend if you have a
dual trace CRO.
F distortion can occur, ie, "integration", ie, further action by
C&R or L&R circuits within the tested amp.
For measuring THD, it is better to use a dedicated low THD sine
wave and following notch filter as described
elsewhere at this website, thd-measurement.html

Because of long time
constants used in the unit, Vdc settlement after turn on takes
about 20 seconds
before 0Vdc appears at the output with negligible Vdc measured.
Some variation of the Vdc offset
will occur with use and switching F ranges or from sine to square.
Vdc will eventually settle. Most good amps have a C&R high
pass filter at their input such as say 0.33uF
plus 68k, giving LF pole at 7.1Hz. The time constant = 0.022
seconds. This will cause serious distortion
of LF square waves. But you will find the distortion where square
wave horizontals become highly sloped
at below 30Hz is of little concern. With this oscillator unit you
can place a bypass link across the 0.33 uF and then the
time constant is set by C24 200uF and 68k plus R41 47k at unit Vo
thus giving time constant of 5.5 seconds,
and LF pole at input at 0.028Hz. The distortion of LF square waves
is then negligible.
At the unit's Vo, the R41 47k, gives a long time constant = 9.4
seconds when unloaded. So where
you have a tube amp with 0.22uF and 470k at input, the TC = 0.103
seconds, LF pole = 1.53Hz and you
will still get square wave distortion at the input. If the 0.22uF
was bypassed with say 22uF NP cap, then
you get a TC = 10.3 seconds and square wave at input will
keep its integrity.
However, tube amps have maybe 3 internal C&R couplings and at
least one R&L OPT coupling
which all cause trouble with square wave inputs. The NFB used will
correct much of the errors
in square waves that are inevitable. But do the so called errors
with square waves at LF matter?
Fortunately, nobody has proved the C&R and R&L coupling
causes bad LF sound, and in fact
well made tube amps just cope, and sound is excellent.

What is important is the HF square wave performance of any amp and
its stability with C loading
or without any load at all.
The amp under test should be able to give a good looking
reproduction of square waves between 100Hz and 5kHz.
There should be very little ringing or overshoot with no amp load
or with a pure R load of nominal value +/- 100%.
So something designed for 8 ohms should work OK with 4 ohms or 16
ohms even though less PO is available.
Some overshoot may be allowed to occur with solely C loads between
0.05uF and 10uF, and where Vo is kept low
to avoid overload.

Many amps fail when tested with square waves and they can
oscillate badly at some RF when no load is connected
or when a pure C load is connected. Most dynamic speakers are
inductive at HF and many do not have a Zobel network
at their X-over filter to ensure there is a nominal R load when F
rises above 20kHz, such as say 8r2 + 0.47u.

The beauty of square wave testing is that you can adjust the the C
value used to bypass the FB resistance from
amp output to its FB input port, and adjust the HF gain and phase
change of the voltage amp to ensure HF gain at
F above 20kHz is reduced, along with phase shift so that FB
remains negative, and not positive, even when C loads
are used. A CRO and a good signal gene are essential tools for
getting any amplifier to work well as it can,
and give optimal bandwidth and complete unconditional stability
when NFB is used. In amps without NFB,
the signal gene is still essential to make sure sine wave response
is as wide as wanted. Without NFB, most amps
have no stability or oscillation problems, but the CRO will tell
you their transient response is much poorer than
their brothers with well designed NFB networks.