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Abstract:

The present invention relates to a supply circuit (1) comprising: a
bridge circuit (3) comprising at least two series-connected switches
(M1, M2) being coupleable to a load circuit (11), a resonant
circuit (5) coupleable at one end to a power source (7) and coupled at
another end to the interconnection (15) of the at least two switches
(M1, M2) of the bridge circuit (3), and at least two diodes
(D1, D2), wherein a diode is coupled between each switch
(M1, M2) and the power source (7).

Claims:

1. Supply circuit comprising: a bridge circuit comprising at least two
series-connected switches being coupleable to a load circuit, a resonant
circuit coupleable at one end to a power source and coupled at another
end to the interconnection of the at least two switches of the bridge
circuit, and at least two diodes, wherein a diode is coupled between each
switch and the power source.

2. Supply circuit as defined in claim 1, wherein one diode is polarized
with respect to the assigned switch so that a flow of negative resonant
current is allowed and another diode is polarized with respect to the
assigned switch so that a flow of positive resonant current is allowed.

3. Supply circuit as defined in claim 1, wherein the resonant circuit is
a series resonant circuit comprising an inductance and a capacitance.

4. Supply circuit as defined in claim 1, wherein the resonant circuit is
coupled to the interconnections between the diodes and the power source.

5. Supply circuit as defined in claim 1, wherein the capacitance is
divided into two partial capacitances, each partial capacitance
comprising half the resonant capacitance, each partial capacitance being
coupled to the interconnections of the diodes and the power source.

6. Supply circuit as defined in claim 1, wherein the at least two
switches of the resonant circuit are MOSFETs.

7. Supply circuit as defined in claim 1, further comprising a control
unit, which is adapted for providing a maximum switching frequency of the
bridge circuit, which is in the range from 10% to 50% of the resonant
frequency of the resonant circuit.

8. Supply circuit as defined in claim 1, further comprising a control
unit, which is adapted for providing a maximum switching frequency of the
bridge circuit, which is in the range of half the resonant frequency of
the resonant circuit.

9. Supply circuit as defined in claim 7, wherein the control unit is
adapted for providing a switching of the switches of the bridge circuit
with a duty-cycle of up to 50%.

10. Device comprising a power supply, a load circuit, and a supply
circuit as defined in claim 1 for supplying said load circuit.

11. Device as claimed in claim 10, further comprising an output filter
between said supply circuit and said load circuit.

Description:

FIELD OF THE INVENTION

[0001] The present invention relates to a supply circuit, and also relates
to a device comprising a supply circuit.

BACKGROUND OF THE INVENTION

[0002] Supply circuits, in particular switched mode power supplies are
well known in the art. Such supply circuits are for example integrated in
consumer and non-consumer products. An exemplary application is the
powering of light-emitting diodes (LEDs) and/or organic light-emitting
diodes (OLEDs), in particular LED/OLED strings used for automotive
LED/OLED lighting and in general, battery powered LED/OLED lighting
systems.

[0003] Supply circuits that are best suited and therefore are preferably
used for the above-named applications are in particular Discontinuous
Series Resonant Converters with a constant average current output I, in
the following denoted as DSRC-I. This type of converter is for example
described in WO2008/110978. The functionality of this type of converter
is well understood by those skilled in the art and is therefore not
explained in more detail. DSRC-I converters provide the advantage of a
constant average current output, furthermore, no current sensing and no
current control loop is required. Consequently, losses caused by a
current sensing are avoided, and the DSRC-I provides a high-efficient,
compact and easy design compared to other commonly known series resonant
converters.

[0004] A disadvantage of the basic DSRC-I converter is that the output
voltage has to be lower than the input voltage if no transformer or
additional components such as an additional voltage doubler circuit are
provided. However, both solutions need space and increase the costs of
the circuitry. As an example, an LED backlight of a car, which consists
of several LEDs in a series connection will need more than 12 V of the
car battery, e.g., 5 LEDs in series require 5×3.3 V=16.5 V. Hence,
the DSRC-I causes problems if several LEDs have to be connected in series
and only a low supply voltage is available, e.g., in automotive
applications.

[0005] Battery powered systems often also stack cells in series to achieve
a higher output voltage. However, sufficient stacking of cells is not
possible in many high voltage applications due to a lack of space.

SUMMARY OF THE INVENTION

[0006] It is an object of the present invention to provide a supply
circuit by which an output voltage can be obtained that is higher than
the input voltage. The supply circuit comprising a boost function
according to the present invention can step up the input voltage, i.e.,
increase the output voltage and so reduce the number of battery cells.

[0007] According to an aspect of the present invention a supply circuit is
provided comprising

[0008] a bridge circuit comprising at least two series-connected switches
being coupleable to a load circuit,

[0009] a resonant circuit coupleable at one end to a power source and
coupled at another end to the interconnection of the at least two
switches of the bridge circuit, and

[0010] at least two diodes, wherein a diode is coupled between each switch
and the power source.

[0011] This converter topology provides a constant average current output
at a higher output voltage than the input voltage. Further, it has a
simple circuit design and does not require a transformer or another
additional component. Altogether, the converter provides the advantage
that no current sensing and current control is required, furthermore, a
very compact circuit design is provided with an integrated voltage boost.
The supply circuit according to the present invention is above all easy
to design, simple to control and provides a high efficiency. The detailed
functionality of the supply circuit will be explained in the context of
the Figures.

[0012] In a first aspect of the present invention a supply circuit is
presented, wherein one diode is polarized with respect to the assigned
switch so that a flow of negative (polarized) resonant current is allowed
and another diode is polarized with respect to the assigned switch so
that a flow of positive (polarized) resonant current is allowed. This
provides the advantage that only a positive current flows through the
output.

[0013] In a further aspect of the present invention a supply circuit is
presented, wherein the resonant circuit is a series resonant circuit
comprising an inductance and a capacitance. This is advantageous as it
assures the advantageous functionality of the DSRC-I as well as zero
current switching (ZCS), which is well known in the art and is therefore
not further explained.

[0014] In yet another aspect of the present invention a supply circuit is
presented, wherein the resonant circuit is coupled to the
interconnections between the diodes and the power source. In particular,
the capacitance is divided into at least two partial capacitances, each
partial capacitance comprising half the resonant capacitance, each
partial capacitance being coupled to the interconnections of the diodes
and the power source. This topology is advantageous as a boost function
is realized and in addition, the major advantageous of the common DSRC-I
are upheld.

[0015] In a further aspect of the present invention a supply circuit is
presented, wherein the at least two switches of the resonant circuit are
MOSFETs. This is advantageous because the MOSFETs are suitable for the
above-named applications and are in addition easy to control.

[0016] In a further aspect of the present invention a supply circuit is
presented, further comprising a control unit, which is adapted for
providing a maximum switching frequency of the bridge circuit, which is
in the range from 10% to 50% of the resonant frequency of the resonant
circuit, in particular in the range of half the resonant frequency of the
resonant circuit.

[0017] Furthermore, the control unit is adapted for providing a switching
of the switches of the bridge circuit with a duty-cycle of up to 50%.
Practically, a duty-cycle of exactly 50% cannot be achieved, but a short
dead-time has preferably to be implemented between the high-side and the
low-side switch, which is preferably in the range from 100 ns to 1 μs.

[0018] According to another aspect of the present invention a device is
provided comprising a power supply, a load circuit, and a supply circuit
as proposed according to the present invention for supplying said load
circuit. It shall be understood that the device comprises the same
advantageous as the supply circuit itself. The device may comprise one or
more loads, whereas the load comprises one or more LEDs, OLEDs or the
like, and the device could, for instance, be a lighting unit.

[0019] Preferably, said output filter is arranged between said supply
circuit and said load circuit. The output filter stabilizes the output
voltage and, hence, guarantees a lower DC ripple of the load current. The
output filter can be implemented simply by a capacitor coupled in
parallel to the load circuit, but more complicated filters are possible,
e.g. comprising series and/or parallel circuits comprising one or more
capacitors and/or inductances, as are generally known in the art.

[0020] It shall be understood that the claimed device has similar and/or
identical preferred embodiments as the claimed supply circuit as defined
in the dependent claims.

BRIEF DESCRIPTION OF THE DRAWINGS

[0021] These and other aspects of the invention will be apparent from and
elucidated with reference to the embodiment(s) described hereinafter. In
the following drawings

[0022]FIG. 1 shows a block diagram of a supply circuit in accordance with
an embodiment of the present invention;

[0023] FIG. 2 shows a simulation schematic of the supply circuit in
accordance with an embodiment of the present invention;

[0024]FIG. 3 shows simulation results for a first set of parameter
values;

[0025]FIG. 4 shows simulation results for a second set of parameter
values;

[0026]FIG. 5 shows simulation results for a third set of parameter
values;

[0027]FIG. 6 shows simulation results for a fourth set of parameter
values;

[0028] FIG. 7 shows a simplified block diagram of a supply circuit in
accordance with an embodiment of the present invention;

[0029]FIG. 8 shows a further simplified block diagram of a supply circuit
in accordance with an embodiment of the present invention;

[0030] FIG. 9 shows a block diagram of the conducting parts of a supply
circuit in accordance with an embodiment of the present invention for a
first time interval;

[0031] FIG. 10 shows a block diagram of the conducting parts of a supply
circuit in accordance with an embodiment of the present invention for a
second time interval;

[0032] FIG. 11 shows a block diagram of the conducting parts of a supply
circuit in accordance with an embodiment of the present invention for a
third time interval;

[0033]FIG. 12 shows a block diagram of the conducting parts of a supply
circuit in accordance with an embodiment of the present invention for a
fourth time interval;

[0035]FIG. 1 shows a block diagram of a supply circuit 1 in accordance
with an embodiment of the present invention. The supply circuit 1
comprises a bridge circuit 3, a resonant circuit 5, which is coupleable
at one end to a power source 7, wherein the power source 7 is preferably
a direct voltage source The supply circuit 1 is coupled to a load circuit
9, which comprises at least one, in FIG. 1 exemplary a total of four,
loads 11 and a smoothing capacitor 13 being connected in parallel to the
loads 11. A load 11 may be a LED, an OLED or the like. The output voltage
Vout is dropped across the array of loads 11.

[0036] The bridge circuit 3 comprises at least two switches M1 and
M2 that are exemplary MOSFETs, which are controlled by a control
unit 14. In response to a direct current from the power source 7, the
bridge circuit 3 communicates a voltage signal to the resonant circuit 5
at a switching frequency fswitch, which in turn communicates an
alternating current Ir to the load circuit 9.

[0037] The switches M1 and M2 of the bridge circuit 3 are
preferably switched by means of the control unit 14, which is adapted to
provide a switching duty-cycle of 50%. Furthermore, the control unit 14
is adapted to provide a maximum switching frequency fswitch of the
bridge circuit 3, which is preferably half the resonant frequency
fres of the resonant circuit 5.

[0038] The switches M1 and M2 are connected in series, whereas
the source contact of switch M1 is coupled to the drain contact of
switch M2 by an interconnection 15.

[0039] The resonant circuit 5 is coupleable at one end to the power source
7 and coupled at another end to the interconnection 15 of the at least
two switches M1 and M2 of the bridge circuit 3. The resonant
circuit 5 comprises an inductance Lres and a capacitance Cres,
whereas the capacitance Cres is exemplary divided into two partial
resonant capacitances Cres/2. Hence, each of the partial
capacitances Cres/2 comprises half the resonant capacitance
Cres.

[0040]FIG. 1 further illustrates that a diode D1 is assigned to the
switch M1 and a diode D2 is assigned to the switch M2. In
particular, the diodes D1 and D2 are interconnected between
each switch M1 and M2 and the power source 7 and are in
particular connected in series with the respective assigned switch
M1 or M2 on the one hand and to the power source 7 on the other
hand. One of the diodes, in particular diode D1 is polarized with
respect to the assigned switch M1 so that a flow of negative
(polarized) resonant current Ir through diode D1 is allowed and
another diode, in particular D2 is polarized with respect to the
assigned switch M2 so that a flow of positive (polarized) resonant
current Ir through diode D2 is allowed.

[0041] As will be explained in more detail later on, the voltage drop
V1 across the resonant circuit 5 depends on the diodes and depends
in particular on which diode is at present conductive. Hence, the voltage
drop across the resonant circuit 5 may be summarized as follows: M1 on,
D1 is conductive: -Vin/2; M1 connected to D2, D2 is conductive:
Vin/2-Vout; M2 connected to D2, D2 is conductive: Vin/2; M2 on, D1 is
conductive: -Vin/2+Vout.

[0042] The partial capacitances Cres/2 are connected in series with
the inductance Lres and further are coupled to the interconnections
between a diode D1 or D2 and the power source 7. Thus, one
partial capacitance Cres/2 is coupled to the interconnection 17
between diode D1 and the power source 7, and the other partial
capacitance Cres/2 is coupled to the interconnection 19 between
diode D2 and the power source 7.

[0043] The above described novel topology of the supply circuit 1 realizes
a DSRC-I comprising most of its major advantages and in addition provides
a boost function, so that the output voltage Vout is higher than the
input voltage Vin without the need for any additional components
such as a transformer.

[0044] It shall be noticed that a device 21 according to the present
invention comprises that supply circuit 1 and may in addition comprise
one or more load circuits 9.

[0045] FIG. 2 shows a simulation schematic of the supply circuit 1 in
accordance with an embodiment of the present invention, whereas FIG. 3 to
FIG. 6 show simulation results for different sets of parameter values.
The simulation schematic of FIG. 2 is based on the supply circuit
topology illustrated in FIG. 1

[0046]FIG. 3 shows simulation results for a second set of parameter
values. In particular, the simulation results are based on an input
voltage Vin=24 V, an output voltage of Vout=30 V and a
switching frequency of the bridge circuit fswitch=fres/2, i.e.,
the switching frequency is half the resonant frequency fres.

[0047] The topmost simulation schematic of FIG. 3 illustrates the currents
I(V1) and I(V4) as a function of time t. Thereby, the voltage
V1 corresponds to the voltage Vin illustrated in FIG. 1 and the
voltage V2 corresponds to the voltage Vout illustrated in FIG.
1. It is obvious that the output current I(V4) is lower than the
input current I(V1).

[0048] The middle simulation schematic of FIG. 3 illustrates the diode
currents I(D1) and I(D4) as a function of time t. As explained
above, that diodes D1 and D2 are connected to their assigned
switches M1 and M2 with opposite polarizations. Therefore, the
diodes D1 and D2 allow current flow alternately depending on
the polarization of the resonant current Ir as will be explained in
more detail hereinafter.

[0049] The lower simulation schematic of FIG. 3 illustrates the resonant
current I(Lres) as a function of time t. The resonant current
I(Lres) corresponds to the resonant current Ir of FIG. 1.

[0050]FIG. 4 shows simulation results for a second set of parameter
values. In particular, the simulation results are based on an input
voltage Vin=24 V, an output voltage of Vout=40 V and a
switching frequency of the bridge circuit fswitch=fres/2, i.e.,
the switching frequency is half the resonant frequency fres.

[0051]FIG. 5 shows simulation results for a third set of parameter
values. In particular, the simulation results are based on an input
voltage Vin=24 V, an output voltage of Vout=50 V and a
switching frequency of the bridge circuit fswitch=fres/2, i.e.,
the switching frequency is half the resonant frequency fres.

[0052]FIG. 6 shows simulation results for a fourth set of parameter
values. In particular, the simulation results are based on an input
voltage Vin=24 V, an output voltage of Vout=40 V and a
switching frequency of the bridge circuit fswitch=fres/3, i.e.,
the switching frequency is one third of the resonant frequency fres.

[0053] In order to describe the functionality of the supply circuit 1, the
topology shown in FIG. 1 can be simplified as illustrated in FIG. 7 and
FIG. 8. In FIG. 7, there are two capacitances Cin1 and Cin2
provided and additionally a resonant capacitance Cres. In FIG. 8 the
partial resonant capacitances Cres/2 of FIG. 1 are combined to one
single capacitance Cres and the power source 7 is virtually split
into two partial power sources 7' and 7'', each providing a direct
voltage Vin/2. It should be noted that taking two partial
capacitances Cres or two capacitances Cin1 and Cin2 and in
addition a resonant capacitance Cres leads to the same result. It
can be seen from FIG. 8 that a voltage dropped across the capacitance
Cres is denoted as VC and a voltage dropped across the
inductance Lres is denoted as VL.

[0054] The resonant circuit 5 can be described with its resonant frequency
fres and its resonant impedance Zres.

Based on simulation results, the circuit behavior can be explained as
follows: For the description in time intervals the half resonant period
τ is defined.

τ = 1 2 T res = 1 2 1 f res ( 3 )
##EQU00002##

[0055] The switching period of the switches M1 and M2 is
Tswitch as can be seen from FIG. 7 and
2*Tres≦Tswitch. The conducting parts in each time
interval are depicted in FIG. 10 to FIG. 13.

[0056] FIG. 9 shows a block diagram of the conducting parts of a supply
circuit 1 in accordance with an embodiment of the present invention for a
first time interval t1: 0<t≦τ, which is illustrated in
FIG. 13. During this time interval, switch M1 is switched on and
switch M2 is switched off. The resonant circuit 5 generates in this
time interval a first, negative sinusoidal half-wave exemplary denoted in
FIG. 13 with W1.

[0057] Hence, switch M1 allows a current flow, which is communicated
from the direct voltage source 7'. The voltage dropped across the series
resonant circuit 3, i.e., across capacitance Cres and inductance
Lres is denoted in FIG. 9 with V1.

[0058] As the resulting current Ir is negative, the diode D1
will be conductive for this current. Diode D2 is polarized opposed
to diode D1 and will therefore not allow a flow of the negative
current Ir in the first time interval.

[0059] Based on simulation results, the conducting components in each time
interval are known and the amplitude of each sinusoidal half wave can be
calculated. From the idealized circuit the voltage dropped across the
resonant capacitor, denoted in FIG. 8 as VC(t), at the beginning of
the first time interval can be calculated. The result is:

VC(t=0)=Vout-Vt (4)

Additionally, the voltage drop V1 across Cres and Lres can
be obtained from FIG. 9: With help of the initial condition and the
voltage of the resonant capacitor VC, the amplitude of each
sinusoidal half-wave and the capacitor's voltage VC after finishing
each cycle can be calculated. For every cycle the voltage V1,
applied to the whole resonant circuit can be obtained from the conducting
parts. For the first cycle V1 is:

V 1 ( 0 < T ≦ τ ) = - V i n 2
( 5 ) ##EQU00003##

Based on idealized circuit behavior, the amplitude of the resulting
first, negative sinusoidal half-wave W1 can be calculated.

I ^ 1 = - V out + V i n 2 Z res ( 6 )
##EQU00004##

Further current flow through D1 after this half-wave is prevented by
diode D1, as current Ir becomes positive.

[0060] FIG. 10 shows a block diagram of the conducting parts of a supply
circuit 1 in accordance with an embodiment of the present invention for a
second time interval t2: τ<t≦Tswitch/2. During
this time interval, switch M1 is still switched on and switch
M2 is still switched off. The resonant circuit 5 generates in this
time interval a second, positive sinusoidal half-wave exemplary denoted
in FIG. 13 with W2.

[0061] Hence, the current Ir is thus positive during this time
interval t2. Consequently, diode D1 does not allow current flow
and thus blocks the positive current Ir. However, diode D2,
which is polarized opposed to diode D2 allows current flow of the
positive current Ir. It is obvious from FIG. 10 that the current
Iout flows through the output.

[0062] From calculations with formulas of the first time interval t1,
the resonant capacitors voltage VC(t) is:

VC(t=τ)=-Vout (7)

And V1:

[0063] V 1 ( τ < t ≦ 2 τ ) = V i
n 2 - V out ( 8 ) ##EQU00005##

This leads to the amplitude of the second, positive sinusoidal half-wave
W2:

I ^ 2 = V in 2 Z res ( 9 ) ##EQU00006##

Further current flow is prevented by diode D2.

[0064] FIG. 11 shows a block diagram of the conducting parts of the supply
circuit 1 in accordance with an embodiment of the present invention for a
third time interval t3:
Tswitch/2<t≦Tswitch/2+τ. During this time
interval, switch M1 is switched off and switch M2 is switched
on. The resonant circuit 5 generates in this time interval a third,
positive sinusoidal half-wave exemplary denoted in FIG. 13 with W3.

[0065] Hence, the current Ir is thus positive during this time interval
t3. Consequently, diode D1 does not allow current flow and thus
blocks the positive current Ir. However, diode D2, which is
polarized opposed to diode D2 allows current flow of the positive
current Ir.

[0066] The behavior in the third and fourth time interval t3 and
t4 is similar to that of the first one and second time interval
t1 and t2. Basically, the current half-waves occur with the
opposite sign.

[0067] The capacitor voltage VC(t) at the beginning of the third time
period t3 is:

Consequently, the third, positive sinusoidal half-wave W3 has the
following amplitude:

I ^ 3 = V out - V in 2 Z res ( 12 )
##EQU00008##

[0069]FIG. 12 shows a block diagram of the conducting parts of a supply
circuit in accordance with an embodiment of the present invention for a
fourth time interval t4:
Tswitch/2+τ<t≦Tswitch. During this time interval,
switch M1 is still switched off and switch M2 is still switched
on. The resonant circuit 5 generates in this time interval a fourth,
negative sinusoidal half-wave exemplary denoted in FIG. 13 with W4.

[0070] Hence, the current Ir is thus negative during this time
interval t4. Consequently, diode D1 does allow current flow of the
negative current Ir. However, diode D2, which is polarized
opposed to diode D2 does not allow current flow of the negative
current Ir. It is obvious from FIG. 12 that the current Ires
again flows through the output.

[0071] Finally, the capacitor voltage VC(t) at the beginning of the
fourth time period t4 is:

[0073] This leads to the amplitude of the fourth, negative, sinusoidal
half-wave W4:

I ^ 4 = - V in 2 Z res ( 15 ) ##EQU00010##

[0074] The circuit behavior shows, that only two sinusoidal half-waves,
namely W2 and W4 flow through the output. Consequently, the
output current Iout consists of two sinusoidal half-waves W2
and W4 per switching period Tswitch.

[0075] The functionality of the supply circuit 1 according to the
invention and the resulting "boost" function will now be explained in
more detail: The topology of the supply circuit 1 causes two of four
half-waves (in particular every second of four half-waves) of the
resonant current Ires not to flow through the output, i.e., the
load. With respect to FIGS. 9 to 13, the first and the third half-waves
W1 and W3 do not flow through the output as can be seen from
FIGS. 9 and 11.

[0076] This respective half-wave is e.g., I1=(-Vout+Vin/2)
Zres, when referring to the first have wave W1 of FIG. 13.
Taking into account the initial condition for the voltage drop across the
capacitance Cres and the voltage drop V1 across the resonant
circuit 5, the amount of the voltage drop across the capacitance
Cres after the first half-wave W1 is equal to the output
voltage Vout. Hence, for the subsequent half-wave W2, the
available voltage is resulting from the series connection of the voltage
drop corresponding to Vout across the capacitance Cres and half
of the input voltage Vin/2.

[0077] However, the output voltage Vout always acts against the
second half-wave W2, and thus, the half of the input voltage, namely
Vin/2 is always left over, enforcing a current flow through the
load. Consequently, the second and fourth half-waves W2 and W4
flow through the load and the amplitude of the current is independent
from the load voltage in case that the load voltage is larger than the
input voltage Vin.

[0078] Consequently, the present invention provides a supply circuit 1, in
particular a converter topology that can be used for automotive LED/OLED
lighting or in general, for battery powered LED/OLED lighting as it not
only constitutes a DSRC-I, which is preferably used for the above-named
applications, but due to the inventive topology, the supply circuit 1 in
addition provides a boost function providing a higher output voltage
Vout than the input voltage Vin without the need for additional
components. Above all, dimming of LEDs/OLEDs can be realized by
decreasing the switching frequency fswitch. Waveforms with decreased
switching frequency are shown in FIG. 6.

[0079] In a further embodiment, a control loop, i.e. a feedback loop, can
be additionally provided. The feedback loop would, for example, measure
the LED current or voltage, send this signal to the controller and adjust
the control signals of the electronic switches accordingly.

[0080] In summary, the novel topology of the supply circuit according to
the present invention offers basically the same major advantages as the
conventional DSRC-I converter, but additionally it provides a higher
output voltage Vout than the input voltage Vin.

[0081] Although the novel supply circuit may be considered to be
disadvantageous because of the conductive part over the two diodes
D1 and D2 for an output voltage Vout lower than the input
voltage Vin, in reality this will cause no problems, as the forward
threshold voltage of the connected load in particular LEDs result in a
high output voltage Vout. This blocks current flow if the converter
is not controlled.

[0082] Altogether, the converter provides the advantage that no current
sensing and current control is required, furthermore, a very compact
circuit design is provided with an integrated voltage boost. The supply
circuit according to the present invention is above all easy to design,
simple to control and provides a high efficiency. It shall be understood
that the same advantageous are valid for a device according to the
invention, comprising the supply circuit.

[0083] While the invention has been illustrated and described in detail in
the drawings and foregoing description, such illustration and description
are to be considered illustrative or exemplary and not restrictive; the
invention is not limited to the disclosed embodiment. Other variations to
the disclosed embodiment can be understood and effected by those skilled
in the art in practicing the claimed invention, from a study of the
drawings, the disclosure, and the appended claims.

[0084] In the claims, the word "comprising" does not exclude other
elements or steps, and the indefinite article "a" or "an" does not
exclude a plurality. A single element or other unit may fulfill the
functions of several items recited in the claims. The mere fact that
certain measures are recited in mutually different dependent claims does
not indicate that a combination of these measured cannot be used to
advantage.

[0085] Any reference signs in the claims should not be construed as
limiting the scope.