Since the publication of my article presenting
EXTRA-2 144 MHz Contest Preamplifier REV5 version [1], a total number of 150
preamps have been produced in 3 months, so the time has come to summarize the
experiences and consider the development possibilities.

First of all I would like to emphasize that
the original circuit met the demands of contesters. The primary design
requirement was the strong signal performance (IMD characteristics) and further
important requirements were good selectivity unconditional stability. However,
the desire for a low noise figure was pushed into the background, so in this
article I examine the possibilities of decreasing the noise figure while keeping
the excellent stability and selectivity of the preamp.

Many thanks to the hams using my preamp for
giving their feedback, and several suggestions which I tried to take into
consideration during the further development of this product. Special thanks to
DC8RI, IZ4BEH, DK5EW and HA1YA. This article describes only
the 144 MHz version, but under the influence of the many words of encouragement,
I have started the development of the 6 m and 70 cm versions using the same
ATF-51389 extra high dynamic range FET.

Built-in
Schottky diode protection for the input and
output (necessary for real-life operation)

Improving the
operation of the protection relay (special PTT
control is not needed)

Design and simulation

By omitting parallel feedback in the original
circuit, the noise figure will decrease by approximately 0.2 dB and the
amplification will increase by 2-3 dB. This increased gain could cause overload
of the following RX in some cases, and would only be acceptable if the
attenuation of the RX cable is high, or if it is possible to put an attenuator
between the preamp and the transverter. To meet the needs of a wider range of
users, a 100 Ohm potentiometer has been included as part of the output
attenuator, so the amplification can be easily adjusted over a range of about 6
dB. Table I shows the average values of the measurements of 20 completed
preamplifiers.

PMIN (0 ohm)

PMAX (100 ohm)

Gain [dB]

13,5

19,5

Noise figure can be decreased to as low as 0.3 dB by modifying
the input matching circuit, but series inductive feedback in the source is then
needed to avoid instability. This is not easy because the FET in its SOT-89
package needs to be cooled efficiently. The special layout developed for the
inductive source degeneration can be seen in the simplified circuit diagram ofFigure 1.A short tapered transmission line connects the source tab
of the FET package to a section of broader transmission line which provides both
the inductive feedback and good heat conduction to the large number of
through-board grounding vias.While working with
the test circuit I found that the optimal width of the broader transmission line
is 200 mil (0.200 inches, 5.08 mm). At this value the cooling and the inductance
are both appropriate. Then I examined the S11 parameter and the K stability
factor with a high frequency circuit simulator for various lengths of line.

Fig. 1: Simplified circuit
diagram

Fig 2: S11 versus source
inductors

Figure 2 shows the S11 results when the source inductance is
decreased from a line length P = 300 mil down to P = 0 (tapered section only)
and also for the case of zero source feedback. The corresponding variations in
the K stability factor over a very wide frequency range are seen in Figure 3.
In order to have suitable input match at 144 MHz, I first chose P = 300 mil,
which seemed to be the most preferable value. The simulated noise figure
increases slightly (0.05 dB), the K stability factor is excellent and the gain
decreases by approximately 2 dB. Ut in the prototype of this layout, the
relatively long series inductor caused too much positive feedback between the
output and the input, leading to oscillation. Based on that practical
experience, I chose P = 110 mil for the further simulations. No more oscillation
was observed and the wideband K stability factor is still adequate.

The high K value allows further optimization
of the input matching circuit to achieve a lower noise figure. At this point a
compromise is needed between s good board layout, acceptable selectivity and a
low noise figure. I examined all the possible input circuit configurations, and
found that the original configuration still met the mentioned requirements the
best, so the further optimizations concentrated on that input circuit.

Fig. 3: Stability
factor versus source inductors

Fig 4: S11 versus C1

Firstly, I examined the variation of S11
with the series input capacitor C1 and found 15 pF to be optimal (Figure
4). With that value fixed, I then optimized the C2 capacitor (Figure
5) and chose the green curve for C2 = 22 pF. The rectangular spiral
inductor then needs to be re-optimized, and the simulator allows the number
of turns (N) to be varied in ¼-turn steps (Figure 6). I chose the
green curve for 3.75 turns as optimum. Attentive readers may notice that I
did not choose the values of L1, C1 or C2 that gave the best possible input
match, for reasons that will be revealed in the next series of simulations.

Fig 5: S11 versus C2

Fig 6: S11 versus number of
inductor turns, N

The input circuit of the preamp needs to
be optimized not only for the good impedance match, but also to give a lower
noise figure. Figure 7 shows the variation of the noise figure with C1, and
15 pF proves to be an acceptable compromise between the impedance match and
the NF. This can be seen from a comparison between Figure 4 and Figure 7.

C2 also needs to be optimized by the same
criteria. Figure 8 shows the variation of NF with C2, and by comparing
Figure 8 with Figure 5 I chose C2 = 22 pF. Finally, the number of turns
of L1 needs to be re-optimized on the basis of Figure 9 and Figure 6.
N = 3.75 remains the optimal value.

Fig. 7: Noise Figure
versus C1

Fig 8: Noise Figure versus
C2

The simulated noise figure of the preamp
using all of these optimized component values can be seen in Figure 10.
The website of the manufacturer does not supply NFMIN data under
500 MHz for NF, so I had to apply linear interpolation during the
simulation. This can cause a small difference compared to the built circuit.
Matching to such high impedance requires very high-Q components to minimize
circuit losses. NF = 0.45 dB can be achieved assuming component Q values of
QC = 300 and QL = 160. To achieve the assumed value of
QC = 300, C1 and C2 will need to be low-loss microwave
capacitors. For L1, QL = 160 is achieved by a spiral inductor
optimized to maximal QL on the popular FR4(TanD =
0.02). This solution also provides easy reproducibility. The box slightly
mistunes the spiral inductor, so C2 consists of a fixed capacitor of 22 pF
with a small trimmer capacitor (CT) to provide accurate tuning.

Fig. 9: NF versus
number of inductor turns

Fig 10: NF using optimized
component values

Figure 11 shows the K stability
factor of the EXTRA-2 REV6 144 MHz contest preamplifier optimized according
to this article, between DC and 4 GHz. S-parameters of the preamp without
output filter can be seen in Figure 12 (this simulation assumes an
output attenuator of -10 dB). Figure 13 shows the characteristics of
the output filter. The input circuit also has some selectivity; it
attenuates -18 dB on 50MHz compared to 145 MHz, which may be useful for
multi-band contests. The S-parameters of the preamplifier with the output
filter and the -10 dB attenuator can be found in Figure 14.

Fig. 11: Stability
Factor

Fig 12: S parameters
without BP filter (Simulation)

Fig. 13: BP filter

Fig 14: Preamplifier
performance with BP filter

The above results were finally achieved
after more than 100 optimization processes, having found the best compromise
between input matching and NF in each process. The noise figure can be
further reduced to even 0.26 dB, but this makes no sense because the noise
of the galaxy is much higher than this at 144 MHz. For this reason, the
design gives more emphasis to input matching than to minimal NF.

I was not able to simulate the IP3
characteristics because the nonlinear circuit model of the ATF-53189 has
only be published by AVAGO Technologies for the companys own (expensive!)
ADS simulator [2]. Despite this, I modified the gate and the drain
decoupling circuits to increase the IP3 value. In the gate circuit C3 = 100
pF provides effective RF decoupling in the GHz range, C4 = 1 nF in VHF band
and C5 = 100 nF for frequencies below a few MHz. In the drain circuit C6, C8
and C9 are responsible for the same functions. The closer together the two
IMD test frequencies F1 and F2 are, the lower
frequency of the 2nd- order the IM product (F2  F1)
will be. Therefore, as input test frequencies F1 and F2
come closer together, higher capacitance is needed to achieve best possible
bypassing of this low frequencyproduct.

I also modified R1 to 22 Ω, being the
optimal value between the noise figure and damping the low-frequency
resonance of the gate bias choke L1, and slightly modified the setup of the
quiescent current resulting in more precise adjustments and even better
temperature stability. The Schottky diodes D1-D5 (VISHAY Semiconductor
BAS70-04) are for the protection of the FET. They have no significant
influence on NF or IP3 but are well proven (and valuable!) in
real-life operation.

By
inverting the DC control of the security relay K1, the input of the preamp
is safely short-circuited when the 12V supply is removed. When 12 V is
connected, the relay will switch off and the preamp will work properly. This
system works equally well when the preamp is powered either through the
coaxial output connection or separately via the feedthrough capacitor C21,
so separate connectors and cables are not needed for the relay control.

At the end of all these detailed design
considerations, the final circuit diagram for the EXTRA-2 REV6 144 MHz
contest preamplifier can be seen in Figure 15.

The layout of the PCB can be found in
Figure16. Dimensions are 72 x 34,5 mm which fits a standard tinplate
box. Figure 17 shows a panel of milled PCBs and Figures 18
and 19 show the finished PCB with its milled notches for the coaxial
connectors and the corners of the tinplate box.

If the PCB has been purchased already
assembled, skip to the next paragraph. The bare PCB as removed from the
milled panel (Figure 17) must be cleaned up with a fine file where
necessary. Solder the components on the PCB using solder paste and a hot air
soldering instrument. For ESD protection it is very important that the FET
must be the last component to be placed on the PCB.

Attach the 2 coaxial connectors to the
drilled box using the upper screws only. Then fit the PCB into the box from
the bottom side and solder the pins of the connectors. Solder the bottom
layer of the PCB to the sides of the box at the points shown on Figure 18.
A Weller PTC-8 soldering iron is recommended for this step. Now the lower
screws of the coax connectors can be fitted. The box has been designed so
that BNC, N-female or N-male connectors can be used. SMA connectors can only
be fixed by soldering, because the holes are not at the right place.

Only two further wires need to be
soldered. The bottom side jumper wire can be seen on Figure 18, and finally
connect the feedthrough capacitor C21 to the pad labeled +12V with a short
Teflon insulated wire.

Fig. 19:
Pre-assembled and pre-tuned PCB

Adjustments

Adjust the current limit of a 13.5 V
power supply to 250 mA and connect the preamplifier to it. The original
factory setting of the potentiometer P1 will require very little change to
achieve the necessary ID = 135 mA
value. In this case 135 mV must be
measured between TP1 and TP2 points. The total current consumption of the
preamplifier should be approximately 152 mA. The power supply voltage can be
varied between +10 and 15 V, without influencing the operation.

After setting the DC conditions,
potentiometer P2 must be set to maximum gain (turn fully clockwise). Tune
the BP filter L3-L4, using a spectrum analyzer and a tracking generator,
until the maximum of the selectivity curve is at 145 MHz. This requires 2
turns inward from the original factory settings of both cores.

Please note that the pre-assembled PCB
(Figure 19) has been pre-tuned on the assumption that the board will be used
inside the specified metal box. The box will move the preamp tuning upward
by 0.5 MHz while the gain will rise by 0.8 dB, and the unboxed boards are
pre-tuned to allow for this later change. Many hams like to use the unboxed
PCB version of the preamp for modernizing their own transverters or outdated
preamplifiers, so some further tuning may be necessary to allow for any
difference in the screening environment. Finally, in absence of a calibrated
noise source, the CT input trimmer capacitor must be adjusted to the middle
of its range. This setting gives the lowest noise figure, and based on
measurements, the gain decreases by only 02 dB compared to GMAX. The
preamp does not require any further adjustment, except for the gain
potentiometer P2 to a level that will maximize the dynamic range of your
particular receiving system.

The ATF-53189 provides the best
performance at frequencies above 500 MHz, which is why it is very difficult
on 144 MHz to achieve both the best input match and the minimum NF together.
In the case of minimal noise figure (0.3 dB) the input return loss is very
unfavorable, as S11 is only -2 dB and the input selectivity is virtually
zero. The input return loss of the REV6 version designed with the necessary
compromises can be seen in Figure 20. The measurement of S22 in
Figure 21 shows the good the good impedance matching of the output
filter. The maximal and minimal gain curves can be seen in Figure 22.
The 1 dB bandwidth of the preamp is shown in Figure 23 and the 3 dB
bandwidth in Figure 24. These figures show the excellent selectivity
and suppression of broadcast frequencies. This is provided not only by the
output BP filter, but also the input matching circuit contributes to it.
Figure 25 shows the transfer function of the EXTRA-2 REV6 preamplifier
between 20 and 500 MHz. The measurement results of the S parameters coincide
with the simulation curves shown in Figure 14. The Third Order Intercept
measurement of the preamp can be seen in Figure 26 (at maximum gain).

Fig. 20: Input
Return Loss

Fig 21: Output Return Loss

Fig. 22: Minimal and
Maximal Gain

Fig 23: 1 dB Bandwidth

Fig. 24: 3 dB
Bandwidth

Fig 25: Transmission in
wideband

The measurement of the noise figure is
shown in Figure 27. This figure shows the measurement results of one
of the noisiest examples tested, including the loss of the input relay K1
(approximately 0.15 dB). Figure 28 shows a possible wiring mode of
the EXTRA series preamplifiers in the antenna relay box.

Fig. 26: TOI
measurement

Fig 27: NF measurement

Results

I have tested approximately 30 pieces of
the EXTRA-2 REV6 preamplifier and the results are nearly the same. The
circuit is stable and easily reproducible. About 20 pieces of the preamp
have been tested by contest stations in IARU Region I VHF Contest, and the
experiences have proved the measurement results and the simulation. It can
be stated that the high-level behavior and the selectivity of the preamp is
outstanding in real-life situations, and the noise figure is certainly low
enough.

It was these positive experiences with the
144 MHz version that encouraged me also to develop the 50 MHz and 432 MHz
versions of the preamp. Figure 29 shows the EXTRA-6 50 MHz contest
preamp, and Figure 30 the EXTRA-70 for
432 MHz. On this latter frequency
it is possible to achieve both an optimum impedance match into the FET at
the same time as a very low noise figure of 0.38 dB.