SE
OUTPUT STAGE CONFIGURATIONS.
This page explores single ended output stage properties with beam
tetrodes and pentodes.
The aim is to produce the best audio fidelity which includes :-

1. Maintaining high enough gain of output tubes with local NFB
while enabling drive voltages at low
at low distortion.
2. Minimizing any reliance on large amounts of global NFB.
3. Locally applied NFB within the output stage to reduce
distortion and Rout to very low levels to
give better performance than triode strapped tubes.
4. Maintaining unconditional stability which means no possible
combination of L, C or R loads, or with
no load at all can cause oscillations at any frequency.
5. Ensuring that stability is possible with critical damping
R&C networks to reduce gain and open loop
phase shift below 20Hz and above 20kHz.
6. Maintaining bandwidth of 10Hz to 65kHz with pure resistance
load at the -6dB output Vac level, ie,
1/4 full rated power.
7. Ensuring that output power at -1dB below clipping is possible
without any stage of the amp being
overloaded of forced into grid current draw between 20Hz and
35kHz.

History.
Single ended output stages have been used for audio amps ever
since the first amplifiers were used for radio
and audio frequencies after the invention of the triode in 1903.
Up until about 1960 nearly everyone in the
world listened to radios and TV sets which had one tube devoted to
powering a loudspeaker.
Push Pull operation could give more power than most ppl ever
needed, and were initially more expensive to
produce than using just one output triode. But once beam tetrode
and pentodes were invented, PP amps
became popular because they could operate in class AB with low
bias current. Such PP amps required less
power from batteries or mains so construction costs declined which
suited manufacturers.
However, if a single large tube or a number of paralleled tubes
are used in pure class A1 audio power, the
sound produced may be better than many PP amps. I'd rather have
one KT88 in triode making 8W than
having two 6V6 in PP making 10W in class AB.

Content of this page is based around schematics.
Fig 1. Three most used basic SE amp stages, 13W to 10W, SE pure
tetrode, SE Ultralinear, SE triode,
1 x KT88/6550.
Fig 2. 20W+ amp with SEUL with 2 x EL34, KT66, KT88, KT90, KT120,
etc.
Fig 3. The Equivalent Model of KT88 with g1 and g2 inputs treated
as current generators.
This allows understanding of operating properties of a KT88, and
its Ra, gm g1, gm g2, and to analyse
all voltages and currents in all electrodes to determine voltage
gain, and effect of local NFB.
The theory may be applied to all power tubes including pure
triodes which do not have a screen, such as
300B and 2A3.
Fig 4. 20W+ amp with CFB, with 2 x EL34, KT66, KT88, KT90, KT120,
etc, using same SEUL OPT in Fig 2.
Fig 5. 36W SE amp with CFB with 3 x KT88 etc, using SEUL OPT, 25%
UL tap, 1k3 : 5r6 Z ratio.
This was designed to include "choke sink" for cathode current, and
choke in anode feed for driver tube.
Probably nobody has ever built an amp like this because they need
to source good quality chokes.
The THD and Rout is much lower than conventional SEUL amps.
Fig 6. 25W PSEUL + CFB amp designed around the Hammond 1640SEA
output transformer with mosfet
CCS at KT88 cathodes.
Fig 7. 25W PSEUL + CFB amp designed around the Hammond 1640SEA
output transformer with choke
at KT88
cathodes.
Fig 8. SE CFB output stage and SEUL output stages with OPT with 3
windings.
Fig 9. Choke Feed SEUL and SET output stages.
Fig 10. Choke Feed SET amp with 845.
Fig 11. Choke Feed SEUL with floating B+ supply.
Fig 12. Choke Feed SET with floating B+ supply.

There are a number or ways to arrange output tubes, OPTs, and
driver stages and NFB.
It is impossible to always consider the output stage entirely
separately and without the interaction of the
driver and input stages and the NFB loops.

BASIC SINGLE ENDED AMPS.
Everyone should be familiar with the basic class A working of a
single vacuum tube, or 2 or more paralleled
in a circuit with tubes, OPT, and PSU all in series, with an idle
current which is varied between zero idle
current and twice idle current, at maximum audio output power.

The operation of the tube will always remain predictable in terms
of applied rail voltages and applied signals to
all electrodes. The Ea, Eg2, Eg1 and Ia may all be varied within a
range to give ideal working with a chosen OPT.
Fig 1. Basic configurations of SE amp stages.
The main 3 varieties of "Single Ended" output stages used in many
simple tube amps are shown here.

I am using a modern Svetlana KT88 to show what is possible but
other brands of KT88 or 6550
could be used similarly. The general principles apply to the range
of power tubes now readily available,
could be KT120, KT88, 6550, KT66, 6L6GC, 5881, 807, 6V6, 6CM5,
6CA7, EL34, EL33, EL84,
EL86, GU50, 13E1,etc, etc.

SE Beam Tetrode or Pentode.
See Far Left side of Fig 1.
The use of single ended amps with one beam tetrode or power
pentode offers high class A efficiency of up
to 45% but with high THD with a dirty mix of both even and odd
numbered harmonic products from low
levels to clipping level. The THD mix of H varies much with load
value. The IMD is appallingly high and
Rout is high and to make such a tube configured here work far
better, at least 20dB of GNFB must be used
from OPT sec back to an input port of an input tube, usually its
cathode. Many such amps have been built
with an output stage like the one shown and all seem to me to be
rather damned awful. Put it this way, if you
use an EL84 to make 4 Watts in pure SE pentode mode, and with 20dB
GNFB, then its sound just is not
as good as using a 2A3 to make 4W with say 12dB GNFB. Millions of
AM radios made before 1955 had
a lone 6V6 for the audio amp and no GNFB. Although the sound of
music was awful the set was cheap
because most ppl were poor, and you could understand the nightly
news and cricket scores, and be
manipulated by politicians and advertizing. Beam or pentode tubes
are easily driven because gain is highest,
and drive for class A1 never involves class A2, where grid voltage
goes positive during input cycles so
grid current is then drawn, with huge reduction in grid input
resistance.

SEUL Beam Tetrode or Pentode.
See Center of Fig 1.
The use of a tap along the anode primary winding was rarely ever
used by any manufacturer because it cost
more money to put the tap there, and CEOs hated innovation because
it always cost money, and CEOs
don't always agree with the marketability of innovation. Besides,
the idea of using an "Ultralinear" tap was not
invented until maybe just after WW2. With a tap at 40% to 65% of
anode primary turns the beam tetrodes
or pentodes acquire triode like harmonic products, less IMD, while
maintaining anode efficiency at over 40%.
Gain is less than pure beam / pentode and grid drive remains free
of grid current. There is nothing "ultra"
about the linearity of having a screen tap for "partial triode"
operation, but in general the UL connection
makes the tube work very much better than otherwise, and
measurements and sound heard will verify it.
Some GNFB is still needed, usually 15dB.

SET - Single Ended Triode.
See Far Right of Fig 1.
In the early days of electronics only one triode was used for
audio amplifiers. The idea remains well
respected as we all see with 45, 2A3, 300B, 211, 845, GM70 etc,
for SET triode power from 2W to 22W.
With real triodes listed, there was no screen grid connection
shown above, and old OPTs never had a
screen tap on OPT. All the common varieties of beam and pentode
power tubes may be configured as a
triode by connecting screen to anode. The screen then carries ALL
the anode signal. This exerts electrostatic
control of the electron stream in similar manner to the control
grid. The gm of the screen g2 is usually
between 1/20 to 1/6 of the gm of g1 control grid. The screen
performs the task which the anode in a real
triode performs. This function is an application of local NFB.

However, the function is not a linear function, so despite the
high amount of inherent NFB in triodes or
triode strapped multi grid tubes, there is usually about 5% THD at
full Po on the one load value which
gives maximum Po. The efficiency of triodes or
triode connected tubes is between 15% to 33%, depending on the
tube operating conditions. Higher
efficiency of up to 40% may be gained by using a direct coupled
cathode follower to drive the triode
grid for class A2 operation. Usually class A2 is more bother than
its worth and if more triode Po is
wanted, use more triodes and stay in class A1. A renowned
exception is Audio Note "Ongaku" amp
with 211, which has a 6SN7 cathode to drive its grid. Not all beam
and pentode OP tubes can be easily
driven in Class A2 such as EL34. Grid current is just too high.

It is difficult to make a bad sounding triode amplifier providing
no tube is driven into any overloaded
operation. Triode mode is the "safest bet to good music". The Ra
of triodes is usually much less than
the anode load driven. At low levels THD and IMD can remain quite
low enough so GNFB may not
be needed. THD at all levels is mainly 2H. Gain of triodes is
perhaps 1/3 that of beam or pentode
tubes and the drive voltage level may be quite high, up to about
110Vrms with 845. But providing
enough drive is never difficult by use of another SE triode driver
tube. The 2H of the driver cancels
part of the 2H produced in the output triode. So at all levels up
to clipping, some SET amps produce
surprisingly low 2H which gives less objectionable THD and IMD
than from a PP amp of the same
power.

Few radio sets ever had slightly larger output tubes than 6V6 or
6F6 such as KT66, 807, 6L6, or
later EL34, KT88, 6550. But fabulous sound quality could be had
from these tubes strapped as
triodes. In fact, in many radios I have re-wired, it is always
possible to get at least 3W from a
single EL34 strapped in triode and power is the same as one 6V6,
and fidelity with trioded EL34 is
far superior to a 6V6.
---------------------------------------------------------------------------------------------------
Russian EH 6550, KT88. Tubes such as KT90EH and KT120EH may always
be used with slightly
higher Ia, cathode bias Rk to suit, and a lower RLa load to be
used while obtaining the same Va,
hence SE Po up to 15Watts is possible. But an OPT meant for 4k2 :
4r,8r,16r and Ia = 100mA
max cannot be pushed by increasing Iadc much beyond 100mAdc.
Notice that for the same B+ supply, and similar Ea and Ia, the
triode load value is lower than for
tetrode or UL. See my pages on load matching to SE triodes and
pentodes to work out operating
conditions different to the above.

To get better triode performance the B+ usually needs to be say
15% higher than for the same
tube with UL. Then the same OPT can best be used with triode. In
the above example for SE Triode,
the OPT can be 4k2 : 4,8,16r.

If the B+ is raised to +490Vdc, Ia +Ig2 reduced to 70mAdc, EK =
+52V, and Pda+g2 remains 30W.
Triode Po then becomes 10W. KT88 or 6550 then give very similar
performance to 300B.
----------------------------------------------------------------------------------------------------
Fig 2. 20W SEUL.
Fig 2 shows a very ordinary 20W Ultralinear output stage using a
pair of EL34 tubes in parallel,
and based on using an available Hammond 1627SEA OPT with ZR = 2k5
: 4r, 8r, 16r.
My comments below are based on using EL34 or 6CA7, but parallel
pairs of other tubes such as
KT88, 6550 can be used.
There is no NFB applied in the EL34 output stage except for the
local "ultralinear screen tap" which is
commonly not considered local NFB, ( although it actually IS NFB
). The UL tap allows the high power
of class A1 pure pentode but with triode like harmonic products,
ie, less odd number H and more even
number H. Ra is reduced greatly so a better damping factor is
possible than for pure pentode / tetrode.

All triodes or triode strapped pentodes or beam tetrodes have
internal NFB by means of the field
effect of the anode interacting with the field effect of the
control grid to give a resultant effect on the
electron stream from cathode to anode. See my equivalent models of
multigrid tube basic operation
with Ra in shunt with g1 and g2 current generators.

For multi-grid tubes such as EL34, KT88, having Eg2 a fixed Vdc
hugely reduces the signal field
effect of anode upon Ia current. The screen g2 may be connected to
a source of signal voltage
which is a fraction of the anode voltage and this has an opposing
action to whatever the g1 signal
voltage tries to do.

A diagram about basic tetrode of pentode operation may explain
more lest I completely confuse everyone.
Please remember that the beam tetrode has no suppressor grid as
found in the pentode, but has beam
forming plates connected to the cathode internally which function
like a suppressor grid g3. There is no
need for mention of the suppressor grid or beam formers again
because the relevant operation facts
are concerned with control grid g1 and screen grid g2.

If you can follow and apply what you find in my Fig 3 diagram, you
will find it useful to analyze possible
signal voltages and currents like our ancestors did, and all
without PCs and simulation programs.
Understanding models frees you from any need for a PC. But
unlike our ancestors who used a slide
rule you may use a $5 pocket calculator.

Fig 3. SEUL basic tube model
Fig 3 above shows an equivalent model for what is happening with 1
x KT88. Other tubes such as
6550, KT90, KT120, 6L6GC, 5881, KT66, 6V6, EL34, EL84, etc, will
show similar basic
operation, and by understanding the model you can analyze any
output stage or design one.
But you MUST start by knowing the g1 gm and g2 gm, and the Ra for
the tube operating
conditions of idle values of Ea Vdc and Ia dc.

In the above Fig 3, the figures for signal current and voltage
amplitudes have been prepared from
measurements of transconductance of g1 and g2 with tested samples
of tubes under the conditions
listed for Ea, Ia, Eg2. The Fig 3 figures above don't apply to all
output tubes but will agree with curves
shown for Svetlana KT88, and other Russian makes of KT88 and 6550.

Testing tubes in a real world amplifier isn't difficult after you
learn what you are doing which may
take days of practice.

Hum voltages present in output stages may spoil the attempts to
measure small signal voltages in an
amp. To avoid noise interfering with measurements, ensure there is
adequate filtering of the B+, and if
not, apply much larger filter caps and perhaps a choke between
rectifier and the existing B+ filter caps.
This is very necessary in SE amps where there is no common mode
rejection of noise in B+ rails.

But let us suppose you have an existing SE amplifier with a
multigrid output tube, and you have well
filtered B+ rails for anode, and g2 and input stages.

To determine g1 gm :-
The g1 gm is the g1 transconductance which is the ability of the
g1 grid to vary the electron flow
and is expressed in mA / Volt.
1. Connect 100r resistance between anode and OPT anode connection.
2. Connect wire shunt link with alligator clips from OPT anode
connection and B+ connection, to short
circuit the primary of OPT.
3. Make sure the screen g2 has the intended fixed supply voltage.
4. Disconnect any global NFB and output speaker loads.
5. Connect audio signal gene to amp input and use a 1kHz sine wave
and increase until there is 1Vrms
at the output tube g1 grid.
6. Measure the signal voltage across the 100r and have a CRO
connected to anode to make sure
distortion is less than 2%.
Suppose you measure 0.64Vrms ( Vac on DMM ) across 100r.
7. Calculate g1 Gm. The current in 100r = 0.64V / 100r = 6.4mAac.
This means the 1V change at g1
produces 6.4 mA change at anode so gm = 6.4mA / Volt. Simple?
The anode signal voltage variation will cause virtually no effect
on the gm measurement because the
gain between g1 and anode is negligible with a very low RLa of
only 100r. Hence there is no action
of internal NFB within the tube.
8. For triode connected tubes, the same set up works OK but g2 is
connected to anode, and the gm
measured will vary slightly, because g2 signal current is
generated in addition to the anode current.
Triode Ra may also be neglected.

The load of 100r is a near vertical load line is drawn upon data
sheet Ra curves for KT88. We wish to
understand the basic current change behaviour caused by grid
voltage change.

Measuring g2 gm :-
The g2 gm is the screen g2 transconductance which is the ability
of the g2 grid to vary the electron flow
and is expressed in mA / Volt.
1. The above set up is used with 100r anode to B+ load, and OPT
primary shunted by a wire link.
2. Ground g1 with 2uF so there is no g1 signal possible, but the
wanted g1 Vdc bias is undisturbed.
3. Disconnect g2 screen is from wherever it is normally connected,
and insert 1k0 in series with g2
and original g2 B+ supply, so that g2 retains its Vdc bias supply.
4. Connect a signal gene between 0V and g2 but use a 2uF DC
blocking cap to enable up to 5Vrms
at 1kHz to be applied to the g2.
5. Measure the signal voltage across the 100r anode to B+
resistor.
Suppose you measure 0.45Vrms at 100r. It means 5V at g2 produces
0.0045 amps of anode current.
6. Calculate g2 gm = 0.0045A / 5.0V = 0.0009A/V = 0.9mA/V.

Measuring Ra :-
Ra is the internal dynamic resistance of the tube when operating
in pure beam tetrode or pentode,
and it always exists in the current gene model and it varies
depending on Ia and to Ea.
Ra is a difficult parameter to measure accurately for beam and
pentode tubes because it is a high
number of ohms. Ra will be easier to measure if there is a UL tap
or triode connection, and then the
Ra is the beam or pentode Ra IN PARALLEL with the effect of the g2
current generator.
When the g2 signal voltage remains the same as the cathode voltage
( with a cap bypass between g2
and k ), then the g2 generator has infinite impedance between the
anode and k.
1. To attempt to measure pure tetrode or pentode Ra, disconnect
all shunts across
OPT winding and have g2 taken to the wanted B+ fixed voltage.
2. Remove any R&C zobel networks across the OPT secondary or
primary.
Remove any secondary load or global NFB connected.
3. Make sure cathode is well bypassed to 0V.
4. Connect 1k0 between anode and a signal gene with DC blocking
cap = 1uF.
((( This can be tricky if you have a solid state signal gene
because such test gear can be fried to a
useless crisp if their outputs are subject to external voltages
exceeding say +/-20V. So you need to
have adequate protection measures on inputs and outputs of ALL
TEST GEAR !!!!!
Using a tubed cathode follower in a separate box is good practice
for connections between delicate
SS test gear and tube gear.)))
5. Ground g1 of power tube with 2uF cap and ensure g2 is grounded
without a series screen stopper
resistance.
6. Apply 10Vrms at 1kHz to input side of 1k0 resistance from
signal gene.
Measure the anode to 0V Va-c, and record this.
7. Measure the Vac across the 1k0.
Suppose you measure 0.4Vac across 1k0. Then current flow from
anode to 0V = 0.4V / 1k0 = 0.4mAac.
Suppose you measure Va-c to 0V = 9.6Vac.
8. Calculate Ra, dynamic anode resistance between anode and
cathode = 9.6Vac / 0.0004Amps ac = 24k.

You will quickly realize that using the published data sheet
values for gm and Ra will lead you to gross
mistakes in calculations of what is going on in your tube. A KT88
with Ia = 60mAdc may have g1 gm =
6.4mA/V, but the data sheets tell us g1 gm = 11mA/V at Ia =
140mAdc. Nobody would ever have a KT88
at idle with 140mAdc.

Let us suppose the OPT Lp = 25H, and at 1 kHz this has inductive
reactance XLp = 157,000 ohms,
with perhaps 400pF of shunt C which makes XC = 397k.
The XLp, and XCsh will be found to have a negligible loading
effect at 1kHz because usually you will find
Ra is always below 50k for most large multigrid output tubes
such as KT88, KT66, EL34. So you may
ignore the current flow from signal gene through primary
inductance and any shunt capacitance.
But of course there are some very badly made amps around and you
can't assume XLP and XC will
be negligible. So measure these parameters as well. Always avoid
making any stupid assumptions.
Measuring triode Ra in the manner above but with g2 tied to anode,
expect Ra to be 1k1 for KT88.

To measure the amplification factor µ.
The amplification factor µ is the calculation of gm x Ra. It is in
fact the voltage gain of any tube where
a grid causes anode voltage change without any current change.
This means the anode load
= infinite ohms, and the loadline for an infinite load is a
horizontal line across the data Ra curve sheets
for Ea and Ia characteristics. The µ of beam and pentode tubes is
the most constant parameter, with large
variations of gm and Ra. Triode operation makes µ much more
constant for wide Va swings.
1. Set up the amp as it would be for normal operation, but with no
global NFB loop, no loads or
zobel networks.
2. Apply 1kHz sine wave to output tube grid g1 via 2uF cap. Use
DMM or hi-Z input volt meter to
measure Va to 0V.
3. Connect CRO to monitor the Va signal distortion.
4. Increase g1 signal to obtain Va 50Vrms. All output tubes should
be able to achieve this much voltage
with no load, and with THD < 5% for multigrids and less than 1%
for triodes.
5. Measure the grid input voltage. Suppose you measure 0.312Vac.
6. Calculate voltage gain and µ without any load = Va / Vg = 50 /
0.312 = 160. Gain is just a number,
no units. It may be considered a negative number because a
positive going Vac at g1 or g2 produces a
negative going Vac at anode.
The voltage gain without any signal load current = amplification
factor, µ.

7. Verify measurements and calculations. For all tubes and for any
g1 or g2 input,
transconductance gm = µ / Ra.
If we knew gm, and Ra, we could calculate µ. From the above
examples, g1 gm = 6.4mA/V and Ra
= 24k, so µ for g1 = 0.0064 x 24,000 = 153.6, close enough to the
160.

The Ra in Fig 3 is always present as a shunt resistance in Fig 3
and is in parallel with the effect of the
g2 current generator if there is a signal voltage applied between
g2 and cathode. Thus the g2 generator
can become equivalent to a resistance.

Consider Fig 3 with an OPT set up with a 25% UL tap for g2 so
while there is +100Va-c+ at anode,
+25Vg2-c appears at g2. The +V at g2 increases current flow. it is
calculated = Vg2 x g2 gm
= 25 x 0.9mA/V = 22.5mA. If you went looking for the 22.5mA, you
would not find it, because it is a
"useful imagined current" that exists in a mathematical model to
help explain what we do measure in a real
circuit.
Consider Fig 3 without the 4k0 RLa anode load. There would still
be the imaginary 22.5mA with Va = 100Vac.
But there would be no load current. The Ra of 25k is there, so the
current in Ra = 100V / 25k = 4mA.
Total Ia model current change = 22.5mA + 4mA = 26.5mA.
The effective Ra = Va / Ia = 100V / 0.0265A = 3,773r.
The portion of this resistance due solely to g2 generator = 100V /
0.0225A = 4,444r.

The effective Ra' for where a screen signal exists = Ra // ( 1 /
UL fraction x g2 gm ), where Ra is that for
pure tetrode or pentode.

From the calculation of Ra' for any % of UL, we can also calculate
the UL g1 µ.
It will always be be less than pure beam tetrode / pentode µ and
more that triode µ.

µg1 = g1 gm x Ra, and if 25% UL Ra = 3,773r then UL g1 µ =
0.0064A/V x 3,773 = 24.15.
The screen has its own transconductance value g2 gm, based on
having no g1 signal. The pure beam or
pentode Ra of the tube remains the same if the screen is used
alone for Ia change and Va change.
µg2 = gm g2 x Ra = 0.0009A/V x 24,000r = 21.6.
Screen gm is so much lower than g1 gm that screen drive is seldom
ever used, so pure screen driven
output tubes are not discussed here. Feel free to experiment to
know more about screen drive properties.
But try as you may, you will find the model in Fig 3 very useful
if you understand the concept of the basic
input voltage controlled current source which has infinite
resistance looking into its output at the top of the
two circles on the drawing.

The voltage gain, A, with a load for UL can also be calculated A =
µ x RL / ( RL + Ra ).
For 25% UL, in this case, A = 24.15 x 4,000 / 7,773 = 12.42
For 100V into 4k0 load, the input voltage = 100 / 12.42 =
8.05Vg-c.
The only confusing thing remaining is the + / - sine for Vac. All
tubes in "common cathode" mode like in
Fig 3 are inverting, ie, a + input voltage causes a - output
voltage. So as said before, in effect, µ is negative,
so -8.05Vg-c produces +100Va-c.

The maximum THD of SE output stages without local FB is highest
for pure pentode or tetrode and can
be 15% for where the RLa is that used to get maximum possible Po
and at 1dB below clipping.
Many even and odd H products are present at all levels.
For class A1, the amount of UL % can be up to 70% for most
multigrid tubes and the Po max without
grid current is nearly the same as pure tetrode or pentode. Odd H
are much reduced and THD will be
around 7% of mainly 2H. But where the UL connected tube produces
the same Po as lesser maximum
for pure triode operation, THD may be less than the triode.

The less THD one has before NFB is applied, the better is the
music. The FB network allows a fraction
of the output distortion H to be applied to an amp's FB input. The
input tubes and output tubes are not
perfectly linear, and the signals fed back create distortion H, so
intermodulation, or "IMD" H are generated
which are the sum and difference between any two frequencies. If
you have 3H fed back with fundamental
test tone of 1H, then the IMD products will be 3+1 = 4H and 3-1 =
2H. Their level will depend on the
non-linearity of the amp devices.

So by means of intermodulation, H products appear at the output
which were not present when the amp
was tested without NFB. It is a very real problem where one has a
pure beam tetrode making 10%
THD and where only 10dB of global NFB is applied, and one will
find some reduction of 2H but increase
of 3H and 5H and perhaps other intermodulation IMD H that were not
produced by the beam tetrode.
Where the signal is music with perhaps 20 frequencies present,
they all react to modulate each other even
without any NFB applied and when NFB is applied in small amounts
the fidelity betterment is less than
expected if the THD for a signal tone was 10% to begin with. With
10% before FB and with only 8dB NFB
applied, the reduction of THD and IMD will not be 8dB, and the
sound may not seem any better. But a tube
making 10% THD without NFB will measure 1% THD with 20dB NFB.

The "second order" IMD H products may be less than 1%. But as the
H number rises, their audibility
increases by factor of N squared / 4, where N is the H number.
Hence it takes very little 7H to make an
amp bad where the input signal = 100Hz, and the 7H = 700Hz. If you
have 100Hz + 500Hz present
at high levels then you get IMD H at 400Hz and 600Hz and if these
H are NOT musical tones in the
music scale their presence lessens perception of fidelity because
they are not in harmony with music.

The THD if considered alone make little difference to perceived
fidelity if it is below a few % because
most musical notes have many H already present and altering their
levels by a few % makes little
difference. But where the THD does measure above 0.5% the
perceived fidelity begins to suffer because
of the INEVITABLE production of IMD as a result of the non
linearity which we measure by using a
single pure sine wave input to assess THD and the resulting H
spectra.

With music, if you could listen to all the IMD products and THD
products but without the original
undistorted music, you would hear a bad sounding non harmonious
noise rising and falling in level
according to the music levels. It is disgusting junk that poisons
our enjoyment.

Applied FB should effective over a wide F range, and open loop
gain of the amp without any FB
should be wide as possible, and phase shift minimized between 10Hz
and 30kHz. What is needed to
minimize production of IMD with output tubes is to get them to be
linear by NFB applied locally
without involving global NFB around 3 amp stages. The use of CFB
and UL taps convert non linear
beam tetrodes and pentodes to being quite different in that the
initial spectral complexity is reduced and
simplified more than NFB theory would predict.

The amount of global NFB applied can then be quite low without
causing significant "extra H" IMD
products providing we make the input and driver stages linear. In
most tube amps, THD produced by
driver tubes is much less than output tubes. So multigrid tubes
benefit with UL taps, and more with triode
connection, and even more with CFB with a tertiary winding on OPT.
Adjusting the signal level applied
between screen and cathode is very important along with setting
the screen grid Vdc, Eg2, to be just right.

Usually, although most beam tetrodes and pentodes produce
atrocious THD levels of up to 15% in pure
beam / pentode mode at just before clipping, they can be cajoled
into producing less than 2% with local NFB.
It means their gain is reduced from say 20 to 3, which means that
instead of 10Vrms grid drive we may need
70Vrms, but this is easy, and at low THD below 0.5%.
-----------------------------------------------------------------------------------------------------------
Local CFB in the OP stage?

SE amps with CFB will normally have the OPT primary winding
divided into 2 windings, with between
70% and 90% of the turns in the "anode circuit" and the remaining
30% to 10% of turns in the "cathode
circuit". The signal Ia flow and Idc flow is equal in both
windings except for where a minor difference
occurs with dc and ac screen current. An example is at se35cfb-monobloc.html
There is no strict ideal ratio of anode turns to cathode turns and
I have used up to 2:1 successfully in my
SE32 amps, 2012 version, with 13E1 tube, ie, anode turns = 66.7%
of total, cathode turns = 33.3%.

Hardly anyone has the patience, time, money or skill to make their
own well designed OPT to suit multiple
paralleled output tubes with CFB windings. Few companies produce
any OPTs with any CFB windings as
a stock item you buy off the shelf. So OPTs for CFB use usually
have to be custom wound at a far higher
price than "normal" stock items, and you wait months and deal with
bullshit artists. There is little doubt my
efforts to make the SE35 and the SE32 creates sound quality to
keep amp owners happy ( 2014 ).

But now let us question Fig 2 above.
Could the standard off the shelf Hammond OPT be used to provide
local cathode feedback in the
output stage, despite the fact there is only ONE "anode" primary
winding?

Is it possible to "build the tubes around the OPT"?

Fig 4.
The above Fig 4 uses a floating B+ anode supply which in effect
works like a 400V battery between
the tube anodes and the anode connection of the OPT primary. The
UL tap is connected to 0V.
The normal connection for B+ is taken to cathodes and their
biasing R&C networks.

This arrangement has anode Idc and signal Iac flowing around the
whole series circuit formed by tubes,
primary winding, and PSU, just like any conventional amp. But here
I have the whole primary at 0Vdc
potential with UL tap taken to 0V and the non grounded power
supply is moved to between tube anodes
and anode connection of OPT. The tubes are operating in exactly
the same manner as they would in Fig 2
and relative signal voltages are the same.

There is a total of +229Vac across all turns of the primary.
With the 40% UL tap, there is -92V from one end of primary to tap,
and +137Vac from tap to the other end
of primary. The tap is connected to 0V, and cathode is connected
to smallest of above Vac, ie, -91Vac.
the tap, o, Vac = 0.4 x 229Vac = 91.6, and there ia UL %But
-91Vac between cathodes and 0V,
and +138Vac between anodes and 0V. The anode and cathode have
oppositely phase Vac.
To make this happen, -22Vac must be applied between grid and
cathode, and it must be same phase as cathode
so that Vg-0V = - ( 91Vac + 22Vac ) = -113Vac. The open loop gain
without the strange connection
= Va-k / Vg-k = 229V / 22V = 10.4. With the strange connection,
there is 40% CFB and closed loop gain
= Va-k / Vg-0V = 229V / 113V = 2.02, so gain reduction factor =
2.02 / 10.4 = 0.194, and THD reduction
may be from 5% at clipping to 1% at clipping, and effective Ra'
< triode Ra.

A normal SEUL stage as in Fig 2 has the screens taken to a 40% UL
tap and there is no CFB with cathodes
at Vac = 0V. open loop gain will be 10.4 with 22Vac at Vg-0V
giving Va-0V = 229Vac. In Fig 4 above the
relative Vac between anode, screen, cathode and grid to cathode
are the same as Fig2, but the UL Vac is
applied in series with grid. Screen g2 must be at a high +Vdc
potential so there must be an additional fixed
B+ supply rail applied between screens and UL tap. This B+ rail
supply is just a normal grounded B+ supply
which will be also used for the input and driver stages.

Fig 4 output stage has low overall gain 2.02 compared to normal UL
and as long as the driver stage can
produce up to about 120Vac with THD < 1% there is a very
worthwhile advantage because EL34 THD
and Ra is much reduced with 40% CFB.
Ra of each UL EL34 = 2,700r, UL µ = 19, and ß = 0.4, so Ra' =
2,700 / [ 1 + ( 19 x 0.4 ) ] = 314r,
and with 2 parallel EL34, Ra' = 157r, and if RLa = 2k5, damping
factor = 15.9.

The driver Vac must be high up to 120Vac so it should be EL34 or
EL84 in triode mode and its 2H should
cancel the similar level of 2H of output EL34. An example of 2H
cancelation is at description of SE32 amp.

There is no absolute need to have a floating PSU B+ supply for the
OP tubes as in Fig 4. A normal grounded
B+ supply may be used as I have it here....
Fig 5. 25% CFB using 25% UL tap.
Fig 5 shows a complete amp using a conventional SEUL OPT with just
one tapped primary winding. I designed
this schematic for my friend Vali in Romania. He wanted to make a
2 channel chassis and using 3 x KT88 for
each channel. He found a supplier for SE OPT for 30W+ into 1k8 :
8r0, with UL taps at 25% and 50%.
The properties of the OPT were analyzed and after some
calculations, I thought 3 x KT88 could make 36W
into primary load of 1,300r , with sec = 5r8. Thus he ought to get
at least 30W at the 5r8 output load.
The amp would tolerate all loads between 3r0 and 20r0.

But I have since made a few improvements to the schematic and I
know that anyone else wanting to make such
an amp may not be able to source the same OPT which my friend Vali
found. So therefore I have drawn the
OPT having UL taps at 25% and 50%, and with tapped secondaries for
speakers nominally 4r, 8r, or 16r.

In Fig 5, there is NO FLOATING B+ anode supply as shown for Fig 4,
and there is just ONE B+ supply used
for all B+ rails for the amp. The B+ apply of +410Vdc is applied
to the 25% UL tap so anode current flows from
B+ to the normal anode connection. The KT88 anodes have 75% of the
total primary voltage, in this case 162Vrms.

Having Idc flow in 75% of the primary winding means the DC core
magnetization Bdc is only 75% of what it would
be if the DC flowed in all turns. Instead of say Bdc = 0.7 Tesla,
it is 0.53 Tesla. So Bac can be 133% higher at 0.93
Tesla, instead of 0.7 Tesla. So the frequency of core saturation
will be 0.75 x original design Fsat, so if Fsat was say
25Hz for full Po level it would be reduced to 19Hz, a considerable
improvement. OR, one could have Fsat at 25Hz
but at a considerably higher signal Vac. In other words, with the
same Idc in fewer primary turns the output power
can be higher.

The signal current produced by KT88 flows in ALL the primary turns
including those without any Idc. The original
B+ connection on the primary is taken to the KT88 cathodes via
C20, 470uF. So where does the Idc flow from
cathodes? It flows down to 0V and back to PSU via L2, 5H choke.

In this case, the OPT had a 25% UL tap which is an ideal % for
deriving CFB. Some available OPTs may have
UL taps at somewhere between 25% and 50%. The 50% UL tap is in
fact the CT of the primary winding.
If the 50% UL tap was used for B+, then in this case the anode
signal would become 108Vrms- and cathode
signal would be 108Vrms+, and the the grid-to-cathode drive signal
would be about 20Vrms+ so the EL34 in
triode mode would need to make 128Vrms+ which is getting a bit
high to achieve at low THD < 0.5%.
But in my SE32 with 13E1 I have an EL34 in triode which must make
over 100Vrms to drive a tube with
33% CFB and it sounds just fine while measuring just fine.

The KT88 in fig 5 are working with relative electrode voltages
identical to a conventional UL amp with normal
25% UL taps. The Fig 5 circuit has 25% of the Va-k fed back to the
cathode. This cathode signal is in series
with the input grid signal. The amplitude of the cathode feedback
is sufficient to be a very effective amount of
local NFB which reduces the THD by a factor of more than 1/4, and
reduces output resistance of the stage
to about 1/2 that of triodes. But the maximum drive voltage needed
by the output stage is still at a moderate
level of 72Vrms and very easily produced by an EL84 strapped in
triode mode.

The L2 5H and C20 470uF form an LC filter with a pole at 3.3Hz,
but I believe this F is so low and that
cathode output resistance is so low that resonant effects of the
C&L are suppressed, ie, damped, and LF
stability should not be threatened. L2 choke should be at least
5H, and with Rw about 50r max. However,
if Rw was 100r, then the top of the choke would be at +26.6Vdc and
the cathodes would be at +60.3Vdc,
so the B+ supply would need to be raised from +410Vdc to +424Vdc.
Amps like this should always be built
with a few taps 15Vrms apart at the end of the winding to allow B+
to be varied depending on winding
resistances and to accommodate changes to the total load value
seen by tubes to best suit an OPT.

L2 choke shunts the cathodes to 0V with its increasing low
reactance as frequency becomes lower. This may
seem to reduce the NFB and increase KT88. The effect is also
increased by C20 whose reactance increases
as F gets lower. But at very low F of say 3Hz, the OPT primary
reactance shunts the tubes to an extent that
whatever weird behaviour occurs with C20 and L2, it is of no
importance.

The C20 should be at least 470uF rated for 450Vwkg, or a pair
paralleled. It should be bypassed with 2uF
to shunt the HF impedance of the electrolytic cap. The choke will
saturate at some low F below 14Hz.
It needs to be designed to take up to 100Vrms with 265mAdc present
without core saturation above 14Hz.
many "off the shelf chokes" may not fulfill the condition, so
perhaps a pair of smaller 2.5H chokes in series
would be better.

At 20Hz, the L2 5H has XL = 630r, and C20 XC = 17r. The inductance
of the whole OPT primary should
be no less than 10H. The Lp acts to shunt cathodes to anodes
and reduce the Va-k. as F goes lower.
The CFB tries to maintain the the Va to k with NFB action, but the
tubes saturate at 20Hz at full Po so
output voltage is limited and the L2 and C20 have little effect on
overload behaviour. This SE amp,
like all others, should have an C&R HPF at input plus LF gain
shelving R&C network after V1 to reduce
the open loop gain to very low levels at below 5Hz.

The use of a choke in Fig 5 is in effect a "cathode current choke
sink" of DC from KT88 cathodes to 0V.
It is a similar technique to using the well known ( but seldom
used ) choke feed from B+ to the anodes with
cap coupling of anodes to OPT. It is also know as "parafeed". The
choke is feeding Idc to anodes and the
choke inductance is in parallel with capacitor coupled primary
inductance. In Fig 5, the L2 5H is in parallel
with the 25% of turns of the OPT, and inductance of 25% of turns
will be about 1.3H, so the extra L2
inductance has negligible extra loading effects at very low F.

The choke L2 is able to do some of the function of an air gapped
OPT and therefore get more audio
power from the OPT. It also would be possible to have a CCS (
constant current sink ) using solid state
and all similar to what I have in Fig 6 below. But the SS devices
must be arranged very carefully because
the + / - cathode V swing means the cathodes go to a negative peak
voltage well below 0V so there
must be a suitably designed negative voltage rail.

If the 36W CFB amp in Fig 5 seems like too much trouble, or you
cannot source the esoteric rarely ever
available well wound OPT, then maybe something else will sound
well.....
Fig 6.
Fig 6 shows a "nearly conventional" SEUL amp and a Hammond 1640SEA
with 40% UL tap.
The specifications for the Hammond 1640 tranny has P : S ratio
1,250r : 4r, 8r, 16r.
The secondary is one winding for 16r and 8r is a tap at 70% of
turns and the 4r0 is at 50%, ie, the
secondary center tap. The specified maximum Idc is 200mAdc. This
means that peak current change in
class A = +/- 200mA peak. So maximum Po can be calculated = 0.5 (
0.2 x 0.2 x 1,250 ) = 25W.
This is a little bit less than what Hammond say, as it is a 30W
rated tranny. But we can forgive them this
minor discrepancy. With 25W, primary signal voltage = 176.7Vrms,
and there is 20Vrms across 16r
secondary.

Now 20Vrms is a large enough voltage to be usable for cathode
feedback if we wished because it is
11.36% of primary signal voltage. Anything over 10% is useful. If
we had just one KT88 producing
8.3W with OPT for 3,750r : 16r, the primary voltage would be the
same 176Vrms, but 16r sec would
have 1.5Vrms, only 6.5% of primary voltage which is an ineffective
amount of CFB. The normal UL
tap and GNFB loop would suffice.

What would happen if we used the 1640 secondary as a CFB winding
AND speaker winding?
The P:S turn ratio will change so that the transformer P:S ratio =
1,549r : 16r, because now the signal
current of tubes flows in secondary turns. This current is much
smaller than the speaker current so the
thicker sec turns can cope easily.

With primary load at 1,549r, it suits the use of 3 x KT88 with
each tube seeing 4,647r, a very nice load
for KT88 operating very comfortably with Ea at +310V and Ia at
60mA. EL34 would also work very
well. Pda + Pg2 per tube = 18.6W + 1.55W = 20.15W. The Idc total
for 3 KT88 is 195mAdc,
nearly Hammond's maximum allowed.

If the 195mAdc flows in 16r winding which may have Rw = 0.8r, the
Vdc across winding is about
0.16Vdc, and be applied to a speaker, unless a DC blocking cap is
used, which we will not to
because there is insufficient Vdc to polarize a large value
electrolytic cap. In addition, the Idc in sec
raises the Bdc of the core which will saturate at a higher
frequency. So there's two good reasons
NOT to have the tube Idc flowing in the sec. But the tube signal
current "helps" things to happen.

Individual cathode biasing R&C networks are good for
paralleled tubes always operating in class A.
But we could have constant current sinks instead of resistors. The
Fig 6 shows IRF610 used for
each output tube cathode. But other CCS with Darlington pair
connected bjts could be used because
the base input resistance is very high and voltage across R27,
R31, R36 remain constant.
The cathode "bypass" caps are still required for each cathode, but
instead of being taken to 0V they
are taken to the top of the 16r secondary winding. The three CCS
each have effective collector
resistance > 50k so having 3 in parallel makes a very high
cathode load which we may consider has
negligible effect in any considerations. The three CCS act like a
choke in Fig 5.

Notice that the 3 KT88 cathodes will settle at about +35Vdc. Just
exactly what Ek will be may vary
but I expect =35V, but samples may vary. Now the cathodes have
20Vrms at clipping, so the
V-swing is +/- 28V peak. The mosfet drain connections will also
have +/- 28V peak, and the
minimum voltage across the mosfets should not become less than
about 10V. The Vg-s should
never go negative. I have the mosfets set up with gates at -14Vdc,
and IRF610 data says gate bias
will be -4Vdc approx at low Id. So sources should be at -10V,
which is -45V below Ek, enabling
+/- 28Vpk cathode swing. I've chosen to have the required negative
rail for mosfets at -24Vdc.
Its not too hard for anyone to make an unregulated -24Vdc rail and
for 200mA. I should not
have to spoon feed you such a detail, and I just won't.

The Fig 6 has a large total amount of applied NFB in 4 "loops".
Each can be considered, and
perhaps discarded if deemed unnecessary.

Loop 1.
It is the use of the UL tap from primary winding to supply KT88
screens with what is normal
Ultra Linear screen FB used now since about 1955. The effect of
this connection makes the effective
Ra of each KT88 = approximately 3k0, a huge reduction from the
pure beam tetrode Ra of 24k.
Odd number H are reduced and spectra is brought closer to triode
operation.
Loop 2.

Local Cathode Feedback from OPT is used to reduce the Ra to less
than KT88 triode value.
The CFB is an external loop involving linear working transformer
windings so that all even and odd
number H are reduced by the same amount. One could say nobody
needs to have any more NFB
in such an amp but the grid input to KT88 requires 38Vrms, so we
MUST use a driver tube because
a preamp or PC sound card cannot make a linear 38Vrms.

Loop 3.
Screen FB to EL34 is used from the OPT secondary. Instead of
connecting the EL34 to anode for
simple triode operation, it is bypassed with C12 to the CFB from
OPT secondary. One might ask
why, well, it can be done, so let us consider the results. The
EL34 has to make 38Vms at anode
and although triode strapping would work just fine, we could have
its screen fed with a signal
voltage that is less than its anode signal so the EL34 is working
in Ultra Linear mode. There is a
convenient signal available and of the correct phase which is the
OPT secondary signal of 20Vrms
that is applied to KT88 cathodes. So EL34 has 52% UL operation.
The load for EL34 is the R17
and L1 in series which becomes high impedance load for most of the
audio band. The EL34 will be
found to be very nearly as linear as triode strapped with such a
load. Gain g1 to anode will be
found to be twice that of triode. Now within the OPT secondary
signal there will be distortion
products generated by KT88 AND those produced by the EL34. All
these distortion products
are amplified x about 10 by the gain between g2 and anode to
create an "error" signal that when
applied to the KT88 grids they are then amplified to oppose their
own production. The screen
NFB is effectively about 20dB NFB, although not a most perfect
form of FB, because the EL34
cannot ever provide less THD than when triode strapped and with
the high RLa load value I have
used. But a typical EL34 in triode mode with RLa > 20Ra can
make 100Vrms at 1% THD, and at
38Vrms perhaps 0.3%, mostly 2H and with 52% UL operation it may be
0.6%. But the KT88
may produce 2% THD, much more than the THD of EL34. The effect of
this screen FB will reduce
THD from 2% to about 0.4% at least, a huge reduction.

OK, so now the amp makes 25Watts at 0.4%, and we need only 2.2
Vrms input to the EL34 grid
so anyone may try all this without adding yet another input tube
V1. At normal listening levels of 1W,
THD can be expected to be 0.07% which is an excellent measured
result for most SE amps, and I
suggest ppl try it out.

What other SE amp amplifier has just TWO active devices with good
linearity?
You say there are 4 tubes total, but the three KT88 act as one
because they are paralleled, and they
could be replaced by 13E1, or 4 x EL34, and all I'm doing is
exploiting the screen properties of an
EL34 so it can be both input and driver. This is possible because
the ratio of g1 gm : g2 gm is quite
low. Put another way, the g2 gm is a useful high value which can
actually do a lot. Suppose we used a
6BX6 / EF80 instead. The g1 gm could be 5mA/V, quite useful, but
g2 gm is only 0.083mA/V, and the
fed back THD content is not amplified many times. The EL34
performs far better, even though it is a
power tube. Other suitable tubes for screen FB applications are
6CA7, EL84, EL86, possibly
EL36 / 6CM5.

Loop 4.
Global NFB from OPT sec to V1 cathode. V1 is a 12AU7 low µ twin
triode with µ = 17.
Gain with CCS anode supply via MJE350 is about 15x. The sound of
this tube is usually just worth
dying for. Here is has little to do, but the amount of NFB = 14dB,
so the 0.4% I spoke of above is
reduced by about 1/5 to 0.08%, say 0.1% at 25W, and at 1W THD may
be 0.014%. Such figures
are typical of very well made pure class A1 PP triode with 2 x
KT88 in triode and with 20dB global
NFB.

However, I have to say, "GEE, what a huge amount of NFB!"
Is it all really needed? Could it ever really be used? Well, in
fact, possibly because I have NOT BUILT
THIS AMP, but I can see already that there could be oscillations
at both LF and HF just outside the
audio band. The fact is that the Hammond 1640SEA does not have
extremely low leakage inductance.
Its barely low enough for general conventional use let alone for
the "sophisticated" schematic I have
proposed here. But if anyone made an OPT with twice the amount of
interleaving used by Hammond,
they may surprised by what might be done. In general, I have found
the 16xx series SE Hammond OPTs
to be very useable and good sounding.

In any amp, as the total amount of open loop gain without any FB
increased, any application of NFB
tries to extend over a wider range of F thus extending bandwidth
beyond the open loop bandwidth.
For example, if OL BW is from 30Hz to 20kHz, 10dB applied GNFB
might extend BW from 10Hz
to 40kHz. 30dB GNFB may increase BW to be 2Hz to 150kHz. But the
phase shift of the open loop
amp at LF and HF will cause the applied FB to become positive and
hence cause oscillations unless
gain shelving networks are used to reduce the phase shift and
reduce the open loop gain outside the
audio band. There is no need at all to have a high amount of NFB
applied outside the audio band.
In the Fig 6 amp, probable open loop F1 pole = 40Hz and F2 pole =
5kHz to get LF and HF stability.
This means that the high amount of NFB only applies to the band of
40Hz to 5kHz. However, perhaps
music is better and amp more stable to have less open loop gain
and less total NFB, with slightly more
THD and slightly lower damping factor and open loop F1 and F2
further apart at 15hz to 20kHz,
resulting in final bandwidth of 7Hz to 65kHz.

There is a simple answer, just leave out the screen NFB and use
the EL34 strapped as a triode.
EL34 triode grid signal will be 4.5Vrms. 12AU7 can have a higher
FB signal applied to its cathode
= 1.5Vrms. Total Va to Vk = 6Vrms, so Vg-k = 0.4Vrms, so input
signal to 12AU7 grids
= 1.9Vrms. This seems high, but is OK, and the amount of GNFB =
13.5dB, and the amp's THD
at 25W = 0.4%, quite good.

Now take note that in all my amps I have supplied to customers, I
have provided protection circuits
to prevent damage to OPTs and other parts if one or more output
tubes decides to conduct far too
much Idc because for one reason or another, the grid bias voltage
ceases to control the Ia flow.
Hence the note on the drawing about tp1, tp2, tp3. A KT88 which
becomes a short circuit could
damage the cathode CCS mosfet. I show no cathode fuse because
before it would blow, the mosfet
may fail with excessive Vd-s and enough current to fuse the mosfet
innards. There is a zener diode
in the mosfet, but it could fry easily.

OK, now you have seen the "traitors way to use Squalid State
devices" in a tube amp. How could I ?
Didn't I know they don't belong ? Well, OK, I hear the complaints,
but constant current sources or
current sinks using SS ARE OK, because they are so good at
providing an extremely high impedance
source of current and as such cannot have any effect on the
signal. The SS devices are friendly slaves,
totally under control of tubes, and they allow tubes to work
better than they other wise might.

3 x KT88 may be used with Hammond 1640SEA but with a single choke
in cathode circuit........
Fig 7.
Fig 7 is the same as Fig 6 but has SS current sinks replaced with
a choke. Ah, so simple! - until
you start thinking about a choke. You might find one for sale from
Hammond.

If you build this type of circuit using an
available-off-the-shelf-stock OPT then all I said above
about how it works applies. The Fig 7 amp like those above have a
high total amount of NFB.
But to prevent oscillations with a high amount of NFB you need to
be able to Nyquist and Bode
graphs and theory intuitively. Its too difficult for me to write a
book about it right now and have
you read 2 pages to understand, So, do your own Googling with Bode
and Nyquist, and lose a
week of your life trying to understand WTF it means.
Basically, as one increases the amount of NFB and or the amount of
open loop, gain, ie, the voltage
gain from input to output without any NFB applied, then the amp
becomes more prone to oscillations
due to the phase shift caused by reactive circuit elements L and C
reacting with R. For most tube
amps there is ONE F where there is no phase shift and it usually
is between 200Hz and 2kHz.
But below 15Hz and above 15kHz, and where the open loop gain has
become attenuated by R+C
Miller effects, C+R couplings, OPT primary L, leakage inductances,
shunt C, etc, and where phase
shift reaches 180 degrees and where gain exceeds 1.0, or "unity",
then the amp will oscillate.
The only way to prevent this is to use Zobel networks at output
and perhaps in output anode circuits
and at V1 output.

In Fig 7, the R+C values affecting stability ARE A GUIDE ONLY, and
YOU have to figure out the
best values for unconditional stability. The critical R+C parts
are R12+C8, R16+C9, C6+R13,
C19+R35, C21+R37.

If you find you just cannot stop oscillations, then abolish the
screen FB to EL34. Connect bottom
of C12 to 0V instead of to the FB from OPT. Reduce value of R16 to
1k0 to increase GNFB.
But you will still have to optimize the bandwidth with a pure R
load and get a good square wave
without severe ringing or HF oscillations with any pure C load
between 4uF and 0.047uF.
------------------------------------------------------------------------------------------------------

Cathode FB with a custom wound OPT.

It is time to mention my favorite output stage configuration for
both SE and PP tube amps. Such stages
have 2 primary windings and the usual secondary. One of the
primary windings has up to 1/3 of the turns of
the other and is called a cathode feedback winding, or tertiary
winding.
Conventional local CFB in SE output stage may be used if you can
find an OPT which has a separate
cathode feedback winding but really it is just part of the whole
primary winding because the tube signal
current flows in both windings. But the CFB winding is usually at
an earthy Vdc potential, while anode
winding is at the B+ Vdc.
Fig 8. "Normal" CFB SE output stages.
Fig 8 has two examples of CFB in output stages.
Fig 8 Left side.
The total signal voltage across both primary windings = 184V + 46V
= 230Vrms. 20% of the total turns
are in the CFB winding and 80% in the anode winding. I show the
formula for working out the effective
UL % as UL% = 100% x ( V ULtap + Vk ) / ( Va + Vk ) where the Vac
are measured between
each of the screen, anode and cathode terminals to 0V.
In this case the V UL = 0V because there is no UL tap. But the
KT88 is still operating as though there
is a UL tap because 46Vac exists between screen and cathode. Pure
beam tetrode or pentode with CFB
is achieved by bypassing the screen 100uF to the cathode so there
is no signal voltage between screen and
cathode.Operation of pure beam or pentode with CFB merely reduces
the complex THD spectra of the
pure beam of pentode. The use of an effective UL tap by means of
bypassing the screen to 0V changes the
THD spectra towards triode with less odd H.
Usually, where the CFB % of total primary turns is close to 10%,
the effective Ra of most beam tetrodes
and pentodes to be about equal to the triode Ra. When the CFB % is
raised above 10%, the Ra can
become considerably lower than triode Ra. Optimum CFB % will be
about 20% but can be from 10% to
40%.

The higher the CFB%, the higher the drive Vac needed so the driver
should have low THD.

Usually, the driver is a trioded EL34 or EL84, but could be 45 or
3A3, although real triodes have very
low gain so a high gain input triode is needed. So in my view the
EL34 or EL84 are winners.
Inevitably, driver triodes produce some 2H. If the driver has to
make 75Vrms, expect 0.7% 2H, and
the output tube may make 1.5%. The 2H generated will cancel, and
2H total at output becomes
1.5% - 0.7% = 0.8%. However, the 2H of all such CFB use with KT88,
EL34 varies with load, and
there is an RLa value just above the RLa value for maximum
possible Po where 2H = 0%.
Below this RLa value the 2H cancels, above the RLa value it adds,
because the relative phase of the
2H of the output tube is the opposite phase ! It is more fully
explained at my pages on the SE35
amp.

Fig 8 Right side.
The operation of the right side has effective UL % = 40% because
there is a UL tap at 20% of total
primary turns and there is 20% CFB. With class A operation there
could be a UL tap at 40% of total
turns which would increase the effective UL total to 60%. Going
beyond this % is pointless because it
restricts the Po available because operation becomes too much like
a triode where the Ea negative swing
is restricted by grid current onset. But at over 40% effective UL,
the odd H are very much reduced
leaving triode like THD which is mainly 2H, and the phenomena of
having 0% 2H at a higher RLa
diminishes and you get good cancelling of 2H produced by the
driver tube.

For SE class A1 and with most beam tetrodes and pentodes the
operation with about 20% CFB and
UL seems to give extraordinarily good sound, so I have been told.
----------------------------------------------------------------------------------------------------
"Shunt feed", aka "parafeed" output stages.
Shunt feed output stages have seldom ever been used for hi-fi
because you need a large air gapped
choke with high Idc flow, high inductance and not likely to
saturate with a high anode signal voltage
at above 14Hz. This choke provides a high impedance source of Idc
to the tube while not consuming
any audio power produced by the tube. The size and its weight may
be larger than the OPT used.

The audio signal power of the tube is conveyed to a primary
winding on an ungapped OPT via a
coupling cap. The OPT primary may be connected to 0V at one end,
and the cap has a large Vdc
across it. The OPT can be a normal PP OPT which is very easy to
source, and its Lp inductance
is usually far more than is actually needed, and it usually does
not saturate above 20Hz at full Po.
The coupling cap capacitance value must be high enough to have a
resonance with OPT LP at
below 3Hz. So if Lp = 30H at low signal levels the C should be
100uF. One might use a number
of C in series with resistance dividers to ensure equal Vdc is
across each cap and Vdc never exceeds
2/3 the Vdc rating for the cap. Electrolytic caps are needed, but
each must be bypassed with 1uF
plastic film caps.

The Shunt Feed advantages are :-
The OPT has no huge Vdc potential between primary and earthy
secondary or 0V or chassis.
The OPT may have no air gap. Laminations may be maximally
interleaved.
OPTs meant for PP amps may be used.
For the maximum Po, the OPT may be smaller than if the OPT was a
conventional air gapped OPT
with Idc flow.

The Shunt Feed disadvantages are :-
A large choke or 2 or 3 series chokes which may have the greater
size and weight as the OPT must
be used between anode and B+ to provide a high impedance feed of
Idc to anode.
Capacitor coupling from anode to OPT must be used which introduces
yet another time constant filter
behaviour which can affect the LF stability of the amp when NFB is
used. GNFB application with a
choke feed amp may be more difficult because unconditional
stability must be assured. The amount of
GNFB is limited by the number of C&R and C&L couplings and
L&R shunts. But with triode output
tubes 12dB GNFB with no output load is usually enough, and
possible, when LF oscillations are most
likely. More careful arrangement of open loop gain shelving
R&C networks are needed. But for HF
stability, there is no extra stability problem compared to using a
simple air gapped OPT and
conventional GNFB arrangement. The capacitors from anode to OPT
must be chosen carefully and
used with respect to their voltage ratings.

Fig 9.
Fig 9 shows how choke feed allows the use of a PP OPT without DC
flow through the primary
winding.
SEUL Choke feed, left.
I show the anode choke L1 = 55H at 80mAdc, and estimate Rw at
200r. You cannot purchase a
Hammond 55H choke equal to what I say is needed. Hammond have have
193C 20H with Rw
180r so 3 in series are needed for 60H. But then Rw total = 540r
giving 46Vdc and B+ must be
raised from +430V to +460Vdc. So before you copy what I show, be
sure you know ALL about
what you are doing!

The screen is cap coupled with 100uF to the CT of OPT primary
giving 50% UL. The screen
requires low Idc of 4mAdc choke feed through L1, perhaps a Hammond
155C, with Rw = 2k7,
L = 60H. I estimate 50H is plenty, and to get the Eg2 nearly equal
to Ea, and to prevent fusing
the L1 choke if screen shorts to cathode or 0V the series R = 1k0,
0.25W rated. If 300Vdc
appears across 1k0, it fuses quickly, and a new R costs 10c.

All electrolytic caps used should be rated for 350Vdc, and each
bypassed with 2uF, plastic film
types rated for 630V.

The C value for coupling anode to OPT seems quite high, 110uF in
fact. The LF pole formed by
110uF and anode load 4k2 is 0.34Hz, but more important is the pole
of HPF formed by the 110uF
and OPT primary L which may become 50H at low levels of signal
typically used for listening.
This F pole would be 2.14Hz, and the peak in response is prevented
by the very low impedance of
choke and OPT at 2.14Hz. Therefore enough global GNFB should be
able to be applied while
maintaining unconditional LF stability with R&C critical
damping networks in input - driver stages.

Triode Choke feed.
The triode use of KT88 can have B+ at a higher voltage for best
triode performance. A 300B could
also be used instead of the KT88, although the B+ has to be +40Vdc
higher because the Ek bias
will be about +88Vdc. The triode use does not need any screen
choke, but anode choke needs to
have high L value as for SEUL. All the same comments made about
SEUL apply about the L1 and caps.
Triode is easier because the screen is simply strapped to anode
via its stopper resistance of 220r.

845 Choke feed.
An 845 may happily work with an easy to buy Hammond 1650P OPT
rated for 60W, but used to
make about 21W. The load match is 6k6 : 4r,8r,16r.
Fig 10.
To make an air gapped OPT for the Fig 10 output stage with an 845
is extremely difficult for 99%
of DIYers. Making their own chokes would also be very difficult.
But they could purchase 1650P
OPT and three Hammond 193C chokes, and the required
capacitors. Possibly MUCH better
chokes and PP OPTs could be found than made by Hammond, but
appraisals of whatever brands
are used demands the full understanding of how the basic item
properties affect the performance.

I won't suggest how an air gapped OPT or choke may be designed
right now but the method and
examples can be found in my other OPT design pages.

I must mention the analysis behind Choke feed.
A choke is defined as a coil of enameled wire. Its basic
properties are described as an amount of
pure inductance in Henrys in series with resistance = winding wire
ohms. There is also capacitance
between turns resulting in a summed effect of an amount C shunting
the L. So all chokes have a
parallel resonance between the L and the C. For a single choke of
55H with an iron core, shunting
C might be only 300pF. The reactance XL of a 55H choke at 10Hz =
3,454r, and far less than the C.
The XL rises with F to a maximum of perhaps 500k ohms at the
theoretical resonance Fo at 1,240Hz.
Above Fo, the XL reduces because the XC declines with rising F and
= 3,454 ohms at 153kHz.
To extend the high XL the choke can be wound on a bobbin divided
into say 3 sections physically
2mm apart so that the 100pF shunt C of each is in series with the
other sections thus reducing C total
to < 50pF.
In practice, the C has little effect when the choke is in the
anode circuit of a tube output stage.
Chokes oppose the flow of AC because the voltage across the choke
sets up a magnetic field which
acts to oppose the flow. Therefore low current flows in the choke
across the audio band. The small
amount of audio power that is lost = Iac squared x Rw, and is
usually negligible. The same applies
to the inductance of any OPT where the power lost as heat in the
OPT = Iac squared x Rw.
The OPT and choke Rw losses are greatest at bass frequencies where
Iac becomes highest and
XL is lowest.
The shunt feed choke may be two or 3 chokes in series to make up
the wanted total choke inductance
as I have shown in Fig 10. The total L is simply the sum of the
individual L values. The L values don't
have to be equal, one could have a 40H choke plus 20H choke to
make 60H. But they MUST not
saturate with the presence of large signal voltages at LF and
combined with high Idc flows.

For Shunt feed SE amps, one aims to ensure core saturation of
choke or OPT does not occur at F higher
than 20Hz and with a signal voltage at the maximum Po level
for the RLa value which gives the highest
Po at the clipping level. For an 845, one might use RLa = 6k6, and
get at least 21W so Va = 372Vrms.
The 1650P OPT is rated for 60W to 6k6 and OK for 629Va-a and
saturation could be at 30Hz.
( I am not exactly sure. ) But saturation F is a voltage dependent
phenomena and occurs independently
to loading and currents. So at 372vrms, expect the Hammond OPT to
saturate at 18Hz. The maximum
primary inductance at high Va-a is probably more than 300H at say
1Tesla, typical with GOSS core
material with high µ. The choke has to cope with 372Vrms at 20Hz
without total Bac and Bdc summing
to more than say 1.2Tesla which is a maximum for medium grade
iron. Top grade GOSS lams or C-cores
may take only 1.5Tesla before onset of core saturation and poor
old iron from 1950 might take only
1.0Tesla. If you make the3 choke, assume the maximum iron µ is
below 2,000, and that it saturates at
1.0Tesla. So one may find one has to wind TWO chokes and connect
them in series. One may buy
chokes, but whatever is manufactured or bought MUST satisfy the
engineering design requirements,
and whatever is used must be tested carefully.

The total L shunting RLa is the choke L plus OPT Lp in parallel.
In all shunt feed SE amps, the air gapped
choke will have much less L than the cap coupled OPT. The total L
should have reactance = RLa at 20Hz
or lower F. Minimum Choke L value = RLa-a / ( 20 x 2pye ).
For RLa = 6,600r, L should be 6,600 / ( 20 x 6.28 ) = 52.5H.
During the amp's life it will be used at less
than 1W, so the Vac across the L will be less than 1/5 the
clipping level. The inductance of iron wound coils
varies with the applied voltage, ie, the lower the Bac, the lower
the permeability µ. Air gapped chokes and
air gapped OPTs have the least variation and reduction of L at low
signal levels. So we don't need to aim
to make the calculated choke L higher to compensate the drop in µ
at low Vac levels. Most variation of L
occurs with non gapped cores for PP amps where inductance at
0.05Watts, 18Vrms across 6k6 load,
may be 1/6 of the 20W Lp level, and so may be 50H. This L in in
parallel with choke L and if the choke
L = 50H then the total L may drop to below 20H at very low levels.
This WILL NOT cause any change
to the F response or bass performance because the tube Ra also
shunts the L and the RLa, and the 845 Ra
= 2k2 plus 6k6 RLa makes the R shunting L = 1k7, so F1 pole with
20H is at 13.5Hz. The 845
or any other tube will easily drive a low inductance load at very
low signal levels.
However, most people will wish to use NFB and the reduction of L
at low levels causes the F1 pole to rise
at low levels despite the Ra and RLa shunting the L and the
additional phase may cause LF oscillations.
Many old mass produced tube amps require a speaker to be connected
while the amp is turned on lest
they begin to oscillate at some low F below 10Hz. I have often
encountered such amps and the
oscillations can be violent enough to cause core saturation and
heavy damaging tube currents, or else
the amp oscillation is limited to a low level because inductance
rises with signal voltage so F1 moves
down and so does circuit gain so the amp in a state of equilibrium
as a phase shift oscillator.
Connecting a speaker puts more R across L and reduces F1 and the
amp stops oscillating.
Such amps have been designed by accountants. But now you see the
need for LF gain shelving R+C
networks to prevent the oscillations at LF.

Above I said total L minimum = 52.5H. If the OPT Lp was in fact
say 300H at 21W, then we should
prefer to ensure total XL = RLa-a at no higher than 20Hz.
Therefore the choke L should be 63.5H, so
when in parallel with 300H the total = 52.5H. Basically, we want
all the choke L we can muster!

Design of a single choke could be....
Ia = 83mAdc, so chose wire size so current density = 2A/sq.mm for
where Idc = 3 x Iadc.
For 249mAdc, wire size = 0.398mm Cu dia. Use high quality
polyester-imide enameled magnetic
winding wire, 0.4mm Cu dia, overall size including enamel = 0.47mm
dia. Winding window available
= 20mm x 72mm = 1,440 sq.mm so random wound turns possible = 1,440
/ ( 0.47 x 0.47 ) = 6,518
turns. Expect to get 6,000 turns on. Experience tells me that
probably this will give enough L at 83mAdc,
and with the correct air gap. But YOU need to read my pages on
choke design to verify that the choke
design is correct. The L is adjusted for a maximum by adjusting
the air gap size with 83mAdc present
plus a 50Vrms Va signal at 50Hz, without any load connected at
OPT. Perhaps you may find choke L
is higher than you wanted, which is great news, and bass will be
fabulous.

To make at least 2 chokes, one for each stereo channel, you would
need about 2Kg of new copper
winding wire, 2mm fiber-board for making bobbins, and enough
E&I laminations. These may be taken
from a large power transformer with a fused winding. I have often
used old laminations from defunct PTs
for chokes. To easily take the laminations apart, the transformer
should be placed in a small wood fire
for just long enough to make the lams appear dull red. The varnish
and all plastics will be vaporized
and burnt. The burnt out core is left to cool slowly and will
easily fall part when the bolts an wire are
cut off. The firing is a messy process best done late on winter
nights if you have a fireplace. The heat
may improve the magnetic properties, but won't worsen magnetic
properties.

There are other methods of avoiding the high +Vdc potential
between an OPT primary and the earthy
secondary. The first uses a "floating" B+ Vdc supply so that
say the earthy rail of the supply connects
to OPT primary at the normal B+ connection, while the other end
connects to 0V. The positive rail of
B+ supply connects to output tube anode/s.
This means the HT winding on PT and its diodes and filter caps and
chokes all are at an elevated B+
potential and all these items carry the anode signal. There is
some capacitance between all this hardware
and 0V and chassis. There is capacitance between secondary HT
winding and other PT windings such
as mains windings which may introduce noise to the anode audio
path.
Therefore the PT should have extremely good insulation between its
HT winding and all others especially
if the B+ supply was 1,200Vdc for 845 or 211 tubes. If the maximum
possible peak signal voltage swing
at any given instant was +1,100V, and the adjacent mains voltage
was -340Vpk, and the B+ was
+1200V, then one could have 2,640Vpk between windings. Hence the
need for good insulation.
To minimize HF diode switching noise signal getting into the
relatively high impedance anode circuit the
PT should have an an electrostatic shield. An ES is usually one
layer of wire turns with one end
connected to 0V and other end left open. Many old radio sets had
such a shield on their PT.
The amp heater windings can function as a shield if one whole
layer of thick wire is designed to
be devoted to heaters which have a CT connected to 0V.
So thus the PT can become a custom wound item.
Building a PSU that is alternative to the normal arrangement of
grounded B+ supply is more difficult
and expensive, and cannot improve sound quality.

Fig 11.
Fig 11 shows a KT88 operating in SEUL with a floating B+ supply,
but also has a choke feed.
It is yet another way to avoid having a special expensive OPT with
air gapped core.
I bet nobody else has ever used the schematic I have here.
Certainly I have not ever seen a mass
produced sample. Manufacturers always will try to avoid excessive
iron wound items and any extra
components that could be avoided by designing a good OPT with Idc
flow and air gap. Often their
attempts remind us that an accountant designed the amp, not an
engineer. Keen DIYers need not
heed any accountant's advice and they may be free to design
according to basic principles and be
creative in the process.

The audio is amplified in the same way as for any standard SEUL
amp. But the choke feed is
arranged differently to standard choke feed. The B+ supply is
floating, ie, not grounded anywhere,
and I show the PT with its HT secondary, diodes and 470uF caps and
filter choke all operating as a
+433Vdc battery between a KT88 anode and a choke at 0Vdc
potential. The Idc path is from
positive rail of B+ supply to anode then down to cathode, through
R&C cathode biasing network,
then to 0V rail then through L1 "choke feed". The B+ PSU for
+433Vdc has the anode signal present
at all parts, yet the floating PSU acts entirely without any
interference from the audio signals, and does
not impart any noise into the audio signal path. There is 0.85Vrms
of 100Hz ripple across reservoir
caps, and 1.1mV across the pair of caps for +433Vdc.

The feature of noiseless floating B+ PSU is made possible with the
use of an electrostatic shield on PT
between the B+ HT winding and other nearby windings with diode
switching noise.
Old amps and radios had a shield made with one layer of thin wire
between Mains input primary and HT
winding. One end of the layer was taken to the amp or radio
chassis, with other end left open.
But the tube heater winding of 6.3V can be made to occupy one
layer and have its CT taken to 0V
rail and it acts as an ES between mains and floating B+ HT
winding. But an ES should be wound over
this floating winding if additional B+ windings are which have
diode rectifiers. An ES need not be a
layer of wire, but can be one turn of thin copper or brass foil of
say 0.1mm thick and overlapped 10mm,
but prevented from being a shorted turn because insulation is
between the overlapped turn ends.
A wire lead is brought out to 0V. Therefore the only energy
transferred to the floating HT winding is
magnetic. I've shown a voltage doubler type of B+ supply with CLC
filtering and 1N5408 Si diodes,
quite good enough.

The tube operates with Ea = 375Vdc, and with +418Vdc at its anode
and with +43Vdc normal cathode
biasing and grid at 0Vdc potential. The anode signal is
transferred to the top of choke L1 55H which has
one end connected to 0V. Notice that the Idc through L1 produces a
NEGATIVE Vdc across the choke.
The higher the choke Rw, the greater this -Vdc will become, and
the higher the B+ supply voltage will
have to be to ensure the wanted Ea is produced. So low
resistance choke is wanted.

The L1 choke of 55H does not significantly load the KT88. 55H at
1kHz has XL = 345k plus some
shunt C of say 300pF = 530k. Now the proposed ideal RLa for
maximum Po = 4k2, so XL = RLa
at 12.15Hz, well below 20Hz, so at full 12.6Watt Vo level the KT88
does not put much signal current
through L1 above 20Hz. Fig 11 shows details of L1 and L3 chokes.
L1 could possibly be 3 x Hammond
193C 20H chokes in series but then Rw = 540r total and the L1 Vdc
= - 46Vdc, so B+ must be raised
from +433Vdc to +464Vdc, which means +496Vdc must be produced by
HT winding so the highest
voltage may not be high enough at 210Vrms. The 193C have maximum
Idc rating = 100mA, which is
a bit low, and the DIYer can make a MUCH better choke at home.

I hope everyone fully understands that if you change one thing
among several properties of one
component, it can seriously affect more than one other thing and
perhaps ruin the optimum
operation of the circuit. SE amps with low max Po ability need all
the optimization possible so that
a wide range of loads can be used.

The SEUL schematic has the anode signal appearing across L1, and
to get the wanted 12.6W of
audio power there must be an R load across L1, and it is the OPT
which is cap coupled to the L
choke with 1,000uF. The 1,000uF prevents any Idc flow in the OPT
primary which always remains
at 0Vdc potential. Now the top of choke L1 is at -15Vdc and
the 1,000uF cap may then be a low
voltage type rated for say 35Vdc and best are those with high
ripple current rating normally used in
PSU for SS amp circuits. The Vdc available will adequately
polarize the electrolytic to give linear
operation. A 2uF plastic film cap should be strapped across the
el-cap to shunt its ESR, and make
the cap work as a pure cap to 10MHz.

The OPT I recommend is Hammond PP type 1615A, rated 5k0 : 4r, 8r,
16r. The primary CT is
used for the UL screen connection. But the screen must have the
same +Vdc voltage as the anode,
so it is coupled to the OPT CT with 100uF el-cap, also bypassed
with say 1uF/630V plastic film.
To get the screen to be fed with Vdc = anode Vdc, I suggest the L2
choke, over 50H. It can be a
Hammond 155C, giving 60H, but its Rw is high at 2k7, and with
screen "stopper" resistor 1k0,
and Ig2 = 5mAdc, the screen Vdc is 18.5Vdc lower than anode Vdc.
Best SEUL operation occurs
with Eg2 = Ea when nearly all power beam tetrodes and pentodes
need Eg2 = Ea lest the maximum
Po be restricted similarly to triode operation. Ea to Eg2
difference should be limited to less than
5% of Ea. There is no doubt that a keen DIYer could make a
slightly larger choke than the
Hammond 155C and with MUCH lower Rw using thicker winding wire and
a bigger core.
Such a choke is less prone to fusing open if a screen becomes
shorted to cathode, or the coupling
el-cap to OPT becomes shorted. This L2 choke also has 1/2 the
anode signal across it, so it must
not saturate with excessive total Bac + Bdc.

Fig 12.
Fig 12 gives the choke feed arrangement for floating B+ and triode
operation. It is very similar to
Fig 10 for SEUL. The KT88 screen is connected to anode with 220r
stopper, so that the screen
is being fed ALL the anode signal. The screen g2 acts as an
additional control grid in the same way
an anode in a "real" triode acts to partially control Ia in
conjunction with the g1 control grid.
Triode operation gives the maximum possible amount of screen NFB
introduced into the tube,
so the voltage gain is lower than UL. The SE triode operation is
obviously simpler than SEUL,
and the maximum output power is not much less than SEUL. But
notice that to get somewhere
near the SEUL Po, the triode Ea and Ek must be higher with Ia
slightly lower for the Pda to be
30W, same as for the SEUL use.

If the KT88 strapped as a triode was used with exactly the same Ea
and Ek as shown for SEUL
in Fig 10, the Pda + Pg2 would still need to be kept at 30W. The
Ia+Ig2 = 85mAdc, so for
maximum Po the RLa = ( 375 / 0.085 ) - ( 2 x 1,100 ) = 2,211r.
This RLa is only 2 x Ra of the
triode. THD is higher than SEUL. Max Po at anode = 8W.
Damping factor = RLa / Ra = 2k2 / 1k1
= 2 which and no better than the SEUL with 4k2 load and 50% UL
tap. Then the OPT which was
chosen for UL has nominal ZR = 5k0 : 4, 8, 16, and for RLa = 2k2
triode anode loading, required
secondary loads = 1r8, 3r5, 7r1. But the winding losses will
double to become 20%, so therefore
the max Po with triode at the secondary becomes about 6.4W only.
Therefore I hope everyone
can see that to get the KT in triode to work nearly as well as
SEUL, you MUST use a higher Ea
and lower Ia to suit the load which is nearly 4 x Ra, and OK for
good triode operation.

Now we might consider the KT88 in SEUL mode with Ea +440V and Ia +
Ig2 69mA. The Ia will
be 64mA, and RLa for max Po = 0.9 x 440 / 0.064 = 6k2. Max Po at
anode = 12.7W.
The OPT secondary loads needed to give RLa = 6k2 are 5r0, 10r0,
20r0. A speaker of "8r"
could be used on the 8r outlet, but the primary loads becomes 5k0,
and Po max = 10W.
But an 8r0 speaker may dip to 5r0 so it is best used on the 4r
outlet and the tube will produce
highest power where the Z dip occurs which is OK. But the use of a
"4r" speaker with a dip to 3r0
will reduce RLa to 3k7, and Po max = 7.5W with high THD. However,
all this isn't as bad as using
the KT88 in triode with Ea = 375V.

SE amps can only give maximum Po at ONE load value. Suppose the
load for 12.7W max is 3r5,
when a "4r" speaker is used on the 4r outlet. Then if speaker Z
dips to 2r0, Po max about 6.7W,
and if Z rises to 8r0, Po max = 6.6W. So although a "12W" amp can
make 12W, the inevitable
variation in speaker Z across its frequency band and the low
sensitivity of modern speakers will
reduce real capability of the amp to 6W. Hence high sensitive
speakers should be used. However,
with say 3 or 4 x KT88, the usable Po range becomes say 24W, and
then the load matching becomes
far less critical and the amp will perform well with any speaker.