Fourier and Spectral Applications part 8

There are many variant procedures that all fall under the rubric of LPC. • If the spectral character of the data is time-variable, then it is best not to use a single set of LP coefﬁcients for the whole data set, but rather to partition the data into segments

Nội dung Text: Fourier and Spectral Applications part 8

572 Chapter 13. Fourier and Spectral Applications
There are many variant procedures that all fall under the rubric of LPC.
• If the spectral character of the data is time-variable, then it is best not
to use a single set of LP coefﬁcients for the whole data set, but rather
to partition the data into segments, computing and storing different LP
coefﬁcients for each segment.
• If the data are really well characterized by their LP coefﬁcients, and you
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can tolerate some small amount of error, then don’t bother storing all of the
residuals. Just do linear prediction until you are outside of tolerances, then
reinitialize (using M sequential stored residuals) and continue predicting.
• In some applications, most notably speech synthesis, one cares only about
the spectral content of the reconstructed signal, not the relative phases.
In this case, one need not store any starting values at all, but only the
LP coefﬁcients for each segment of the data. The output is reconstructed
by driving these coefﬁcients with initial conditions consisting of all zeros
except for one nonzero spike. A speech synthesizer chip may have of
order 10 LP coefﬁcients, which change perhaps 20 to 50 times per second.
• Some people believe that it is interesting to analyze a signal by LPC, even
when the residuals xi are not small. The xi ’s are then interpreted as the
underlying “input signal” which, when ﬁltered through the all-poles ﬁlter
deﬁned by the LP coefﬁcients (see §13.7), produces the observed “output
signal.” LPC reveals simultaneously, it is said, the nature of the ﬁlter and
the particular input that is driving it. We are skeptical of these applications;
the literature, however, is full of extravagant claims.
CITED REFERENCES AND FURTHER READING:
Childers, D.G. (ed.) 1978, Modern Spectrum Analysis (New York: IEEE Press), especially the
paper by J. Makhoul (reprinted from Proceedings of the IEEE, vol. 63, p. 561, 1975).
Burg, J.P. 1968, reprinted in Childers, 1978. [1]
Anderson, N. 1974, reprinted in Childers, 1978. [2]
Cressie, N. 1991, in Spatial Statistics and Digital Image Analysis (Washington: National Academy
Press). [3]
Press, W.H., and Rybicki, G.B. 1992, Astrophysical Journal, vol. 398, pp. 169–176. [4]
13.7 Power Spectrum Estimation by the
Maximum Entropy (All Poles) Method
The FFT is not the only way to estimate the power spectrum of a process, nor is it
necessarily the best way for all purposes. To see how one might devise another method,
let us enlarge our view for a moment, so that it includes not only real frequencies in the
Nyquist interval −fc < f < fc , but also the entire complex frequency plane. From that
vantage point, let us transform the complex f -plane to a new plane, called the z-transform
plane or z-plane, by the relation
z ≡ e2πif ∆ (13.7.1)
where ∆ is, as usual, the sampling interval in the time domain. Notice that the Nyquist interval
on the real axis of the f -plane maps one-to-one onto the unit circle in the complex z-plane.

13.7 Maximum Entropy (All Poles) Method 573
If we now compare (13.7.1) to equations (13.4.4) and (13.4.6), we see that the FFT
power spectrum estimate (13.4.5) for any real sampled function ck ≡ c(tk ) can be written,
except for normalization convention, as
2
N/2−1
k
P (f ) = ck z (13.7.2)
k=−N/2
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Of course, (13.7.2) is not the true power spectrum of the underlying function c(t), but only an
estimate. We can see in two related ways why the estimate is not likely to be exact. First, in the
time domain, the estimate is based on only a ﬁnite range of the function c(t) which may, for all
we know, have continued from t = −∞ to ∞. Second, in the z-plane of equation (13.7.2), the
ﬁnite Laurent series offers, in general, only an approximation to a general analytic function of
z. In fact, a formal expression for representing “true” power spectra (up to normalization) is
∞ 2
P (f ) = ck z k (13.7.3)
k=−∞
This is an inﬁnite Laurent series which depends on an inﬁnite number of values ck . Equation
(13.7.2) is just one kind of analytic approximation to the analytic function of z represented
by (13.7.3); the kind, in fact, that is implicit in the use of FFTs to estimate power spectra by
periodogram methods. It goes under several names, including direct method, all-zero model,
and moving average (MA) model. The term “all-zero” in particular refers to the fact that the
model spectrum can have zeros in the z-plane, but not poles.
If we look at the problem of approximating (13.7.3) more generally it seems clear that
we could do a better job with a rational function, one with a series of type (13.7.2) in both the
numerator and the denominator. Less obviously, it turns out that there are some advantages in
an approximation whose free parameters all lie in the denominator, namely,
1 a0
P (f ) ≈ 2 = 2 (13.7.4)
M/2 M
bk z k 1+ ak zk
k=−M/2 k=1
Here the second equality brings in a new set of coefﬁcients ak ’s, which can be determined
from the bk ’s using the fact that z lies on the unit circle. The bk ’s can be thought of as
being determined by the condition that power series expansion of (13.7.4) agree with the
ﬁrst M + 1 terms of (13.7.3). In practice, as we shall see, one determines the bk ’s or
ak ’s by another method.
The differences between the approximations (13.7.2) and (13.7.4) are not just cosmetic.
They are approximations with very different character. Most notable is the fact that (13.7.4)
can have poles, corresponding to inﬁnite power spectral density, on the unit z-circle, i.e., at
real frequencies in the Nyquist interval. Such poles can provide an accurate representation
for underlying power spectra that have sharp, discrete “lines” or delta-functions. By contrast,
(13.7.2) can have only zeros, not poles, at real frequencies in the Nyquist interval, and must
thus attempt to ﬁt sharp spectral features with, essentially, a polynomial. The approximation
(13.7.4) goes under several names: all-poles model, maximum entropy method (MEM),
autoregressive model (AR). We need only ﬁnd out how to compute the coefﬁcients a0 and the
ak ’s from a data set, so that we can actually use (13.7.4) to obtain spectral estimates.
A pleasant surprise is that we already know how! Look at equation (13.6.11) for linear
prediction. Compare it with linear ﬁlter equations (13.5.1) and (13.5.2), and you will see that,
viewed as a ﬁlter that takes input x’s into output y’s, linear prediction has a ﬁlter function
1
H(f ) = N
(13.7.5)
1− dj z j
j=1
Thus, the power spectrum of the y’s should be equal to the power spectrum of the x’s
multiplied by |H(f )|2. Now let us think about what the spectrum of the input x’s is, when

574 Chapter 13. Fourier and Spectral Applications
they are residual discrepancies from linear prediction. Although we will not prove it formally,
it is intuitively believable that the x’s are independently random and therefore have a ﬂat
(white noise) spectrum. (Roughly speaking, any residual correlations left in the x’s would
have allowed a more accurate linear prediction, and would have been removed.) The overall
normalization of this ﬂat spectrum is just the mean square amplitude of the x’s. But this is
exactly the quantity computed in equation (13.6.13) and returned by the routine memcof as
xms. Thus, the coefﬁcients a0 and ak in equation (13.7.4) are related to the LP coefﬁcients
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returned by memcof simply by
a0 = xms ak = −d(k), k = 1, . . . , M (13.7.6)
There is also another way to describe the relation between the ak ’s and the autocorrelation
components φk . The Wiener-Khinchin theorem (12.0.12) says that the Fourier transform of
the autocorrelation is equal to the power spectrum. In z-transform language, this Fourier
transform is just a Laurent series in z. The equation that is to be satisﬁed by the coefﬁcients
in equation (13.7.4) is thus
M
a0
2 ≈ φj z j (13.7.7)
M
j=−M
1+ ak zk
k=1
The approximately equal sign in (13.7.7) has a somewhat special interpretation. It means
that the series expansion of the left-hand side is supposed to agree with the right-hand side
term by term from z −M to z M . Outside this range of terms, the right-hand side is obviously
zero, while the left-hand side will still have nonzero terms. Notice that M , the number of
coefﬁcients in the approximation on the left-hand side, can be any integer up to N , the total
number of autocorrelations available. (In practice, one often chooses M much smaller than
N .) M is called the order or number of poles of the approximation.
Whatever the chosen value of M , the series expansion of the left-hand side of (13.7.7)
deﬁnes a certain sort of extrapolation of the autocorrelation function to lags larger than M , in
fact even to lags larger than N , i.e., larger than the run of data can actually measure. It turns
out that this particular extrapolation can be shown to have, among all possible extrapolations,
the maximum entropy in a deﬁnable information-theoretic sense. Hence the name maximum
entropy method, or MEM. The maximum entropy property has caused MEM to acquire a
certain “cult” popularity; one sometimes hears that it gives an intrinsically “better” estimate
than is given by other methods. Don’t believe it. MEM has the very cute property of
being able to ﬁt sharp spectral features, but there is nothing else magical about its power
spectrum estimates.
The operations count in memcof scales as the product of N (the number of data points)
and M (the desired order of the MEM approximation). If M were chosen to be as large as
N , then the method would be much slower than the N log N FFT methods of the previous
section. In practice, however, one usually wants to limit the order (or number of poles) of the
MEM approximation to a few times the number of sharp spectral features that one desires it
to ﬁt. With this restricted number of poles, the method will smooth the spectrum somewhat,
but this is often a desirable property. While exact values depend on the application, one
might take M = 10 or 20 or 50 for N = 1000 or 10000. In that case MEM estimation is
not much slower than FFT estimation.
We feel obliged to warn you that memcof can be a bit quirky at times. If the number of
poles or number of data points is too large, roundoff error can be a problem, even in double
precision. With “peaky” data (i.e., data with extremely sharp spectral features), the algorithm
may suggest split peaks even at modest orders, and the peaks may shift with the phase of the
sine wave. Also, with noisy input functions, if you choose too high an order, you will ﬁnd
spurious peaks galore! Some experts recommend the use of this algorithm in conjunction with
more conservative methods, like periodograms, to help choose the correct model order, and to
avoid getting too fooled by spurious spectral features. MEM can be ﬁnicky, but it can also do
remarkable things. We recommend that you try it out, cautiously, on your own problems. We
now turn to the evaluation of the MEM spectral estimate from its coefﬁcients.
The MEM estimation (13.7.4) is a function of continuously varying frequency f . There
is no special signiﬁcance to speciﬁc equally spaced frequencies as there was in the FFT case.

13.8 Spectral Analysis of Unevenly Sampled Data 575
In fact, since the MEM estimate may have very sharp spectral features, one wants to be able to
evaluate it on a very ﬁne mesh near to those features, but perhaps only more coarsely farther
away from them. Here is a function which, given the coefﬁcients already computed, evaluates
(13.7.4) and returns the estimated power spectrum as a function of f ∆ (the frequency times
the sampling interval). Of course, f ∆ should lie in the Nyquist range between −1/2 and 1/2.
#include
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float evlmem(float fdt, float d[], int m, float xms)
Given d[1..m], m, xms as returned by memcof, this function returns the power spectrum
estimate P (f ) as a function of fdt = f ∆.
{
int i;
float sumr=1.0,sumi=0.0;
double wr=1.0,wi=0.0,wpr,wpi,wtemp,theta; Trig. recurrences in double precision.
theta=6.28318530717959*fdt;
wpr=cos(theta); Set up for recurrence relations.
wpi=sin(theta);
for (i=1;i