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Abstract:

A method for detecting a digital radio signal includes the steps of
receiving the digital radio signal including a series of symbols,
developing a correlation waveform having a peak that corresponds to a
symbol boundary, normalizing the correlation waveform, and calculating a
peak value of the normalized correlation waveform, wherein the peak value
represents the quality of the received digital radio signal. A receiver
that performs the method is also provided.

Claims:

1. A method for detecting the quality of a digital radio signal, the
method comprising the steps of:receiving a digital radio signal including
a series of symbols;developing a correlation waveform having a peak that
corresponds to a symbol boundary;normalizing the correlation waveform;
andcalculating a peak value of the normalized correlation waveform,
wherein the peak value represents the quality of the received digital
radio signal.

2. The method of claim 1, wherein the step of developing a correlation
waveform is performed for upper and lower sidebands of the digital radio
signal to produce an upper sideband correlation waveform and a lower
sideband correlation waveform.

3. The method of claim 2, wherein the step of normalizing the correlation
waveform is performed for the upper and lower sideband correlation
waveforms.

4. The method of claim 3, wherein the step of calculating the peak value
of the normalized correlation waveform is performed for the normalized
upper and lower sideband correlation waveforms.

5. The method of claim 4, further comprising the steps of:comparing the
peak values of the normalized upper and lower sideband correlation
waveforms to a first predetermined threshold, andcomparing the sum of the
peak values of the normalized upper and lower sideband correlation
waveforms to a second predetermined threshold.

6. The method of claim 4, further comprising the steps of:determining a
peak index of the normalized upper sideband correlation waveform and a
peak index of the normalized lower sideband correlation waveform;
andcalculating a peak index delta representative of the difference
between the peak indices for the normalized upper and lower sideband
correlation waveforms.

7. The method of claim 6, further comprising the steps of:comparing the
sum of the peak values of the normalized upper and lower sideband
correlation waveforms to a first predetermined threshold; andcomparing
the peak index delta to a second predetermined threshold.

8. The method of claim 1, further comprising the step of:comparing the
peak value of the normalized correlation waveform to a predetermined
threshold.

9. The method of claim 1, further comprising the step of:setting a status
flag to indicate if the received digital radio signal exceeds a
predetermined quality threshold.

10. The method of claim 1, wherein the digital radio signal includes upper
and lower sidebands, and the symbols received on the upper and lower
sidebands are processed separately.

11. The method of claim 10, further comprising the step of:filtering each
sideband in the digital radio signal prior to the step of developing a
correlation waveform.

12. The method of claim 11, wherein the filtering step is performed using
a finite impulse response filter.

13. The method of claim 1, wherein the correlation waveform is based on
amplitudes of samples of leading and trailing portions of orthogonal
frequency division multiplexed symbols.

14. The method of claim 13, wherein the amplitudes of the leading and
trailing portions of the orthogonal frequency division multiplexed
symbols are tapered.

15. The method of claim 1, wherein the symbols comprise orthogonal
frequency division multiplexed symbols and the correlation waveform is
based on a cyclic prefix applied to the symbols.

16. The method of claim 2, further comprising the step of calculating a
frequency offset difference for the correlation waveforms for the upper
and lower sidebands.

17. The method of claim 16, further comprising the step of comparing the
frequency offset difference to a predetermined threshold.

18. A receiver for detecting a digital radio signal, the receiver
comprising:an input for receiving a digital radio signal including a
series of symbols; anda processor for calculating a peak value that
corresponds to a symbol boundary of a normalized correlation waveform,
wherein the peak value represents the quality of the received digital
radio signal.

19. The receiver of claim 18, wherein the digital radio signal includes
upper and lower sidebands, and the processor calculates the peak values
of a normalized upper sideband correlation waveform and a normalized
lower sideband correlation waveform.

20. The receiver of claim 19, wherein the processor compares at least one
of the peak values of the normalized upper and lower sideband correlation
waveforms to a predetermined threshold.

21. The receiver of claim 19, wherein the processor compares the peak
values of the normalized upper and lower sideband correlation waveforms
to a first predetermined threshold and compares the sum of the peak
values of the normalized upper and lower sideband correlation waveforms
to a second predetermined threshold.

22. The receiver of claim 19, wherein the processor determines a peak
index for the normalized upper sideband correlation waveform and a peak
index of the normalized lower sideband correlation waveform, and a peak
index delta representative of the difference between the peak indices for
the normalized upper and lower sideband correlation waveforms.

23. The receiver of claim 22, wherein the processor compares the sum of
the peak values of the normalized upper and lower sideband correlation
waveforms to a first predetermined threshold and the peak index delta to
a second predetermined threshold.

24. The receiver of claim 18, wherein the processor sets a status flag to
indicate that the received digital radio signal exceeds a predetermined
quality threshold.

25. The receiver of claim 18, wherein the digital radio signal includes
upper and lower sidebands, and the samples received on the upper and
lower sidebands are processed separately.

26. The receiver of claim 25, further comprising:a filter for filtering
each sideband in the digital radio signal prior to the processor
calculating the peak value of a normalized correlation waveform.

28. The receiver of claim 18, wherein the correlation waveform is based on
amplitudes of samples of leading and trailing portions of orthogonal
frequency division multiplexed symbols.

29. The receiver of claim 28, wherein the amplitudes of the leading and
trailing portions of the symbols are tapered.

30. The receiver of claim 18, wherein the symbols comprise orthogonal
frequency division multiplexed symbols and the correlation waveform is
based on a cyclic prefix applied to the symbols.

31. The receiver of claim 19, wherein the processor calculates a frequency
offset difference for the correlation waveforms for the upper and lower
sidebands.

32. The method of claim 31, wherein the processor compares the frequency
offset difference to a predetermined threshold.

Description:

FIELD OF THE INVENTION

[0001]This invention relates to digital radio broadcasting receivers, and
more particularly to methods and apparatus for implementing in a digital
radio receiver a signal quality metric for an OFDM digital signal.

BACKGROUND OF THE INVENTION

[0002]Digital radio broadcasting technology delivers digital audio and
data services to mobile, portable, and fixed receivers. One type of
digital radio broadcasting, referred to as in-band on-channel (IBOC)
digital audio broadcasting (DAB), uses terrestrial transmitters in the
existing Medium Frequency (MF) and Very High Frequency (VHF) radio bands.
HD Radio® technology, developed by iBiquity Digital Corporation, is
one example of an IBOC implementation for digital radio broadcasting and
reception.

[0003]IBOC DAB signals can be transmitted in a hybrid format including an
analog modulated carrier in combination with a plurality of digitally
modulated carriers or in an all-digital format wherein the analog
modulated carrier is not used. Using the hybrid mode, broadcasters may
continue to transmit analog AM and FM simultaneously with higher-quality
and more robust digital signals, allowing themselves and their listeners
to convert from analog-to-digital radio while maintaining their current
frequency allocations.

[0004]One feature of digital transmission systems is the inherent ability
to simultaneously transmit both digitized audio and data. Thus the
technology also allows for wireless data services from AM and FM radio
stations. The broadcast signals can include metadata, such as the artist,
song title, or station call letters. Special messages about events,
traffic, and weather can also be included. For example, traffic
information, weather forecasts, news, and sports scores can all be
scrolled across a radio receiver's display while the user listens to a
radio station.

[0005]IBOC DAB technology can provide digital quality audio, superior to
existing analog broadcasting formats. Because each IBOC DAB signal is
transmitted within the spectral mask of an existing AM or FM channel
allocation, it requires no new spectral allocations. IBOC DAB promotes
economy of spectrum while enabling broadcasters to supply digital quality
audio to the present base of listeners.

[0006]Multicasting, the ability to deliver several programs or data
streams over one channel in the AM or FM spectrum, enables stations to
broadcast multiple streams of data on separate supplemental or
sub-channels of the main frequency. For example, multiple streams of data
can include alternative music formats, local traffic, weather, news, and
sports. The supplemental channels can be accessed in the same manner as
the traditional station frequency using tuning or seeking functions. For
example, if the analog modulated signal is centered at 94.1 MHz, the same
broadcast in IBOC DAB can include supplemental channels 94.1-1, 94.1-2,
and 94.1-3. Highly specialized programming on supplemental channels can
be delivered to tightly targeted audiences, creating more opportunities
for advertisers to integrate their brand with program content. As used
herein, multicasting includes the transmission of one or more programs in
a single digital radio broadcasting channel or on a single digital radio
broadcasting signal. Multicast content can include a main program service
(MPS), supplemental program services (SPS), program service data (PSD),
and/or other broadcast data.

[0007]The National Radio Systems Committee, a standard-setting
organization sponsored by the National Association of Broadcasters and
the Consumer Electronics Association, adopted an IBOC standard,
designated NRSC-5A, in September 2005. NRSC-5A, the disclosure of which
is incorporated herein by reference, sets forth the requirements for
broadcasting digital audio and ancillary data over AM and FM broadcast
channels. The standard and its reference documents contain detailed
explanations of the RF/transmission subsystem and the transport and
service multiplex subsystems. Copies of the standard can be obtained from
the NRSC at http://www.nrscstandards.org/standards.asp. iBiquity's HD
Radio® technology is an implementation of the NRSC-5A IBOC standard.
Further information regarding HD Radio® technology can be found at
www.hdradio.com and www.ibiquity.com.

[0008]Other types of digital radio broadcasting systems include satellite
systems such as XM Radio, Sirius and WorldSpace, and terrestrial systems
such as Digital Radio Mondiale (DRM), Eureka 147 (branded as DAB), DAB
Version 2, and FMeXtra. As used herein, the phrase "digital radio
broadcasting" encompasses digital audio broadcasting including in-band
on-channel broadcasting, as well as other digital terrestrial
broadcasting and satellite broadcasting.

[0009]It would be desirable to have a metric for the quality of a received
digital signal because a number of applications require an accurate
indication of signal quality, including a seek-scan function, resolution
of 300-kHz-spaced interferers, first adjacent interferer sideband
selection, and diversity switching, for example. It would also be
desirable for this metric to be quickly obtained, and to be effective and
reliable for FM hybrid and all-digital signals. It would also be
desirable to minimize any changes to existing HD Radio® receiver
hardware or software when implementing the metric calculation.

SUMMARY OF THE INVENTION

[0010]In a first aspect, the invention provides a method for detecting a
digital radio signal. The digital radio signal includes a series of
symbols, each of which is comprised of a plurality of samples. The method
includes the steps of receiving the digital radio signal, developing a
correlation waveform having a peak that corresponds to a symbol boundary,
normalizing the correlation waveform, and calculating a peak value of the
normalized correlation waveform, wherein the peak value represents the
quality of the received digital radio signal.

[0011]The digital radio signal can comprise upper and lower sidebands, and
the method can be applied independently to each of the sidebands to
produce the peak values of normalized correlation waveforms for each of
the sidebands. The digital signal quality metric can be validated by
calculating a peak index delta. The method can include calculating the
peak index corresponding to the peak value for the normalized correlation
waveforms for the upper and lower sidebands. Then a peak index delta
representative of the difference between the peak indices for the upper
and lower sidebands can be determined and the peak index delta and the
peak values for the upper and lower sidebands can be compared to
thresholds. The digital signal quality metric can also be validated by
calculating a frequency offset difference between the upper and lower
sidebands and determining whether the difference meets a certain
threshold, thereby indicating whether a detected signal is a desired
signal of interest or an adjacent interfering signal.

[0012]In another aspect, the invention provides a receiver for detecting a
digital radio signal. The digital radio signal includes a series of
symbols, each of which is comprised of a plurality of samples. The
receiver includes an input for receiving a digital radio signal, and a
processor for calculating a peak value that corresponds to a symbol
boundary of a normalized correlation waveform, wherein the peak value
represents the quality of the received digital radio signal.

BRIEF DESCRIPTION OF THE DRAWINGS

[0013]FIG. 1 is a block diagram of a transmitter for use in an in-band
on-channel digital radio broadcasting system.

[0014]FIG. 2 is a schematic representation of a hybrid FM IBOC waveform.

[0015]FIG. 3 is a schematic representation of an extended hybrid FM IBOC
waveform.

[0016]FIG. 4 is a schematic representation of an all-digital FM IBOC
waveform.

[0017]FIG. 5 is a schematic representation of a hybrid AM IBOC DAB
waveform.

[0018]FIG. 6 is a schematic representation of an all-digital AM IBOC DAB
waveform.

[0019]FIG. 7 is a functional block diagram of an AM IBOC DAB receiver.

[0020]FIG. 8 is a functional block diagram of an FM IBOC DAB receiver.

[0021]FIGS. 9a and 9b are diagrams of an IBOC DAB logical protocol stack
from the broadcast perspective.

[0022]FIG. 10 is a diagram of an IBOC DAB logical protocol stack from the
receiver perspective.

[0023]FIG. 11a is a graphical representation of an OFDM signal in the
frequency domain.

[0024]FIG. 11b is a graphical representation of the OFDM signal in the
time domain.

[0026]FIG. 11d is a graphical illustration of the conjugate products
multiplied by respective amplitude tapers.

[0027]FIG. 12 is a block diagram of one embodiment of an acquisition
module.

[0028]FIGS. 13a, 13b, and 13c are graphical representations of symbol
timing for a peak development module.

[0029]FIG. 14 is a flow diagram of a first portion of signal acquisition
processing.

[0030]FIG. 15 is a functional block diagram that illustrates an
acquisition algorithm.

[0031]FIG. 16 is a functional block diagram of sideband combination.

[0032]FIG. 17 is a diagram that illustrates waveform normalization near a
symbol boundary.

[0033]FIG. 18 is a graph of a normalized correlation peak.

[0034]FIG. 19 is a flow diagram of a second portion of signal acquisition
processing.

[0035]FIGS. 20 through 24 are graphs of the probability of stopping at a
particular frequency for various conditions in a seek-scan application of
a digital signal quality metric according to the present invention.

DETAILED DESCRIPTION OF THE INVENTION

[0036]FIGS. 1-13 and the accompanying description herein provide a general
description of an IBOC system, including broadcasting equipment structure
and operation, receiver structure and operation, and the structure of
IBOC DAB waveforms. FIGS. 14-24 and the accompanying description herein
provide a detailed description of the structure and operation of an
acquisition module for implementing a digital signal quality metric
according to an aspect of the present invention.

IBOC System and Waveforms

[0037]Referring to the drawings, FIG. 1 is a functional block diagram of
the relevant components of a studio site 10, an FM transmitter site 12,
and a studio transmitter link (STL) 14 that can be used to broadcast an
FM IBOC DAB signal. The studio site includes, among other things, studio
automation equipment 34, an Ensemble Operations Center (EOC) 16 that
includes an importer 18, an exporter 20, an exciter auxiliary service
unit (EASU) 22, and an STL transmitter 48. The transmitter site includes
an STL receiver 54, a digital exciter 56 that includes an exciter engine
(exgine) subsystem 58, and an analog exciter 60. While in FIG. 1 the
exporter is resident at a radio station's studio site and the exciter is
located at the transmission site, these elements may be co-located at the
transmission site.

[0038]At the studio site, the studio automation equipment supplies main
program service (MPS) audio 42 to the EASU, MPS data 40 to the exporter,
supplemental program service (SPS) audio 38 to the importer, and SPS data
36 to the importer. MPS audio serves as the main audio programming
source. In hybrid modes, it preserves the existing analog radio
programming formats in both the analog and digital transmissions. MPS
data, also known as program service data (PSD), includes information such
as music title, artist, album name, etc. Supplemental program service can
include supplementary audio content as well as program associated data.

[0039]The importer contains hardware and software for supplying advanced
application services (AAS). A "service" is content that is delivered to
users via an IBOC DAB broadcast, and AAS can include any type of data
that is not classified as MPS, SPS, or Station Information Service (SIS).
SIS provides station information, such as call sign, absolute time,
position correlated to GPS, etc. Examples of AAS data include real-time
traffic and weather information, navigation map updates or other images,
electronic program guides, multimedia programming, other audio services,
and other content. The content for AAS can be supplied by service
providers 44, which provide service data 46 to the importer via an
application program interface (API). The service providers may be a
broadcaster located at the studio site or externally sourced third-party
providers of services and content. The importer can establish session
connections between multiple service providers. The importer encodes and
multiplexes service data 46, SPS audio 38, and SPS data 36 to produce
exporter link data 24, which is output to the exporter via a data link.

[0040]The exporter 20 contains the hardware and software necessary to
supply the main program service and SIS for broadcasting. The exporter
accepts digital MPS audio 26 over an audio interface and compresses the
audio. The exporter also multiplexes MPS data 40, exporter link data 24,
and the compressed digital MPS audio to produce exciter link data 52. In
addition, the exporter accepts analog MPS audio 28 over its audio
interface and applies a pre-programmed delay to it to produce a delayed
analog MPS audio signal 30. This analog audio can be broadcast as a
backup channel for hybrid IBOC DAB broadcasts. The delay compensates for
the system delay of the digital MPS audio, allowing receivers to blend
between the digital and analog program without a shift in time. In an AM
transmission system, the delayed MPS audio signal 30 is converted by the
exporter to a mono signal and sent directly to the STL as part of the
exciter link data 52.

[0041]The EASU 22 accepts MPS audio 42 from the studio automation
equipment, rate converts it to the proper system clock, and outputs two
copies of the signal, one digital (26) and one analog (28). The EASU
includes a GPS receiver that is connected to an antenna 25. The GPS
receiver allows the EASU to derive a master clock signal, which is
synchronized to the exciter's clock by use of GPS units. The EASU
provides the master system clock used by the exporter. The EASU is also
used to bypass (or redirect) the analog MPS audio from being passed
through the exporter in the event the exporter has a catastrophic fault
and is no longer operational. The bypassed audio 32 can be fed directly
into the STL transmitter, eliminating a dead-air event.

[0042]STL transmitter 48 receives delayed analog MPS audio 50 and exciter
link data 52. It outputs exciter link data and delayed analog MPS audio
over STL link 14, which may be either unidirectional or bidirectional.
The STL link may be a digital microwave or Ethernet link, for example,
and may use the standard User Datagram Protocol or the standard TCP/IP.

[0043]The transmitter site includes an STL receiver 54, an exciter 56 and
an analog exciter 60. The STL receiver 54 receives exciter link data,
including audio and data signals as well as command and control messages,
over the STL link 14. The-exciter link data is passed to the exciter 56,
which produces the IBOC DAB waveform. The exciter includes a host
processor, digital up-converter, RF up-converter, and exgine subsystem
58. The exgine accepts exciter link data and modulates the digital
portion of the IBOC DAB waveform. The digital up-converter of exciter 56
converts from digital-to-analog the baseband portion of the exgine
output. The digital-to-analog conversion is based on a GPS clock, common
to that of the exporter's GPS-based clock derived from the EASU. Thus,
the exciter 56 includes a GPS unit and antenna 57. An alternative method
for synchronizing the exporter and exciter clocks can be found in U.S.
patent application Ser. No. 11/081,267 (Publication No. 2006/0209941 A1),
the disclosure of which is hereby incorporated by reference. The RF
up-converter of the exciter up-converts the analog signal to the proper
in-band channel frequency. The up-converted signal is then passed to the
high power amplifier 62 and antenna 64 for broadcast. In an AM
transmission system, the exgine subsystem coherently adds the backup
analog MPS audio to the digital waveform in the hybrid mode; thus, the AM
transmission system does not include the analog exciter 60. In addition,
the exciter 56 produces phase and magnitude information and the analog
signal is output directly to the high power amplifier.

[0044]IBOC DAB signals can be transmitted in both AM and FM radio bands,
using a variety of waveforms. The waveforms include an FM hybrid IBOC DAB
waveform, an FM all-digital IBOC DAB waveform, an AM hybrid IBOC DAB
waveform, and an AM all-digital IBOC DAB waveform.

[0045]FIG. 2 is a schematic representation of a hybrid FM IBOC waveform
70. The waveform includes an analog modulated signal 72 located in the
center of a broadcast channel 74, a first plurality of evenly spaced
orthogonally frequency division multiplexed subcarriers 76 in an upper
sideband 78, and a second plurality of evenly spaced orthogonally
frequency division multiplexed subcarriers 80 in a lower sideband 82. The
digitally modulated subcarriers are divided into partitions and various
subcarriers are designated as reference subcarriers. A frequency
partition is a group of 19 OFDM subcarriers containing 18 data
subcarriers and one reference subcarrier.

[0046]The hybrid waveform includes an analog FM-modulated signal, plus
digitally modulated primary main subcarriers. The subcarriers are located
at evenly spaced frequency locations. The subcarrier locations are
numbered from -546 to +546. In the waveform of FIG. 2, the subcarriers
are at locations +356 to +546 and -356 to -546. Each primary main
sideband is comprised of ten frequency partitions. Subcarriers 546 and
-546, also included in the primary main sidebands, are additional
reference subcarriers. The amplitude of each subcarrier can be scaled by
an amplitude scale factor.

[0047]FIG. 3 is a schematic representation of an extended hybrid FM IBOC
waveform 90. The extended hybrid waveform is created by adding primary
extended sidebands 92, 94 to the primary main sidebands present in the
hybrid waveform. One, two, or four frequency partitions can be added to
the inner edge of each primary main sideband. The extended hybrid
waveform includes the analog FM signal plus digitally modulated primary
main subcarriers (subcarriers +356 to +546 and -356 to -546) and some or
all primary extended subcarriers (subcarriers +280 to +355 and -280 to
-355).

[0049]FIG. 4 is a schematic representation of an all-digital FM IBOC
waveform 100. The all-digital waveform is constructed by disabling the
analog signal, fully expanding the bandwidth of the primary digital
sidebands 102, 104, and adding lower-power secondary sidebands 106, 108
in the spectrum vacated by the analog signal. The all-digital waveform in
the illustrated embodiment includes digitally modulated subcarriers at
subcarrier locations -546 to +546, without an analog FM signal.

[0050]In addition to the ten main frequency partitions, all four extended
frequency partitions are present in each primary sideband of the
all-digital waveform. Each secondary sideband also has ten secondary main
(SM) and four secondary extended (SX) frequency partitions. Unlike the
primary sidebands, however, the secondary main frequency partitions are
mapped nearer to the channel center with the extended frequency
partitions farther from the center.

[0051]Each secondary sideband also supports a small secondary protected
(SP) region 110, 112 including 12 OFDM subcarriers and reference
subcarriers 279 and -279. The sidebands are referred to as "protected"
because they are located in the area of spectrum least likely to be
affected by analog or digital interference. An additional reference
subcarrier is placed at the center of the channel (0). Frequency
partition ordering of the SP region does not apply since the SP region
does not contain frequency partitions.

[0052]Each secondary main sideband spans subcarriers 1 through 190 or -1
through -190. The upper secondary extended sideband includes subcarriers
191 through 266, and the upper secondary protected sideband includes
subcarriers 267 through 278, plus additional reference subcarrier 279.
The lower secondary extended sideband includes subcarriers -191 through
-266, and the lower secondary protected sideband includes subcarriers
-267 through -278, plus additional reference subcarrier -279. The total
frequency span of the entire all-digital spectrum is 396,803 Hz. The
amplitude of each subcarrier can be scaled by an amplitude scale factor.
The secondary sideband amplitude scale factors can be user selectable.
Any one of the four may be selected for application to the secondary
sidebands.

[0053]In each of the waveforms, the digital signal is modulated using
orthogonal frequency division multiplexing (OFDM). OFDM is a parallel
modulation scheme in which the data stream modulates a large number of
orthogonal subcarriers, which are transmitted simultaneously. OFDM is
inherently flexible, readily allowing the mapping of logical channels to
different groups of subcarriers.

[0054]In the hybrid waveform, the digital signal is transmitted in primary
main (PM) sidebands on either side of the analog FM signal in the hybrid
waveform. The power level of each sideband is appreciably below the total
power in the analog FM signal. The analog signal may be monophonic or
stereo, and may include subsidiary communications authorization (SCA)
channels.

[0055]In the extended hybrid waveform, the bandwidth of the hybrid
sidebands can be extended toward the analog FM signal to increase digital
capacity. This additional spectrum, allocated to the inner edge of each
primary main sideband, is termed the primary extended (PX) sideband.

[0056]In the all-digital waveform, the analog signal is removed and the
bandwidth of the primary digital sidebands is fully extended as in the
extended hybrid waveform. In addition, this waveform allows lower-power
digital secondary sidebands to be transmitted in the spectrum vacated by
the analog FM signal.

[0057]FIG. 5 is a schematic representation of an AM hybrid IBOC DAB
waveform 120. The hybrid format includes the conventional AM analog
signal 122 (bandlimited to about ±5 kHz) along with a nearly 30 kHz
wide DAB signal 124. The spectrum is contained within a channel 126
having a bandwidth of about 30 kHz. The channel is divided into upper 130
and lower 132 frequency bands. The upper band extends from the center
frequency of the channel to about ±15 kHz from the center frequency.
The lower band extends from the center frequency to about -15 kHz from
the center frequency.

[0058]The AM hybrid IBOC DAB signal format in one example comprises the
analog modulated carrier signal 134 plus OFDM subcarrier locations
spanning the upper and lower bands. Coded digital information
representative of the audio or data signals to be transmitted (program
material), is transmitted on the subcarriers. The symbol rate is less
than the subcarrier spacing due to a guard time between symbols.

[0059]As shown in FIG. 5, the upper band is divided into a primary section
136, a secondary section 138, and a tertiary section 144. The lower band
is divided into a primary section 140, a secondary section 142, and a
tertiary section 143. For the purpose of this explanation, the tertiary
sections 143 and 144 can be considered to include a plurality of groups
of subcarriers labeled 146, 148, 150 and 152 in FIG. 5. Subcarriers
within the tertiary sections that are positioned near the center of the
channel are referred to as inner subcarriers, and subcarriers within the
tertiary sections that are positioned farther from the center of the
channel are referred to as outer subcarriers. In this example, the power
level of the inner subcarriers in groups 148 and 150 is shown to decrease
linearly with frequency spacing from the center frequency. The remaining
groups of subcarriers 146 and 152 in the tertiary sections have
substantially constant power levels. FIG. 5 also shows two reference
subcarriers 154 and 156 for system control, whose levels are fixed at a
value that is different from the other sidebands.

[0060]The power of subcarriers in the digital sidebands is significantly
below the total power in the analog AM signal. The level of each OFDM
subcarrier within a given primary or secondary section is fixed at a
constant value. Primary or secondary sections may be scaled relative to
each other. In addition, status and control information is transmitted on
reference subcarriers located on either side of the main carrier. A
separate logical channel, such as an IBOC Data Service (IDS) channel can
be transmitted in individual subcarriers just above and below the
frequency edges of the upper and lower secondary sidebands. The power
level of each primary OFDM subcarrier is fixed relative to the
unmodulated main analog carrier. However, the power level of the
secondary subcarriers, logical channel subcarriers, and tertiary
subcarriers is adjustable.

[0061]Using the modulation format of FIG. 5, the analog modulated carrier
and the digitally modulated subcarriers are transmitted within the
channel mask specified for standard AM broadcasting in the United States.
The hybrid system uses the analog AM signal for tuning and backup.

[0062]FIG. 6 is a schematic representation of the subcarrier assignments
for an all-digital AM IBOC DAB waveform. The all-digital AM IBOC DAB
signal 160 includes first and second groups 162 and 164 of evenly spaced
subcarriers, referred to as the primary subcarriers, that are positioned
in upper and lower bands; 166 and 168. Third and fourth groups 170 and
172 of subcarriers, referred to as secondary and tertiary subcarriers
respectively, are also positioned in upper and lower bands 166 and 168.
Two reference subcarriers 174 and 176 of the third group lie closest to
the center of the channel. Subcarriers 178 and 180 can be used to
transmit program information data.

[0063]FIG. 7 is a simplified functional block diagram of an AM IBOC DAB
receiver 200. The receiver includes an input 202 connected to an antenna
204, a tuner or front end 206, and a digital down converter 208 for
producing a baseband signal on line 210. An analog demodulator 212
demodulates the analog modulated portion of the baseband signal to
produce an analog audio signal on line 214. A digital demodulator 216
demodulates the digitally modulated portion of the baseband signal. Then
the digital signal is deinterleaved by a deinterleaver 218, and decoded
by a Viterbi decoder 220. A service demultiplexer 222 separates main and
supplemental program signals from data signals. A processor 224 processes
the program signals to produce a digital audio signal on line 226. The
analog and main digital audio signals are blended as shown in block 228,
or a supplemental digital audio signal is passed through, to produce an
audio output on line 230. A data processor 232 processes the data signals
and produces data output signals on lines 234, 236 and 238. The data
signals can include, for example, a station information service (SIS),
main program service data (MPSD), supplemental program service data
(SPSD), and one or more auxiliary application services (AAS).

[0064]FIG. 8 is a simplified functional block diagram of an FM IBOC DAB
receiver 250. The receiver includes an input 252 connected to an antenna
254 and a tuner or front end 256. A received signal is provided to an
analog-to-digital converter and digital down converter 258 to produce a
baseband signal at output 260 comprising a series of complex signal
samples. The signal samples are complex in that each sample comprises a
"real" component and an "imaginary" component, which is sampled in
quadrature to the real component. An analog demodulator 262 demodulates
the analog modulated portion of the baseband signal to produce an analog
audio signal on line 264. The digitally modulated portion of the sampled
baseband signal is next filtered by sideband isolation filter 266, which
has a pass-band frequency response comprising the collective set of
subcarriers f1-fn present in the received OFDM signal. Filter
268 suppresses the effects of a first-adjacent interferer. Complex signal
298 is routed to the input of acquisition module 296, which acquires or
recovers OFDM symbol timing offset or error and carrier frequency offset
or error from the received OFDM symbols as represented in received
complex signal 298. Acquisition module 296 develops a symbol timing
offset At and carrier frequency offset Δf, as well as status and
control information. The signal is then demodulated (block 272) to
demodulate the digitally modulated portion of the baseband signal. Then
the digital signal is deinterleaved by a deinterleaver 274, and decoded
by a Viterbi decoder 276. A service demultiplexer 278 separates main and
supplemental program signals from data signals. A processor 280 processes
the main and supplemental program signals to produce a digital audio
signal on line 282. The analog and main digital audio signals are blended
as shown in block 284, or the supplemental program signal is passed
through, to produce an audio output on line 286. A data processor 288
processes the data signals and produces data output signals on lines 290,
292 and 294. The data signals can include, for example, a station
information service (SIS), main program service data (MPSD), supplemental
program service data (SPSD), and one or more advanced application
services (AAS).

[0065]In practice, many of the signal processing functions shown in the
receivers of FIGS. 7 and 8 can be implemented using one or more
integrated circuits.

[0066]FIGS. 9a and 9b are diagrams of an IBOC DAB logical protocol stack
from the transmitter perspective. From the receiver perspective, the
logical stack will be traversed in the opposite direction. Most of the
data being passed between the various entities within the protocol stack
are in the form of protocol data units (PDUs). A PDU is a structured data
block that is produced by a specific layer (or process within a layer) of
the protocol stack. The PDUs of a given layer may encapsulate PDUs from
the next higher layer of the stack and/or include content data and
protocol control information originating in the layer (or process)
itself. The PDUs generated by each layer (or process) in the transmitter
protocol stack are inputs to a corresponding layer (or process) in the
receiver protocol stack.

[0067]As shown in FIGS. 9a and 9b, there is a configuration administrator
330, which is a system function that supplies configuration and control
information to the various entities within the protocol stack. The
configuration/control information can include user defined settings, as
well as information generated from within the system such as GPS time and
position. The service interfaces 331 represent the interfaces for all
services except SIS. The service interface may be different for each of
the various types of services. For example, for MPS audio and SPS audio,
the service interface may be an audio card. For MPS data and SPS data the
interfaces may be in the form of different application program interfaces
(APIs). For all other data services the interface is in the form of a
single API. An audio codec 332 encodes both MPS audio and SPS audio to
produce core (Stream 0) and optional enhancement (Stream 1) streams of
MPS and SPS audio encoded packets, which are passed to audio transport
333. Audio codec 332 also relays unused capacity status to other parts of
the system, thus allowing the inclusion of opportunistic data. MPS and
SPS data is processed by program service data (PSD) transport 334 to
produce MPS and SPS data PDUs, which are passed to audio transport 333.
Audio transport 333 receives encoded audio packets and PSD PDUs and
outputs bit streams containing both compressed audio and program service
data. The SIS transport 335 receives SIS data from the configuration
administrator and generates SIS PDUs. A SIS PDU can contain station
identification and location information, program type, as well as
absolute time and position correlated to GPS. The AAS data transport 336
receives AAS data from the service interface, as well as opportunistic
bandwidth data from the audio transport, and generates AAS data PDUs,
which can be based on quality of service parameters. The transport and
encoding functions are collectively referred to as Layer 4 of the
protocol stack and the corresponding transport PDUs are referred to as
Layer 4 PDUs or L4 PDUs. Layer 2, which is the channel multiplex layer,
(337) receives transport PDUs from the SIS transport, AAS data transport,
and audio transport, and formats them into Layer 2 PDUs. A Layer 2 PDU
includes protocol control information and a payload, which can be audio,
data, or a combination of audio and data. Layer 2 PDUs are routed through
the correct logical channels to Layer 1 (338), wherein a logical channel
is a signal path that conducts L1 PDUs through Layer 1 with a specified
grade of service. There are multiple Layer 1 logical channels based on
service mode, wherein a service mode is a specific configuration of
operating parameters specifying throughput, performance level, and
selected logical channels. The number of active Layer 1 logical channels
and the characteristics defining them vary for each service mode. Status
information is also passed between Layer 2 and Layer 1. Layer 1 converts
the PDUs from Layer 2 and system control information into an AM or FM
IBOC DAB waveform for transmission. Layer 1 processing can include
scrambling, channel encoding, interleaving, OFDM subcarrier mapping, and
OFDM signal generation. The output of OFDM signal generation is a
complex, baseband, time domain pulse representing the digital portion of
an IBOC signal for a particular symbol. Discrete symbols are concatenated
to form a continuous time domain waveform, which is modulated to create
an IBOC waveform for transmission.

[0068]FIG. 10 shows the logical protocol stack from the receiver
perspective. An IBOC waveform is received by the physical layer, Layer 1
(560), which demodulates the signal and processes it to separate the
signal into logical channels. The number and kind of logical channels
will depend on the service mode, and may include logical channels P1-P3,
PIDS, S1-S5, and SIDS. Layer 1 produces L1 PDUs corresponding to the
logical channels and sends the PDUs to Layer 2 (565), which demultiplexes
the L1 PDUs to produce SIS PDUs, AAS PDUs, PSD PDUs for the main program
service and any supplemental program services, and Stream 0 (core) audio
PDUs and Stream 1 (optional enhanced) audio PDUs. The SIS PDUs are then
processed by the SIS transport 570 to produce SIS data, the AAS PDUs are
processed by the AAS transport 575 to produce AAS data, and the PSD PDUs
are processed by the PSD transport 580 to produce MPS data (MPSD) and any
SPS data (SPSD). The SIS data, AAS data, MPSD and SPSD are then sent to a
user interface 590. The SIS data, if requested by a user, can then be
displayed. Likewise, MPSD, SPSD, and any text based or graphical AAS data
can be displayed. The Stream 0 and Stream 1 PDUs are processed by Layer
4, comprised of audio transport 590 and audio decoder 595. There may be
up to N audio transports corresponding to the number of programs received
on the IBOC waveform. Each audio transport produces encoded MPS packets
or SPS packets, corresponding to each of the received programs. Layer 4
receives control information from the user interface, including commands
such as to store or play programs, and to seek or scan for radio stations
broadcasting an all-digital or hybrid IBOC signal. Layer 4 also provides
status information to the user interface.

[0069]As previously described, the digital portion of an IBOC signal is
modulated using orthogonal frequency division multiplexing (OFDM).
Referring to FIG. 11a, an OFDM signal used in the present invention is
characterized as a multi-frequency carrier signal comprising the
plurality of equidistantly spaced subcarriers f1-fn. Adjacent
subcarriers, such as f1 and f2, are separated each from the
other by a predetermined frequency increment such that adjacent
subcarriers are orthogonal, each to the other. By orthogonal, it is meant
that when properly Nyquist weighted, the subcarriers exhibit no
crosstalk. In one hybrid system incorporating the instant invention and
using both digital and analog transmission channels, there are 191
carriers in each sideband with a 70 kHz bandwidth for each sideband. In
one all-digital implementation of the instant invention there are 267
carriers in each sideband with a 97 kHz bandwidth for each sideband.

[0070]FIG. 11b shows an OFDM symbol 5 in the time domain. The symbol has
an effective symbol period or temporal width T, and a full symbol period
T.sub.α. The OFDM subcarrier orthogonality requirement creates a
functional interdependency between the effective symbol period T and the
frequency spacing between adjacent OFDM subcarriers. Specifically, the
frequency separation between adjacent subcarriers is constrained to be
equivalent to the inverse of the effective symbol period T of each OFDM
symbol 5. That is, the frequency separation is equal to 1/T. Extending
across the effective symbol period T of each OFDM symbol 5 is a
predetermined number N of equidistantly spaced temporal symbol samples
(not shown in the figure). Further, extending across the full period
T.sub.α, of each OFDM symbol 5 are a predetermined number
N.sub.α=N(1+α) of equidistantly spaced temporal symbol
samples. α is the amplitude tapering factor for the symbol, and can
be considered here as a fractional multiplier. During modulation, an OFDM
modulator generates a series of OFDM symbols 5, each of which comprises a
predetermined number of temporal symbol samples N.sub.α
corresponding to full symbol period T.sub.α, wherein the first
αN samples and the last αN samples of each symbol are tapered
and have equal phases. In one embodiment, the predetermined number
N.sub.α of temporal samples extending across each full symbol
period T.sub.α, is 1080, the predetermined number N of temporal
samples extending across each effective symbol period T is 1024, and the
number of samples in each of the first αN samples and last αN
samples is 56. These values are merely exemplary and may be varied in
accordance with system requirements. Also during modulation, a cyclic
prefix is applied such that the leading and trailing portions of each
transmitted symbol are highly correlated.

[0071]Predetermined amplitude-time profile or envelope 11, 15, 13 is
imposed upon the signal levels of these samples. This amplitude profile
includes symmetrically ascending and descending amplitude tapers 11, 15
at the leading portion and trailing portion of each symbol 5,
respectively, and a flat amplitude profile 13 extending therebetween.
These rounded or tapered edges provided in the time domain serve to
substantially reduce undesirable side-lobe energy in the frequency
domain, to thus provide a more spectrally efficient OFDM signal. Although
the full symbol period T.sub.α of symbol 5 extends beyond the
effective symbol period T, orthogonality between adjacent subcarriers in
the frequency domain (FIG. 11a) is not compromised so long as amplitude
tapers 11, 15 of symbol 5 follow a Nyquist or raised-cosine tapering
function. More specifically, orthogonality is maintained in the present
invention through a combination of root-raised cosine weighting (or
amplitude tapering) of transmitted symbols and root-raised cosine matched
filtering of received symbols.

[0072]The leading and trailing portions of OFDM symbol 5 share an
additional important feature, namely, the first αN OFDM symbol
samples extending across the leading portion of OFDM symbol 5, which has
a temporal duration αT, have substantially equivalent phases as the
last αN symbol samples extending across the trailing portion of
OFDM symbol 5, which also has a temporal duration αT. Note again
that α is the amplitude tapering factor for the symbol, and can be
considered here as a fractional multiplier.

Acquisition Module Structure and Operation

[0073]One embodiment of a basic acquisition module 296, described in U.S.
Pat. Nos. 6,539,063 and 6,891,898, is shown in FIG. 12. Received complex
signal 298 is provided to the input of peak development module 1100,
which provides the first stage of signal processing for acquiring the
symbol timing offset of the received OFDM signal. Peak development module
1100 develops a boundary signal 1300 at an output thereof, which has a
plurality of signal peaks therein, each signal peak representing a
received symbol boundary position for each received OFDM symbol
represented in received signal 298, input to peak development module
1100. Because these signal peaks represent received symbol boundary
positions, their temporal positions are indicative of received symbol
timing offset. More specifically, because the receiver has no initial or
a priori knowledge of the true or actual received symbol boundary
position, such a position is initially assumed or arbitrarily created to
enable receiver processing to operate. Acquisition module 296 establishes
the symbol timing offset At that exists between this a priori assumption
and the true, received symbol boundary position, thus enabling the
receiver to recover and track symbol timing.

[0074]In developing the signal peaks representing OFDM symbol boundaries,
peak development module 1100 exploits the cyclic prefix applied by the
transmitter, as well as the predetermined amplitude tapering and phase
properties inherent in the leading and trailing portions of each received
OFDM symbol. Particularly, complex conjugate products are formed between
the current sample and the sample preceding it by N samples. Such
products, formed between the first αN samples and the last αN
samples in each symbol, produce a signal peak corresponding to each
symbol comprising the αN conjugate products so formed.

[0075]Mathematically, the formation of the conjugate products is
represented as follows. Let D(t) denote the received OFDM signal, and let
T.sub.α=(1+α)T denote the full OFDM symbol duration or period
where 1/T is the OFDM channel spacing and cc is the amplitude tapering
factor for the symbol. The signal peaks in boundary signal 1300 appear as
a train of pulses or signal peaks in the conjugate products of
D(t)D*(t-T). As a result of the Nyquist amplitude tapering imposed on the
leading and trailing portions of each OFDM symbol, each of the pulses or
signal peaks has a half-sine-wave amplitude profile of the form

w(t)={1/2 sin (πt(αT)), for 0≦t≦αT, and

w(t)={0, otherwise.

[0076]Further, the periodicity of signal 1300, that is, the period of the
train of signal peaks, is T.sub.α. Referring to FIG. 11c, the train
of signal peaks included in boundary signal 1300 has amplitude envelope
w(t) and the peaks are spaced by a period of T.sub.α. Referring to
FIG. 11d, the product of the overlapping leading and trailing portion
amplitude tapers 11, 15 multiplies the squared magnitudes in the
conjugate products, resulting in the half-sine-wave, w(t) which has a
durational width αT corresponding to αN samples.

[0077]Returning again to FIG. 12, for each signal sample input to peak
development module 1100, one product sample is output from multiplier
circuit 1250 representing a conjugate product between that input sample
and a predecessor sample, spaced T samples therefrom. Complex conjugate
developer 1200 produces at its output the complex conjugate of each input
sample, which output is provided as one input to multiplier 1250. The
conjugate samples at this output are multiplied against the delayed
sample output from delay circuit 1150. In this way, complex conjugate
products are formed between the received signal 298 and a delayed replica
thereof obtained by delaying the received signal 298 by the predetermined
time T using delay circuit 1150.

[0078]Referring to FIGS. 13a, 13b, and 13c, the relevant symbol timing for
peak development module 1100 is illustrated. FIG. 13a represents
consecutive OFDM symbols 1 and 2 provided at the input to peak
development module 1100. FIG. 13b illustrates the delayed versions of
OFDM symbols 1 and 2 as output from delay circuit 1150. FIG. 13c
represents the signal peak developed for each corresponding set of
N.sub.α=N(1+α) product samples (which in one working
embodiment equals 1080 samples), the train of signal peaks being produced
responsive to the conjugate multiplication between the received signal of
FIG. 13a and the delayed version thereof in FIG. 13b.

[0079]By way of specific example, if the received OFDM symbol period
T.sub.α corresponds to N.sub.α=1080 signal samples, and the
αN samples at each of the leading and trailing portions of the
symbol correspond to 56 signal samples, then for each 1080-sample OFDM
symbol input to peak development module 1100, there appears a
corresponding set of 1080 product samples in boundary signal 1300. In
this example, delay circuit 1150 imparts a 1024-(N) sample delay so that
each sample input to multiplier 1250 is multiplied by its predecessor
1024 samples away. The signal peak so developed for each corresponding
set of 1080 product samples comprises only 56 conjugate products formed
between the first and last 56 samples of each corresponding symbol.

[0080]Peak development module 1100 can be implemented in any number of
ways as long as the correspondence between the leading and trailing
portions of each symbol is exploited in the manner previously described.
For instance, peak development module 1100 may operate on each sample as
it arrives, so that for each sample in, a product sample is provided at
the output thereof. Alternatively, a plurality of samples may be stored,
such as in vector form, thus creating present sample vectors and delayed
sample vectors, which vectors can be input to a vector multiplier to form
vector product samples at an output thereof. Alternatively, the peak
development module can be implemented to operate on continuous rather
than sampled discrete time signals. However, in such an approach, it
would be desirable that input received signal 298 also be a continuous
rather than a sampled signal.

[0081]Ideally, boundary signal 1300 has easily identifiable signal peaks
therein, as illustrated in FIGS. 11c and 13c. However, in reality, each
signal peak is virtually indistinguishable from the undesired noisy
products of samples lying in adjacent symbols. Since peak development
module 1100 continually forms products between samples extending across
each received symbol and predecessor samples delayed therefrom, boundary
signal 1300 includes both desired signal peaks as well as the noisy
conjugate products. For example, the first αN (56) samples in each
symbol are multiplied against the last αN samples therein, to
produce the desired signal peak αN samples in duration. However,
the remaining N (1024) samples are multiplied against N samples from the
adjacent symbol responsive to the delay imparted thereto by delay circuit
1150 (see FIG. 13). These additional unwanted products have the effect of
filling in noise between the occurrences of the desired signal peaks.
Thus, noisy products corresponding to OFDM signals can be appreciable.

[0082]In addition to the presence of the aforementioned product noise in
boundary signal 1300, there is noise derived from other sources well
known in the art of digital communications. Such noise is imparted to the
signal during propagation thereof through the atmosphere by ambient
noise, scattering, multipath and fading, and signal interferences. The
front end of the receiver also adds noise to the signal.

[0083]Subsequent signal processing stages are dedicated, in part, to
combating the depreciating effect of the aforementioned noise with
respect to the desired signal peaks in boundary signal 1300, or more
specifically, to improve the signal-to-noise ratio of the signal peaks
present in boundary signal 1300. Signal enhancing module 1350 is provided
at the output of peak development module 1100, and comprises first and
second stage signal enhancing circuits or modules. The first stage signal
enhancing circuit is an additive superposition circuit or module 1400 and
the second stage enhancing circuit is a matched filter 1450, provided at
the output of the first stage enhancing circuit.

[0084]Additive superposition circuit 1400 additively superimposes a
predetermined number of signal peaks and their surrounding noisy
products, to enhance signal peak detectability by increasing the
signal-to-noise ratio of the signal peaks in boundary signal 1300. To
implement this process of additive superposition, a predetermined number
of consecutive segments of boundary signal 1300 are first superimposed or
overlapped in time. Each of these superimposed segments comprises a
symbol period's worth of conjugate product samples as are output from
peak development module 1100, and includes a desired signal peak
surrounded by undesired noisy product samples.

[0085]After the predetermined number or block of signal segments have been
time overlapped, the product samples occupying a predetermined temporal
position in the superimposed set of segments are accumulated to form a
cumulative signal sample for that predetermined position. In this way, a
cumulative signal is developed comprising a cumulative signal sample for
each of the predetermined sample positions extending across the
superimposed boundary signal segments.

[0086]If, for example, 32 contiguous boundary signal segments are to be
superimposed, and if each segment includes a symbol period's worth of
1080 samples, then additive superposition circuit 1400 produces 1080
cumulative samples for each contiguous block of 32 segments (1080 samples
per segment) input thereto. In this manner, the conjugate products of 32
segments (each segment including 1080 samples, a signal peak and noise
therein) are additively superimposed or "folded" on top of one another,
by pointwise adding the superimposed conjugate products of the 32
segments. Essentially, in this folding process, the products of the 32
segments are pointwise added to corresponding conjugate products one
symbol period (or 1080 samples) away, over the 32 contiguous symbols, to
produce a cumulative signal segment comprising 1080 cumulative samples
therein. The signal processing is then repeated for the next contiguous
block of 32 boundary signal segments, to produce another cumulative
signal segment, and so on.

[0087]The cumulative signal segment produced by additively superimposing
the predetermined number of contiguous segments of boundary signal 1300
includes an enhanced signal peak therein, which exhibits an increased
signal-to-noise ratio with respect to the signal peaks in each of the
constituent input boundary signal segments. The reason for this
enhancement is that the superposition of the boundary signal segments
aligns their respective signal peaks, so that when the segments are
accumulated, each signal peak adds to the next, thus achieving a form of
coherent processing gain based upon the repetitive nature of the boundary
signal peaks.

[0088]Whereas the aligned, repetitive signal peaks in the boundary signal
segments coherently accumulate to form an enhanced (cumulative) signal
peak at the output of the additive superposition module 1400, by
contrast, the random nature of the noisy conjugate products surrounding
the signal peak in each of the boundary signal segments produce
incoherent addition thereof during the additive superposition process.
Because the signal peaks add coherently and the surrounding noisy
products having zero mean add incoherently and are thus averaged, the
enhanced signal peak output from the additive superposition module 1400
exhibits, overall, an improved signal-to-noise ratio. The processing gain
and signal-to-noise ratio enhancement achieved by the additive
superposition module increases along with the number of boundary signal
segments superimposed to produce the cumulative signal segment.
Offsetting this advantage is a corresponding disadvantageous increase in
acquisition delay, since more boundary signal segments are collected to
produce the cumulative signal peak. Thus, the particular predetermined
number, for instance 16 or 32, represents in any application a balancing
between these two competing interests, wherein the number of averages is
ultimately limited by the fading bandwidth.

[0089]In mathematical terms, the additive superposition of contiguous
segments of the conjugate products present in boundary signal 1300 can be
expressed by the following:

F ( t ) = k = 0 K - 1 D ( t + k T α
) D * ( t - T + k T α )

where k is the number of superimposed segments, D is input 298 to the peak
development module 1100, and K is the number of segments, such as 16, for
example. An important aspect of the foregoing signal processing is that
symbol timing is preserved at each stage thereof: OFDM symbols input to
peak development module 1100, boundary signal segments input to additive
superposition circuit 1400, and cumulative signal segments output
therefrom, each have a temporal period of T.sub.α (corresponding to
N=1080 samples). In this way, symbol timing offset, as indicated by the
positioning of the signal peaks within a signal segment, is preserved
throughout.

[0090]In operation, the additive superposition module 1400, summation
module 1600 and feedback delay module 1650, together provide the additive
superposition functions. That is, summation module 1600 adds a present
input sample to the result of an accumulation of samples in contiguous
symbols, each of the samples being temporally spaced by one symbol period
T.sub.α (corresponding to 1080 samples). Delay 1650 imparts the one
symbol period delay between accumulations. Stated otherwise, each
accumulated result output by summation module 1600 is delayed by 1 symbol
period T.sub.α, and then fed back as an input to summation module
1600, where it is added to the next input sample. The process repeats for
all input samples across each input symbol.

[0091]Stated otherwise, the first cumulative sample in the cumulative
signal segment represents an accumulation of all of the first samples of
all of the 32 boundary signal segments. The second cumulative sample
represents an accumulation of all of the second samples of all of the 32
boundary signal segments, and so on, across the cumulative signal
segment.

[0092]Reset generator 1700 provides a reset signal to delay module 1650
after the predetermined number of signal segments has been accumulated to
produce the cumulative signal segment. For example, if the predetermined
number of boundary signal segments to be accumulated is 32, the reset
generator 1700 asserts a reset to feedback delay module 1650 every 32
signal segments. Responsive to assertion of the reset, additive
superposition module 1400 accumulates the next predetermined number of
contiguous boundary signal segments.

[0093]As previously described, the output of additive superposition module
1400 is a cumulative signal comprising a series of cumulative signal
segments, each segment including an enhanced signal peak 1550 therein. In
a high-noise environment, enhanced signal peak 1550, although exhibiting
an improved signal-to-noise ratio, can still be virtually
indistinguishable from the surrounding noise. Thus, it is desirable to
further enhance the signal-to-noise ratio of the enhanced signal peak.

[0094]To further enhance the signal-to-noise ratio of enhanced signal peak
1550, the cumulative signal output from additive superposition module
1400 is input to matched filter 1450. The temporal impulse response of
matched filter 1450 is matched to the shape or amplitude envelope of the
enhanced signal peak input thereto, and in one embodiment of the present
invention, follows a root-raised cosine profile. Specifically, the
impulse response of the matched filter corresponds to the function w(t),
as shown in FIG. 11d, and is determined by pointwise multiplying the
first αN samples of symbol 5 with the last αN samples
thereof. See FIGS. 11b and 11d.

[0095]Although a non-matched low-pass filter could be used to smooth the
noise present in the cumulative signal, the matched filter 1450 provides
the optimum signal-to-noise improvement for the desired signal, enhanced
signal peak 1550, in a Gaussian noise environment. Matched filter 1450 is
implemented as a finite impulse response (FIR) digital filter that
provides at an output thereof a filtered version of the complex samples
input thereto.

[0096]Briefly summarizing the signal processing stages leading up to the
output of the matched filter, peak development module 1100 produces a
plurality of signal peaks, the temporal positions of which represent
symbol boundary positions which represent symbol timing offset for each
received OFDM symbol. Signal enhancing module 1350 enhances the
detectability of the signal peaks by first additively superimposing a
predetermined number of input signal segments to produce a cumulative
signal segment having an enhanced peak therein, and then second, matched
filtering the cumulative signal segment to produce a cumulative,
matched-filtered signal segment that is optimally ready for subsequent
peak detection processing. This process continually operates to produce a
plurality of filtered enhanced signal peaks at the output of signal
enhancing module 1350. The temporal positions of these filtered enhanced
signal peaks within the match-filtered, cumulative signal segments output
from signal enhancing module 1350, are indicative of symbol boundary
positions or OFDM symbol timing offset.

[0097]Taken individually, and especially in combination, the additive
superposition module and matched filter advantageously enhance signal
peak detectability. Their introduction subsequent to the peak development
stage permits the effective use of an OFDM signal comprising a large
number of frequency carriers, and operating in a propagationally noisy
signal environment.

[0098]The next stage of signal processing required to establish symbol
timing offset is to detect the temporal position of the signal peak
output from signal enhancing module 1350. The temporal position of the
signal peak is, in actuality, the sample index, or sample number, of the
enhanced signal peak within the filtered, cumulative signal segment
output from the matched filter.

[0099]Filtered complex signal 1750 output from matched filter 1450 is
provided as an input to peak selector module 1900, which detects the
enhanced filtered signal peak and the temporal position, or sample index,
thereof. In operation, squared magnitude generator 1950 of peak selector
1900 squares the magnitude of the complex signal samples input thereto to
generate a signal waveform at the output thereof. The output of squared
magnitude generator 1950 is provided as an input to max finder 2000 which
examines the sample magnitudes input thereto and identifies the temporal
position or sample index corresponding to the signal peak.

[0100]This temporal position of the signal peak is provided, essentially,
as the symbol timing offset that is provided by acquisition module 296 to
an input of a symbol timing correction module (not shown). It should be
appreciated that the temporal position provided as the timing offset
Δt may require slight adjustments to compensate for various
processing delays introduced by the preceding signal processing stages.
For example, initialization delays in loading filters, etc., can add
delays that need to be calibrated out of the final timing offset
estimate. However, such delays are generally small and implementation
specific.

[0101]After the temporal position of the signal peak has been determined
(to establish symbol timing offset), the next stage in signal processing
is to determine the carrier phase error and corresponding carrier
frequency error of the received OFDM signal. The matched-filtered,
enhanced signal peak in complex signal 1750 represents the cleanest
point, or point of maximum signal-to-noise ratio, at which to determine
the carrier phase error and frequency error. The phase of the complex
sample at this peak position gives an indication of the frequency error
existing between the transmitter and receiver, since the conjugate
product at this point, as developed by peak development module 1100,
should have yielded a zero-phase value in the absence of carrier
frequency error. The conjugate product at this point of the signal peak,
and in fact at every other point in the signal peak, should yield a
zero-phase value because, mathematically, the conjugate product between
symbol samples having equivalent phase (as do the samples at the leading
and trailing portions of each received symbol) eliminates phase, in the
absence of carrier frequency error. Any residual phase present at the
peak of the signal output from the matched filter is proportional to
carrier frequency error, and the frequency error is simple to calculate
once the residual phase is determined.

[0102]Mathematically, the carrier frequency error Δf produces the
residual phase shift of 2πΔfT between the samples at the leading
and trailing portions of an OFDM symbol that form a conjugate product
peak. Thus, the frequency error is represented by the following equation:

Δ f = Arg ( G Max ) 2 π T

where GMax is the peak of the matched filter output and Arg denotes
the argument (phase) of a complex number--the complex sample--at the
signal peak. The Arg function is equivalent to the four quadrant
arctangent. Since the arctangent cannot detect angles outside of a 2π
window, the frequency estimate is ambiguous up to a multiple of the
channel spacing, 1/T. Nevertheless, this frequency error estimate,
together with the timing offset estimate provided by the location of the
signal peak, is sufficient to allow the commencement of symbol
demodulation. As demodulation proceeds, subsequent receiver frame
boundary processing, not part of the present invention, resolves the
frequency ambiguity.

[0103]In FIG. 12, both the matched-filtered, complex signal 1750 and the
temporal position or sample index, are provided as inputs to phase
extractor 2050. Phase extractor 2050 extracts the residual phase from the
complex sample representing the enhanced signal peak output from the
matched filter. The extracted phase is provided to the input of frequency
generator 2100, which simply scales the extracted phase input thereto to
produce the carrier frequency error Δf which is then provided by
acquisition module 296 to a frequency correction module (not shown).
Thus, the temporal position of the filtered signal peak provided at the
output of matched filter 1450 is indicative of symbol timing offset, and
from the phase of this signal peak, carrier frequency error is derived.

FM Digital Signal Quality Metric

[0104]The foregoing method and apparatus for acquiring or recovering
symbol timing offset and carrier frequency error from a received OFDM
signal provide a basic technique for determining unqualified symbol
timing offset and carrier frequency error. U.S. Pat. Nos. 6,539,063 and
6,891,898 describe additional techniques for acquiring or recovering
symbol timing offset and carrier frequency error from a received OFDM
signal, any of which may be used to implement a digital signal quality
metric according to the present invention. Because the acquisition
function as described in these patents is a time-domain process that
occurs near the start of the baseband processing chain and before OFDM
demodulation, it can be exploited to provide an effective digital signal
quality metric.

[0105]Moreover, the predetermined amplitude and phase properties described
above and inherent in the leading and trailing portions of the OFDM
symbol, namely, the tapering of sample amplitudes in the leading and
trailing portions of each OFDM symbol and the equivalent phases thereof,
are advantageously exploited by existing IBOC systems in order to
efficiently acquire OFDM symbol timing and frequency in the receiver.
These properties can be used according to the present invention for
implementing a digital signal quality metric. Thus, in one aspect, this
invention utilizes these symbol characteristics to provide a digital
signal quality metric using a previously existing FM acquisition module.

[0106]Preferably, the acquisition algorithm used for the digital signal
quality metric is comprised of two operations: pre-acquisition filtering
and acquisition processing. Pre-acquisition filtering is used to prevent
falsely acquiring on large second-adjacent channels. Each primary
sideband is filtered prior to acquisition processing. In one example, the
pre-acquisition filter is an 85-tap finite impulse response (FIR) filter,
designed to provide 40 dB stopband rejection while limiting the impact on
the desired primary sideband. Existing pre-acquisition filters can be
completely reused, without modification, when calculating the quality
metric of this invention. After the input samples have been filtered,
they are passed to the acquisition processing functional component.

[0107]The acquisition processing functional component takes advantage of
correlation within the symbol resulting from the cycle prefix applied to
each symbol by the transmitter to construct acquisition peaks. As
previously described, the position of the peaks indicates the location of
the true symbol boundary within the input samples, while the phase of the
peaks is used to derive the frequency error. Moreover, frequency
diversity can be achieved by independently processing the upper and lower
primary sidebands of the digital radio signal.

[0108]Each of the symbols includes a plurality of samples. The inputs to
acquisition processing are blocks of upper and lower primary sideband
samples. In one example, each block is comprised of 940 real or imaginary
samples, at a rate of 372,093.75 samples per second.

[0109]The acquisition algorithm as modified for calculating a digital
signal quality metric is shown in FIGS. 14 and 19. Referring first to
FIG. 14, 940-sample filtered data blocks are buffered into 1080-sample
symbols, as shown in block 370. As previously described, the first and
last 56 samples of each transmitted symbol are highly correlated due to
the cyclic prefix. Acquisition processing reveals this correlation by
complex-conjugate multiplying each sample in an arbitrary symbol with its
predecessor 1024 samples away (block 372). To enhance the detectability
of the resulting 56-sample peak, the corresponding products of 16
contiguous symbols are "folded" on top of one another to form a
1080-sample acquisition block (block 374). Sixteen symbols are used in
this embodiment, instead of the 32 symbols as described with respect to
the previously described acquisition methods, in order to expedite
calculation of the digital signal quality metric, but fewer symbols such
as 8 may be desirable and any other suitable number of symbols may be
used.

[0110]The 56-sample folded peak, although visible within the acquisition
block, is very noisy. Therefore, block 376 shows that it is smoothed with
a 57-tap FIR filter whose impulse response is matched to the shape of the
peak:

y [ n ] = k = 0 56 x [ n + 57 - k ] h [ k ]
for n = 0 , 1 , , 1079

where n is the output sample index, x is the matched-filter input, y is
the matched-filtered output, and h[k] is the filter impulse response,
defined below.

h [ k ] = cos ( - π 2 + k π 56 ) for k =
0 , 1 , 56 .

[0111]Taking the magnitude squared of the matched-filtered outputs (block
378) simplifies symbol boundary detection by converting complex values to
real values. This computation increases the dynamic range of the input,
making the symbol boundary peak even less ambiguous and allowing the peak
search to be conducted over a single dimension (versus two dimensions for
the I and Q values). The magnitude-squared calculation is:

y[n]=I[n]2+Q[n]2 for n=0,1, . . . , 1079

where I is the real portion of the input, Q is the imaginary portion of
the input, y is the magnitude-squared output, and n is the sample index.
The upper sideband and lower sideband matched-filtered, magnitude-squared
output waveforms for each 16-symbol block are used to generate the
digital signal quality metric. As shown in block 380, the acquisition
process continues, as described above, and the quality metric algorithm
continues, as shown in FIG. 19 (block 450).

[0112]The next step in the quality metric algorithm is to calculate a
normalized correlation peak (blocks 452-458) in order to achieve improved
discrimination of the symbol boundary peak. Normalizing the correlation
peak provides a basis for assessing the quality of the signal and
indicates the probability that there is a digital signal present. The
peak value of the normalized correlation peak can range from zero to one,
with a value of one indicating the maximum likelihood that a digital
signal is present. The peak value of the normalized correlation peak
thereby provides a digital signal quality metric.

[0113]Circuitry according to the existing acquisition algorithm for
calculating a correlation peak is shown in box 382 of FIG. 15. The input
384 is a 1080-sample symbol received on either the upper or lower
sideband. The input samples are shifted by 1024 samples 386 and the
complex conjugate 388 of the shifted samples is multiplied 390 by the
input samples. Sixteen symbols are folded as shown by block 392 and adder
394. The folded sums are filtered 396 by root-raised cosine matched
filter and magnitude squared 398 to produce a correlation peak 399. Thus,
the acquisition algorithm finds a symbol boundary by multiplying a
current input sample by the complex conjugate of the input delayed by
1024 samples. At the start of a symbol, the phase of the conjugate
product over the next 56 samples is effectively zero for each OFDM
subcarrier. The constituent OFDM subcarriers combine coherently over this
period, but not over the remainder of samples in the symbol. The result
is a discernible correlation peak 399 after 16 symbols are folded and
matched filtering is applied.

[0114]Referring again to FIG. 19, additional processing steps according to
the present invention are shown. The normalized correlation peak is
determined by first calculating a normalization waveform for each of the
upper and lower sideband waveforms (block 452). This normalization
waveform exploits an amplitude correlation between the first and last 56
samples of an OFDM symbol due to the root-raised cosine pulse shaping
applied at the transmitter. Referring to FIG. 15, block 400 illustrates
the computation of the normalized waveform 416. The magnitude squared 406
of each input symbol is delayed 386 by 1024 samples and added 404 to the
current magnitude-squared samples 402. Sixteen symbols are folded as
shown by block 408 and adder 410. The folded sums are raised-cosine
matched filtered 412, and squared and reciprocated 414 to produce a
normalization waveform 416. The folding and matched filtering of the
normalization waveform is identical to that performed in the existing
acquisition algorithm, except the existing matched filter taps are
squared and halved to ensure proper normalization:

g [ k ] = h [ k ] 2 2 for k = 0 56

where k is the index of taps in the matched filters, h[k] are the existing
taps for the conjugate-multiplied correlation peak, and g[k] are the new
taps for the normalization waveform. After folding the first 16 symbols
and matched filtering, a symbol boundary is apparent. As shown in FIG.
17, the location of the symbol boundary is marked by a reduction in
amplitude of the resultant waveform.

[0115]Referring again to FIG. 19, once the normalization waveform is
calculated, the next step is normalization of the correlation peak, block
458. Normalization of the correlation peak 399 with the normalization
waveform from block 452 enhances the correlation peak by reducing the
level of all samples except those coincident with the symbol boundary.
Referring again to FIG. 15, the correlation peak 399 is multiplied 418 by
the normalization waveform 416 to produce a normalized correlation peak
420. FIG. 18 shows an example of a normalized correlation peak in a
relatively clean environment, where the x-axis represents the sample
number and the y-axis is the normalized correlation value.

[0116]Once the correlation peak is normalized, the next step in the
quality metric algorithm is to find peak indices PU and PL and
peak values QU and QL (FIG. 19, block 460). The peak index is
the sample number corresponding to the maximum value of the normalized
correlation waveform. PU and PL are the peak indices of the
normalized correlation waveform for the upper and lower sidebands,
respectively. Peak value is the maximum value of the normalized
correlation waveform and provides a digital signal quality metric.

[0117]A quality estimate from each sideband can be independently
calculated. The peak values of the normalized correlation waveform are
representative of the relative quality of that sideband:

QU=x(PU)

QL=x(PL)

where x is the normalized correlation waveform, QU is the upper
sideband quality, and QL is the lower sideband quality. Referring to
FIG. 15, the peak index 424 is identified and peak quality value 422 is
calculated for a sideband by 426.

[0118]In order to validate the digital signal quality metric, optionally a
peak index delta can be found and wrapped. The peak index delta compares
the peak indices of the upper and lower sidebands for each sixteen-symbol
block:

=|PU-PL|.

[0119]Because the symbol boundaries are modulo-1080 values, the computed
deltas must be appropriately wrapped to ensure that the minimum
difference is used:

If Δ>540, then Δ=1080-Δ.

[0120]A peak index delta of zero indicates that the peak indices from each
sideband are identical, thereby representing the maximum assurance that
the normalized correlation peaks from each sideband correspond to the
presence of a valid digital signal.

[0121]As an additional method for validating the digital signal quality
metric, optionally a frequency offset difference can be calculated for
the upper and lower sidebands. According to the previously described
acquisition algorithm, the phase of the complex sample at the peak
position of signal 1750 gives an indication of the frequency error
existing between the transmitter and receiver, since the conjugate
product at this point, as developed by peak development module 1100,
should have yielded a zero-phase value in the absence of carrier
frequency error. The conjugate product at this point of the signal peak,
and in fact at every other point in the signal peak, should yield a
zero-phase value because, mathematically, the conjugate product between
symbol samples having equivalent phase (as do the samples at the leading
and trailing portions of each received symbol) eliminates phase, in the
absence of carrier frequency error. Any residual phase present at the
peak of the signal output from the matched filter is proportional to
carrier frequency error, and the frequency error is simple to calculate
once the residual phase is determined. The range of frequency offset
measured on either sideband is ±1/2 FFT bin spacing, which is
equivalent to ±1/(2T), for a channel spacing of 1/T, as shown in FIG.
11a. If the frequency offset estimated difference between the upper and
lower sidebands is within a certain threshold, such as ± 1/16 FFT bin
spacing, for example, then it is unlikely that any adjacent interferer
has the same frequency offset (as well as peak index) as the desired
signal of interest. As such, the frequency offset difference indicates
that the detected signal is in fact the desired signal of interest.

[0122]Referring to FIG. 16, the peak values and indices from the
individual sidebands (FIG. 15, items 422 and 424) are combined to produce
the peak delta and quality estimates. The peak correlation value 430 from
the upper sideband signal processing is representative of the upper
sideband signal quality. The peak correlation value 432 from the lower
sideband signal processing is representative of the lower sideband signal
quality. Optionally, the difference between the peak index 434 from the
upper sideband signal processing and the peak index 436 from the lower
sideband signal processing is determined by subtracting one index from
the other as shown by subtraction point 438. The absolute value of the
difference is determined (block 440) and the signal is wrapped to
≦540 samples (block 442) to produce a peak index delta 444. The
signal is wrapped to ≦540 samples because the symbol boundary
offset is modulo-1/2 symbol, meaning that the distance to the nearest
symbol boundary is always ≦540 samples.

[0123]Once the peak index delta and quality estimates have been computed,
optionally they can be compared to thresholds in order to implement
appropriate decision rules. The quality for each individual sideband can
be separately compared to a threshold, in addition to optionally
evaluating the peak index delta and sum of the quality estimates from
both sidebands. This allows for a quality assessment of a signal even
when one of its sidebands has been destroyed by interference. In
addition, a quality status parameter reflecting different levels of
sensitivity can be used. In one example, the quality status parameter is
a 2-bit value that indicates to the host controller of a digital radio
receiver the quality of the currently tuned channel. In this example, the
quality of the received signal increases as the status bits change from
00→11. This allows receiver manufacturers to have the ability to
adjust the sensitivity of the quality algorithm by varying the threshold
of the quality status bits.

[0124]The digital signal quality metric can also be used to generate a
visual indication on a receiver's display of the quality of a received
signal. Presently, a series of bars known as a digital audio availability
indicator (DAAI) indicate the strength of a received digital signal. The
status bits of the quality status parameter can be correlated to the
number and size of the bars in such an indicator.

[0125]As will be appreciated from reading the above description, the
simplicity of the algorithm of this invention limits the required changes
to previously known receivers. The extent of the impacts on the baseband
processor and host controller of the receiver are as follows.

[0126]While processing the first acquisition block, the baseband processor
must now calculate the normalization waveform, as illustrated in FIG. 15.
This entails computing the magnitude squared of both the current
1080-sample input symbol and a 1024-sample delayed version, adding the
magnitude-squared vectors, accumulating the sum over 16 symbols, matched
filtering it, and squaring the resulting vector. Besides the increase in
MIPS (million instructions per second), additional memory must be
allocated for the delay, accumulation, and FIR-filtering operations.
Other changes include normalizing the correlation peak via vector
division, finding the peak value and index of the normalized correlation
peak, and computing the peak index delta. The baseband processor may then
apply a decision rule and appropriately set quality status parameter
based on the digital signal quality metric.

[0127]The digital signal quality metric can be applied to many areas of
interest, such as an FM seek-scan function, resolution of 300-kHz-spaced
interferers, first adjacent interferer sideband selection, and diversity
switching, for example. The algorithm has been implemented in a reference
receiver, and tested in a variety of environments over a range of
carrier-to-noise ratios, in order to implement a digital seek-scan
function. Specifically, performance was tested within a number of dB of
digital audio threshold in additive white Gaussian noise (AWGN), AWGN
with one sideband, urban fast (UF) Rayleigh fading, and UF Rayleigh
fading with a -6 dB first-adjacent signal.

[0128]At each point, at least 300 re-acquisitions were forced. The peak
index delta and quality estimates were logged for each attempt, and the
following decision rule was applied:

(QU≧TQ)

OR

(QL≧TQ)

OR

(QL+QU≧TQ+0.2 AND Δ≦T.sub.Δ).

[0129]The probability of stopping was then computed and plotted over a
range of TQ and T.sub.Δ, to allow judicious selection of those
thresholds.

[0130]The probability of stopping versus the carrier-to-noise ratio Cd/No
in the various environments is shown in FIG. 20 through FIG. 23 with
T.sub.Δ=8 and TQ ranging from 0.4 to 0.6. Over this range of
thresholds, the probability of stopping with no input signal is virtually
nil. In each figure, digital audio threshold is indicated by a vertical
line 500.

[0131]After reviewing the plots in FIG. 20 through FIG. 23, the
recommended default thresholds were set as follows: T.sub.Δ=8,
TQ=0.5. Over all environments and carrier-to-noise ratios, these
thresholds yield the performance that best minimizes the probability of
missing strong stations while simultaneously minimizing the probability
of falsely stopping on a weak signal. The probability of stopping in the
various environments using these default thresholds is shown in FIG. 24.
On each curve, digital audio threshold is depicted by a square.

[0132]The curves in FIG. 24 indicate that performance in AWGN is quite
good. At high carrier-to-noise ratios, the probability of detection is
high. Likewise, at low values of Cd/No, the false alarm rate is very low.
The steep transition region around digital audio threshold is desirable.
In a fading environment, a longer dwell time can be employed to reduce
false alarms, at the expense of increased band scanning durations.

[0133]This invention provides a method and apparatus that provides fast
and accurate seek and scan functions for detecting the presence of an FM
digital HD Radio® signal. The algorithm could be merged with the
existing analog FM seek and scan techniques to provide an improved
approach to general FM seek and scan functions (for analog, hybrid, and
all-digital signals). The methods described herein may be implemented
utilizing either a software-programmable digital signal processor, or a
programmable/hardwired logic device, or any other combination of hardware
and software sufficient to carry out the described functionality.

[0134]While the present invention has been described in terms of its
preferred embodiment, it will be understood by those skilled in the art
that various modifications can be made to the disclosed embodiment
without departing from the scope of the invention as set forth in the
claims.